Data Sheet
December 1999

CSP1027 Voice Band Codec for

Cellular Handset and Modem Applications

1 Features

(delta-sigma) A/D and D/A converters with stan-

dard 16-bit serial I/O interface.

On-chip filters meet ITU-T G.712 voice band fre-
quency response and signal to distortion plus noise
specifications. Suitable for IS-54, GSM, and JDC dig-
ital cellular applications.

Low-profile package (<1.5 mm) 48-pin thin quad flat
pack (TQFP) available or 44-pin EIAJ quad flat pack
(QFP).

Operates in systems with a 3.0 V to 5.0 V digital
power supply and a 5.0 V analog supply.

Low-power 0.9 µm CMOS technology, fully static
design, typical power of 68 mW when active and
0.05 mW in standby with a 3.3 V digital supply and a
5.0 V analog supply.

A low-power inactive (standby) state without stopping
clock or removing power supply.

Sampling rates up to 24 kHz.

On-chip programmable sampling clock generator
allows input clock to be an integer multiple of
125 times the sampling rate or an integer multiple of
the sampling rate.

Programmable phase adjust of both codec sampling
clock and baseband codec clock.

Two on-chip clock dividers for generating the output
clock for the baseband codec and the output clock
for other processors.

Regulated microphone power supply.

Microphone preamplifier, with programmable input
ranges of 0.16 Vp and 0.5 Vp.

Output amplifier, with programmable gain settings,
0 dB to 45 dB in 3 dB steps.

High-pass filters selectable via control registers.

Power-on reset pulse generator.

Standard 16-bit serial I/O interface.

Serial I/O multiprocessor mode compatible with the
Lucent Technologies Microelectronics Group's
DSP16A and DSP1610/1616/1617/1618 Digital Sig-
nal Processors.

2 Description

The Lucent CSP1027 is a high-precision linear voice-
band

(delta-sigma) codec designed for cellular

handset and modem applications. The device is fabri-
cated in low-power CMOS technology and designed for
low-voltage (3.0 V to 5.0 V) digital systems. The
CSP1027 is packaged in a 44-pin EIAJ quad flat pack
(QFP) or a 48-pin EIAJ thin quad flat pack (TQFP). In
the 48-pin TQFP, the CSP1027 occupies a total volume
of 0.0784 cm

The CSP1027 has a variety of significant programma-
ble features not found in standard voice band codecs.
The analog interface includes a microphone preampli-
fier with programmable gain settings, an output ampli-
fier with gain programmable in 3 dB steps over a 45 dB
range, and a regulated microphone power supply. An
inactive mode allows a low-power standby state, and a
mute function provides suppression of the analog out-
put. On-chip antialiasing and anti-imaging filtering
includes a selectable high-pass filter. The CSP1027
meets ITU-T G.712 voice band specifications.

The programmable features of the CSP1027 are set by
writing four on-chip control registers through the serial
I/O interface. The codec's digital input/output uses a
linear 16-bit two's complement data format that is also
transferred through the serial I/O interface. The
CSP1027 interfaces easily to the 16-bit serial ports of
digital signal processors and other devices. The serial
interface supports the Lucent fixed-point DSP family
serial multiprocessor mode. This allows up to eight
compatible devices, including two CSP1027s, to inter-
face to each other on a common 4-wire bus using a
time-division-multiplexing scheme.

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

Table of Contents

Contents

Page

Features ...................................................................................................................................................... 1

Description .................................................................................................................................................. 1

Pin Information ........................................................................................................................................... 3

Architectural Information ............................................................................................................................ 5
4.1

Overview........................................................................................................................................... 6

4.2

Description of Signal Paths............................................................................................................... 6

4.3

Programmable Features ................................................................................................................. 13

4.4

Power-On Reset ............................................................................................................................. 14

4.5

Clock Generation ............................................................................................................................ 16

4.6

Serial I/O Configurations................................................................................................................. 20

Register Information.................................................................................................................................. 26
5.1

Codec I/O Control 0 (cioc0) Register ............................................................................................. 26

5.2

Codec I/O Control 1 (cioc1) Register ............................................................................................. 27

5.3

Codec I/O Control 2 (cioc2) Register ............................................................................................. 28

5.4

Codec I/O Control 3 (cioc3) Register ............................................................................................. 29

Signal Descriptions ................................................................................................................................... 30
6.1

Clock Interface................................................................................................................................ 30

6.2

Reset Interface ............................................................................................................................... 31

6.3

Serial I/O Interface.......................................................................................................................... 31

6.4

External Gain Control Interface ...................................................................................................... 32

6.5

Digital Power and Ground............................................................................................................... 32

6.6

Analog Interface.............................................................................................................................. 32

6.7

Analog Power and Ground ............................................................................................................. 32

Application Information ............................................................................................................................. 33
7.1

Analog Information.......................................................................................................................... 33

7.2

Power Supply Configuration ........................................................................................................... 36

7.3

The Need for Fully Synchronous Operation ................................................................................... 36

7.4

Crystal Oscillator............................................................................................................................. 38

7.5

Programmable Clock Generation ................................................................................................... 45

Device Characteristics .............................................................................................................................. 47
8.1

Absolute Maximum Ratings ............................................................................................................ 47

8.2

Handling Precautions...................................................................................................................... 47

8.3

Recommended Operating Conditions............................................................................................. 47

Electrical Characteristics and Requirements ............................................................................................ 48
9.1

Power Dissipation ........................................................................................................................... 50

10 Analog Characteristics and Requirements................................................................................................ 51

10.1

Analog Input and Microphone Regulator ........................................................................................ 51

10.2

Analog-to-Digital Path..................................................................................................................... 52

10.3

Digital-to-Analog Path..................................................................................................................... 53

10.4

Miscellaneous ................................................................................................................................. 54

11 Timing Characteristics and Requirements ................................................................................................ 55

11.1

Clock Generation ............................................................................................................................ 56

11.2

Power-On Reset ............................................................................................................................. 57

11.3

Reset .............................................................................................................................................. 58

11.4

Serial I/O Communication .............................................................................................................. 59

11.5

Serial Multiprocessor Communication ............................................................................................ 61

12 Outline Diagrams ...................................................................................................................................... 62

12.1

44-Pin EIAJ Quad Flat Pack (QFP) ................................................................................................ 62

12.2

48-Pin EIAJ Thin Quad Flat Pack (TQFP) ...................................................................................... 63

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

3 Pin Information

(continued)

Functional descriptions of the pins are found in Section 6 on page 30.

Table 1. Pin Descriptions

QFP Pin

TQFP Pin

Symbol

Type

Name/Function

1, 2, 3

RES

NC*

Reserved.

SMODE1

Serial Mode Select 1.

SMODE0

Serial Mode Select 0.

6, 7, 8,

9, 10

6, 7, 8,

9, 10, 11

RES

NC*

Reserved.

CKO1

Clock Output 1.

CLK

Clock Input.

XLO

Crystal Input.

XHI

Crystal Output.

XOSCEN

Crystal Oscillator Enable.

CKO2

Clock Output 2.

Digital Ground.

SADD

I/O

Serial Address.

Serial Input Data.

Serial Output Data.

Digital Power Supply.

RES

NC*

Reserved.

SYNC

I/O

Serial Input/Output Load Strobe and Synchronization.

IOCK

Serial

Clock.

24, 25,

26, 27, 28

26, 27, 28,

29, 30, 31

RES

NC*

Reserved.

RSTB

Reset.

PORB

Power-On Reset Output.

PORCAP

External Capacitor Connection for Power-On Reset.

SMODE2

Serial Mode Select 2.

EIGS

External Input Gain Select.

RES

NC*

Reserved.

SSA

Analog Ground.

REG

Regulated Output Voltage for Electrect Condenser Microphone.

DDA

Analog 5.0 V Power Supply.

AOUTN

Inverting Analog Output of Output Amplifier.

RES

NC*

Reserved.

AOUTP

Noninverting Analog Output of Output Amplifier.

SSA

Analog Ground.

MICIN

Analog Input for Microphone.

REFC

External Capacitor Connection for Internal Voltage Regulator.

AUXIN

Analog Input from Auxiliary.

DDA

Analog 5.0 V Power Supply.

Indicates no connection.

Indicates 3-state output.
Indicates pull-up device on input.
§ Indicates pull-up resistor on input.
** Indicates pull-down device on input.

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

4.1 Overview

The CSP1027 is a complete analog-to-digital and digi-
tal-to-analog acquisition and conversion system (see
Figure 3 on page 5) that band limits and encodes ana-
log input signals into 16-bit PCM, and takes 16-bit PCM
inputs and reconstructs and filters the resultant analog
output signal. The selectable A/D input circuits, pro-
grammable sample rates, and digital filter options allow
the user to optimize the codec configuration for either
speech coding or voice band data communications.
The on-chip digital filters meet the ITU-T G.712 voice
band frequency response and signal to distortion plus
noise specifications and are suitable for IS-54, GSM,
and JDC digital cellular applications. In addition, the
small supply current drain, when powered down,
extends battery life in mobile communication applica-
tions.

The CSP1027 is intended for both voice band voice and
data communication systems. As a result, this codec
has a variety of features not found in standard voice
band codecs:

3.0 V regulated power supply for a condenser micro-
phone.

Microphone preamplifier with programmable input
ranges.

Mute control of D/A output.

Programmable output gain in 3 dB increments.

Output speaker driver.

Programmable master clock divider to set A/D and
D/A conversion rate.

Testability loopback mode.

High-quality dither scheme to eliminate idle channel
tones.

4.2 Description of Signal Paths

4.2.1 Sampling Frequency

The oversampling ratio of the codec is 125:1; this is the
ratio of the frequency of the oversampling clock to the
frequency of the sampling clock. Most speech applica-
tions specify a sampling frequency of 8 kHz, yielding an
oversampling frequency of 8 kHz x 125 = 1.0 MHz. The
codec will operate at sampling frequencies up to
24 kHz, with the frequency response of the digital filters
being changed proportionally. For this architectural
description, the sampling frequency, f

, is assumed to

be 8 kHz, with an oversampling frequency, f

, of

1 MHz, unless otherwise stated.

4.2.2 Analog-to-Digital Path

The analog-to-digital (A/D) conversion signal path (see
Figure 3 on page 5) begins with the analog input driving
the input block. The signal from the input block is then
encoded by a second-order

modulator A/D. The

bulk of the antialiasing filtering is done in the digital
domain in two stages following the

modulator to

give a 16-bit result. The blocks will next be covered in
more detail.

4.2.3 Analog Input Block

The A/D input block operates in two modes: when the
external input gain select (EIGS) pin is low or left
unconnected, the input goes through a preamplifier and
is band limited by a second-order 30 kHz low-pass anti-
aliasing filter (see Figure 4 on page 7). When EIGS is
high, external resistors, Rin and Rfb, are used to set the
gain of an inverting amplifier (see Figure 5 on page 7).
These resistors, in combination with Cin and Cfb, cre-
ate a bandpass antialiasing filter. Note that EIGS is a
digital pin whose input levels are relative to digital
power and ground (V

and V

4.2.4 A/D Modulator and Digital Filters

A second-order

modulator quantizes the analog

signal to 1 bit (see Figure 3 on page 5). At the same
time, the resulting quantization noise is shaped such
that most of this noise lies outside of the baseband.
The modulator output is then digitally low-pass filtered
to remove the out-of-band quantization noise. After this
filtering, the output samples are decimated down to the
output sampling frequency. In the CSP1027, the filter-
ing and decimation are completed in two stages. The
first-stage low-pass filter shapes the modulator output
according to the sinc-cubic transfer function:

The output sampling frequency of the sinc-cubic filter is
reduced by a factor of 25 from 1 MHz to 40 kHz. The
sinc-cubic filter places nulls in the frequency response
at multiples of 40 kHz, and removes most of the quanti-
zation noise above 20 kHz so that very little energy is
aliased as a result of the decimation.

The sinc-cubic filter output is then processed by a
seventh-order IIR digital low-pass filter. This filter
removes the out-of-band quantization noise between
3.4 kHz and 20 kHz, compensates for the passband
droop caused by the sinc-cubic decimator, and deci-
mates the sampling frequency by a factor of five from
40 kHz to 8 kHz.

H z

( )

------

(

)

(

)

-------------------------

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

Following the low-pass filtering and decimation to 8 kHz, the 16-bit two's complement PCM can go directly to the
output register, cdx(A/D), or go to a third-order IIR digital high-pass filter and then to the output register. The 3 dB
corner frequency of the high-pass filter is approximately 270 Hz. This filter exceeds the VSELP preprocessing
requirements of IS-54 for attenuation of 60 Hz and 120 Hz signals. The high-pass filter is selected by writing the
HPFE field in the cioc3 register (see Table 10 on page 29). The default value upon reset is the high-pass filter
enabled (HPFE = 0).

Figure 4. CSP1027 A/D Path When in the Preamplifier Mode (EIGS = 0)

Figure 5. CSP1027 A/D Path in the External Gain Select Mode (EIGS = 1)

4.2.5 A/D Path Frequency Response

The composite digital filters (decimator, LPF, and HPF)
meet the ITU-T G.712 voice band frequency response
specifications and are suitable for IS-54, JDC, and
GSM digital cellular applications. Figures 6 through 9
show the A/D and D/A frequency response without the
optional high-pass filter (HPF). Figures 10 and 11 show
the group delay characteristics of the A/D and D/A with-
out the high-pass filter. Figures 12 and 13 show the fre-
quency response of the high-pass filter. Figures 14 and
15 show the group delay characteristics of the high-
pass filter. In all figures, the frequency is normalized to
the sampling frequency f

(i.e., frequency/f

). To get the

actual frequency, multiply the normalized frequency by
f

. The absolute delay and delay distortion have been

normalized to the sampling period 1/f

(i.e., delay x f

To obtain the actual delay, divide the normalized delay
by f

. The templates shown in Figures 7 through 9,

11 through 13, and 15 correspond to the limits in the
ITU-T G.712 specification where f

= 8.0 kHz.

4.2.6 PCM Saturation Versus Analog Input Levels

16-bit two's complement saturation is employed to pre-
vent wraparound during input overload conditions. The
saturation is hard-limiting:

0x7fff = maximum positive level

0x8000 = minimum negative level

The analog levels that correspond to the saturation lev-
els for the three input modes are outlined in Table 14 on
page 51.

MICIN

PREAMPLIFIER

AND AA-FILTER

A/D

AND

FILTERS

Rin

Cin

Vin1

AUXIN

Rin

Cin

Vin2

EXTERNAL

COMPONENTS

5-7592 (F)

MICIN

A/D

AND

FILTERS

Rfb

Cfb

AUXIN

EXTERNAL

COMPONENTS

INTERNAL

SIGNAL GND

INTERNAL

SIGNAL GND

Rin

Cin

5-7593 (F)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

4.2.7 Digital-to-Analog Path

Starting at the bottom right of Figure 3 on page 5, the

D/A conversion process begins with a 16-bit two's

complement PCM signal read from the DI serial input.
The PCM is interpolated up to 1 MHz in two stages and
low-pass filtered at each stage to attenuate 8 kHz
images.

The PCM input is latched into the cdx(D/A) register at
a nominal word rate of 8 kHz. The signal is then option-
ally high-pass filtered. This filter has the same transfer
function as the A/D high-pass filter.

A digital sample-and-hold increases the word rate by a
factor of 5 from 8 kHz to 40 kHz. The seventh-order IIR
digital low-pass filter then removes the spectral images
between 4 kHz and 20 kHz and predistorts the pass-
band to compensate for the filtering done during the
interpolation up to the 1 MHz word rate. The transfer
function of this low-pass filter is the same as the one
employed in the A/D converter.

The output of the low-pass filter feeds a programmable
gain adjustment block that serves as a volume control.
The gain can be changed in 3 dB increments from
0 dB to 45 dB. The attenuation level is set by writing
the OGSEL field in the cioc0 register (see Table 7 on
page 26).

The digital modulator block further increases the word
rate by a factor of 25 from 40 kHz to 1 MHz. Through
quantization and noise shaping, the digital

modula-

tor creates 1-bit output words at 1 MHz.

The modulator 1-bit output drives a structure combining
a 1-bit D/A converter and a second-order switched-
capacitor filter having a cutoff frequency of 8 kHz
(based on a 1 MHz clock). This is all shown as the D/A
block in Figure 3 on page 5.

This is followed by a second-order active Chebychev fil-
ter having a cutoff frequency of 35 kHz.

The passband ripple of the analog filters is small
enough such that they have virtually no effect on the
passband response.

The output amplifier buffers the analog filter output.

The frequency responses of the A/D and D/A paths are
essentially the same. See Figures 6 through 15 for the
magnitude and delay responses versus frequency.

4.3 Programmable Features

4.3.1 Active/Inactive Modes

The CSP1027 has active and inactive modes of opera-
tion which are selected by the ACTIVE field in the
cioc0 register (see Table 7 on page 26). The default
value upon reset and powerup is ACTIVE = 0 (i.e.,
inactive). In the inactive mode, the codec clocks are
disabled, data transfers by the codec are disabled, and
analog bias currents are shut off. This state is useful in
battery-powered applications when prolonged periods
of inactivity are expected. It takes approximately
600 ms for the codec to reach full steady-state perfor-
mance in going from inactive to active. This is primarily
due to the charging of the large external capacitors,
C

REF

and C

REG

. However, the codec is functionally

useful after 100 ms.

4.3.2 Input Select

When the A/D preamplifier is selected (EIGS = 0), the
INSEL field of cioc0 (see Table 7 on page 26) switches
the preamp input between the MICIN and AUXIN
inputs. When external gain select is used (EIGS = 1),
the INSEL field has no effect.

4.3.3 A/D Input Ranges

When the preamplifier is used (EIGS = 0), the IRSEL
field of the cioc0 register (see Table 7 on page 26)
selects the 500 mVp range when IRSEL = 0 and the
160 mVp range when IRSEL = 1. IRSEL has no effect
when the external gain select mode is used (EIGS = 1).

When EIGS = 1, the inverting amplifier of Figure 5 on
page 7 replaces the preamplifier. The input range in
this mode is the following:

FULL-SCALE

4.3.4 Output Mute Function

The D/A converter output can be selectively muted with
the MUTE field in the cioc0 register (see Table 7 on
page 26). The default value upon reset is muted
(MUTE = 0). The mute function is implemented (Figure
3 on page 5) internally by a MUX following the D/A
input. Placing the mute function here causes the signal
at the analog output to gradually decay/rise over
approximately 1 ms upon muting/unmuting. This effect
is due to the impulse response and group delay of the
digital filters. This implementation will reduce any
potentially undesirable transient effects such as pops,
when the D/A is muted.

Rin
Rfb

----------

1.578 Vp

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

4.3.5 Output Gains

The D/A converter output can be programmed in 3 dB
increments with the OGSEL field in the cioc0 register
(see Table 7 on page 26) to serve as a volume control.

4.3.6 Loopback Mode

The codec has a programmable loopback mode, repre-
sented by the TEST field in the cioc0 register, (see
Table 7 on page 26). As shown in Figure 3 on page 5,
when TEST = 0, the codec is in its normal mode of
operation. When TEST = 1, the loopback mode is acti-
vated. In loopback mode, the 1-bit PDM output signal
from the analog modulator is received by the analog
demodulator. At the same time, the 1-bit signal output
from the digital modulator is received by the sinc-cubic
filter in the A/D. This results in the analog input being
looped back to the analog output through the A/D and
D/A, and the digital input being looped back to the digi-
tal output through the digital filters. The loopback mode
can be useful for evaluating analog performance of the
codec in the target system without going through the
digital filters. This mode is also useful for evaluating the
response of the digital filters or in evaluating the read/
write functions of the codec and cdx registers without
having to provide an analog input to the A/D.

4.3.7 High-Pass Filter Select

The high-pass filter in the A/D and D/A can be enabled
or disabled with the HPFE field in the cioc3 register
(see Table 10 on page 29).

4.3.8 Dither

A dithering scheme is employed in the CSP1027 which
decorrelates the periodic quantization noise of the D/A
modulator to make it white noise.

converters are popular due to their high tolerance

to component mismatch present in integrated circuit
fabrication processes. However,

converters may

suffer from periodic noise and spurious tone generation
(in-band and out-of-band) due to the coarse quantiza-
tion and feedback of the

modulator. Although this

periodic noise may exist at very low levels (for example,
at about 90 dBm), it may be very objectionable to the
listener while having virtually no impact on the resolu-
tion of the converter. The CSP1027 D/A uses a robust
dithering scheme which eliminates any potential prob-
lems due to this phenomenon.

The DITHER field in the cioc3 register (see Table 10
on page 29) disables this feature. The default value
upon reset is DITHER = 0 (i.e., enabled). When the
DITHER is disabled, the signal-to-noise ratio will gener-
ally be about 2 dB higher. The DITHER should be
enabled if the CSP1027 is used in an audio application,
i.e., where this device interfaces to an audio trans-
ducer. If the CSP1027 is used in an application other
than audio, such as data communications, the DITHER
can be disabled if so desired.

4.4 Power-On Reset

4.4.1 Internal

The CSP1027 has a power-on reset circuit that is
ORed internally with the inversion of the reset pin,
RSTB, to form the internal reset (see Figure 16 on
page 15). The power-on reset circuit's inverted output
is also an output pin, PORB. The PORB can be used to
provide power-on reset to the system.

The power-on reset circuit is composed of two pulse-
generating elements, its output being the OR of the
two. One element is entirely internal and generates a
power-on pulse of 1.5 ms to 7.0 ms. The second ele-
ment is composed of an input pin, PORCAP, a resistor
connected between PORCAP and V

, and an invert-

ing input buffer. The user selects the capacitor value to
connect between PORCAP and ground that will gener-
ate a power-on pulse of desired width. The pin
PORCAP allows the user to lengthen the power-on
reset pulse to a width greater than the internal power-
on element provides. The nominal value of the resistor
is 155 k

, and the threshold of the inverting input buffer

is 0.6 x V

. The formula that relates the power-on

reset pulse delay to the PORCAP capacitor is as fol-
lows:

= R x C x log

(1 0.6)

= 0.9163 x R x C

Hence, to generate a 14.2 ms power-on reset pulse,
one would use a 0.1 µF capacitor connected between
PORCAP and V

An internal power-on pulse can be initiated after power-
on by writing a one to the TSTPOR field in the cioc3
register (see Table 10 on page 29). This causes the
internal power-on pulse of 1.5 ms to 7.0 ms to be gen-
erated. The pulse resets the device and appears on the
PORB output pin.

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

4.5 Clock Generation

Figure 17 on page 15 shows the clock generation and
distribution for the CSP1027. The programmable divid-
ers can customize the codec sample and master clock
rates for a variety of applications in addition to standard
8 kHz sampling, while allowing a range of values for
the crystal-controlled input clock. In Figure 17 on page
15, XOSCEN is a chip input to enable the crystal oscil-
lator circuit. XLO and XHI are the two leads for the
crystal. CLK is the chip clock input if the crystal is not
used. CK

is the internal codec sample clock, typically

8 kHz. CK

is the internal codec oversampled clock,

typically 1 MHz. CKO1 and CKO2 are general-purpose
clocks brought out to chip pins. CDIV1 and CDIV2 are
programmable dividers with a range from 1 to 31.
CDIV0 is programmed to be 1 or 2, but extra clock
pulses can be added or subtracted at the output for one
period of time following a write to the control register
cioc1. This one-time increase or decrease in the num-
ber of clocks is programmed by ADJMOD and ADJ and
causes a phase shift in the CKO1 and CK

output. F1

is an integral or a fractional divider controlled by the
five programmable coefficients shown connected to it.
With the fractional divide, the period of CK

will vary,

but the period of CK

will be constant.

The following discussion begins with the crystal oscilla-
tor and is followed by a detailed description of each
divisor block. Section 7.5 on page 45 provides some
examples of how to program the clocks.

4.5.1 Crystal Oscillator

The CSP1027 has a selectable on-chip clock oscillator.
A logic 1 on the XOSCEN pin enables the crystal oscil-
lator. A logic 0 disables the oscillator, powers it down,
and selects the input buffer connected to the CLK pin.

To use the oscillator, select a 20 MHz to 30 MHz funda-
mental-mode crystal with a series resistance less than
60

and a mutual capacitance less than 7.0 pF. Con-

nect the crystal between the XLO and XHI pins, and
add 10 pF capacitors between XLO and ground, and
XHI and ground. The XOSCEN pin enables and dis-
ables the crystal oscillator. See the application informa-
tion on optimizing the oscillator performance.

4.5.2 Clock Divider 2

Figure 18. Clock Divider 2

The CDIV2 field in cioc0 (see Table 7 on page 26) sets
the clock divider that generates the output clock,
CKO2. The clock output is a general-purpose clock that
can be used to clock external logic or processors.
CDIV2 ranges from 1 to 31, with 0 holding the output
low. RSTB going low sets CDIV2 to ÷16. CKO2 is
active while RSTB is low and synchronized by RSTB
going high.

4.5.3 Clock Divider 0

Figure 19. Clock Divider 0

The CDIV0 field in cioc1 (see Table 8 on page 27) sets
the clock divider that generates the internal clock 0
(ICLK0) to either divide by one or divide by two. The
ADJMOD and ADJ fields in cioc1 are used to adjust
the phase of ICLK0 by increasing or decreasing the
rate of ICLK0 for a burst of pulses, one time only. This
event occurs each time control register cioc1 is written
with nonzero values of ADJ. For example, let CDIV0 be
set to ÷2, ADJ to seven, and ADJMOD to one
(advance). After this word is written to the cioc1 regis-
ter, seven ICLK0 pulses will occur at the same rate as
ICLK, not divided by two. These seven clock pulses
shift the phase of CK

, CK

, and CKO1 earlier, thus

advancing these clocks. If ADJMOD is set to zero
(retard), the ÷2 becomes a ÷3 for seven pulses of
ICLK0. The CDIV0 clock divider is temporarily changed
internally so that it divides by one greater, to retard the
clocks, or one less, to advance the clocks, for the spec-
ified number of ICLK0 cycles. Note that the CDIV0
clock divider must be set to divide by two in order to
advance and retard the clocks. If CDIV0 clock divider is
set to divide by 1, one can only retard the clocks.

CDIV2

ICLK

CKO2

5-7589 (F)

CDIV0

ICLK

ICLK0

ADJMOD,

ADJ

5-7588 (F)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

CDIV0 has values of 1 or 2, ADJMOD is 0 or 1, and
ADJMOD ranges from 1 to 127, with 0 selecting no
clock adjust. RSTB going low sets CDIV0 to ÷2. ICLK0
is active while RSTB is low and synchronized by RSTB
going high.

4.5.4 Clock Divider 1

Figure 20. Clock Divider 1

The CDIV1 field in cioc1 (see Table 8 on page 27) sets
a clock divider that generates the CKO1 output clock.
This general-purpose clock output can be used for
clocking another codec in the system, such as the
CSP1084. The ability to phase adjust the output clock
and the codec sampling clock simultaneously is an
important feature. CDIV1 ranges from 1 to 31, with 0
disabling the output. RSTB going low sets CDIV1
to ÷16. CKO1 is active while RSTB is low and synchro-
nized by RSTB going high.

4.5.5 Sampling Clocks Generation

Figure 21. Sampling Clocks Generation

The oversampling codec clock CK

, typically 1 MHz,

is used in the front sections of the A/D and the back
sections of the D/A. The lower-frequency codec clock,
CK

, typically 8 kHz, is the sample clock at the output

of the A/D and the input to the D/A. The sampling clock
frequency, f

, is the oversampling clock frequency, f

divided by 125 (the fixed oversampling ratio). The
divide by 125 must remain fixed, since it is constrained
by the architecture of the codec digital filters. Many sys-
tems, however, have fixed high-frequency clocks and
fixed sampling clocks, so it is necessary to have a great
deal of flexibility in the creation of the codec clock CK

The CSP1027 solves this problem in a unique way, by
providing a programmable, fractional divider, F1.

F1 is the programmable ratio between ICLK0 and
CK

. The equation for F1 is:

where 3

64, 0

62, and S = {1, 1};

or M = 2, 0

62, and S = 1;

or M = 1, N = 0, and S = 1.

M is encoded by CDIV3 (see Table 3 on page 18), N is
encoded by CDIF0, CDIF1, and CDIF2 (see Table 5 on
page 19), and S is encoded by CDIFS (see Table 4 on
page 18).

is generated by dividing CK

by 125. The fre-

quency of CK

can be described by:

Note that when N = 0, ICLK0 is simply divided by the
integer M to create the oversampling clock, CK

. This

is the preferred method for generating the sampling
clock. If N

0, the fractional division results in an over-

sampling clock, CK

, whose period varies with time

such that the average period is the desired fraction.
This variation in the oversampling clock period is mini-
mized by the clock generator but can cause distortion
in the codec. Because the denominator of the fraction
is fixed at 125, the period of the sampling clock, CK

will be an integer multiple of the period of the internal
clock, ICLK0, and will not vary. This is more clearly
shown by the following equation:

The expanded equation below explains what is hap-
pening in the time domain:

During each sampling period,

, there are (125 N)

oversampling clock cycles of period

and N

oversampling clock cycles of period

. The N

oversampling clock cycles are evenly distributed
among the (125 N) oversampling clock cycles to min-
imize the distortion due to oversampling clock cycles of
differing period. The values for CDIF[0--2] in Table 5
on page 19 have been selected to achieve the even
distribution.

CDIV1

ICLK0

CKO1

5-7587 (F)

ICLK0

125

CDIFS, CDIF0, DSIF1,

CDIF2, CDIV3

5-7586 (F)

125

----------

125

ICLK

125

----------

125

ICLK

----------------

125

(

)

(

)

-----

125

(

)

ICLK

----------------

(

)

ICLK

--------------------

-----

ICLK

----------------

(

)

ICLK

--------------------

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

Table 5. CDIF0, CDIF1, CDIF2 Values for Each N

CDIF0

CDIF1

CDIF2

CDIF0

CDIF1

CDIF2

00 0000

0 0000

00 0100

10 1000

0 0010

11 1111

00 0000

0 0000

00 0100

10 0100

0 0010

11 1101

00 0010

0 0000

00 0011

00 0010

1 0010

10 1001

00 0010

0 0000

00 0011

00 0010

1 0100

01 1111

00 0000

0 0000

00 0011

00 0010

0 0100

01 1001

00 0000

0 0000

00 0011

1 0010

01 0101

10 0110

0 0000

00 0011

0 0010

01 0010

10 0100

0 0000

00 0011

00 0100

0 0010

00 1111

00 0010

0 0000

00 0011

00 0111

0 0000

00 1110

10 0101

0 0000

00 0011

00 1110

0 0000

00 1100

00 0010

0 0000

00 0011

11 0110

0 0000

00 1011

00 0010

0 0010

00 0011

10 1001

0 0000

00 1010

00 0010

0 0010

00 0011

10 0110

0 0000

00 1001

00 0010

1 0010

00 0011

10 0100

0 0010

00 1001

10 0111

0 0010

00 0011

10 0011

0 0010

00 1000

00 0011

0 0000

00 0010

1 0010

00 1000

10 0100

0 0010

00 0010

1 0011

00 0111

00 0011

0 0000

00 0010

1 0101

00 0111

10 1001

0 0010

00 0010

0 0000

00 0110

00 0010

1 0011

00 0010

0 0011

00 0110

00 0011

0 0010

00 0010

0 0010

00 0110

10 1011

0 0000

00 0010

00 0011

1 0011

00 0101

00 0010

1 0010

00 0010

00 0011

0 0011

00 0101

00 0010

0 0010

00 0010

00 0011

0 0010

00 0101

00 0100

0 0010

00 0010

00 0100

0 0010

00 0101

00 0000

0 0000

00 0010

00 0101

0 0000

00 0101

10 0100

0 0010

00 0010

00 0110

0 0000

00 0100

00 0010

1 0011

00 0010

00 0111

0 0010

00 0100

00 0010

0 0011

00 0010

00 1010

0 0010

00 0100

00 0011

0 0000

00 0010

00 1111

0 0010

00 0100

00 0101

0 0010

00 0010

00 0000

0 0000

00 0100

00 0000

0 0000

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

4.6 Serial I/O Configurations

4.6.1 Codec Data Transfer

When the codec is active, ACTIVE = 1 (see Table 7 on
page 26), it loads data into the cdx(A/D) and empties
data from the cdx(D/A) register (see Figure 3 on page
5) at the sampling frequency, f

(which is 8 kHz based

on a 1 MHz oversampling frequency). The codec data
transfers occur independent of the serial input/output
data transfers described below. The data is double
buffered, allowing the codec to transfer data to or from
the cdx while the serial I/O is shifting data into or out of
the shift registers (isr and osr). When the codec is set
to inactive, ACTIVE = 0, there are no codec data trans-
fers to the cdx(A/D) or from the cdx(D/A).

The internal STATUS flag is set high when cdx(A/D) is
loaded and cdx(D/A) is emptied. Loading data from the
cdx(A/D) into the output shift register (osr) or loading
data from the input shift register (isr) into the cdx(D/A)
due to a serial I/O transaction, clears the internal STA-
TUS flag. The internal STATUS flag can be observed
on the data output (DO) pin in the passive mode and
causes data transfers in the active and multiprocessor
modes.

4.6.2 Codec Control Writes

The four control registers are written through the serial
port. The serial address (SADD) selects between con-
trol and data transfers. Bits 15 and 14 of the control
word being transferred select which control register,
cioc0, cioc1, cioc2, or cioc3, is written (i.e., cioc0:
Bit[15:14] = 00, cioc1: Bit[15:14] = 01, etc.).

4.6.3 Serial I/O Port Overview

The CSP1027 serial I/O unit is an asynchronous, full-
duplex, double-buffered channel operating at up to
20 Mbits/s that easily interfaces with other Lucent fixed-
point DSPs (i.e., DSP16A and DSP1610/1616/1617/
1618) in a single or multiple DSP environment. Com-
mercially available codecs and time-division multi-
plexed (TDM) channels can be interfaced to the
CSP1027 device with little, if any, external logic.

The serial interface is a subset of the standard Lucent
DSP serial I/O and is comprised of eight pins:

A single passive serial input/output clock (IOCK).

A combined input load, output load, and synchroni-
zation (SYNC).

Serial data input (DI).

Serial data output (DO).

Serial address (SADD).

Three serial mode select pins (SMODE[2:0]).

The CSP1027's serial I/O is different from the standard
Lucent serial I/O in a number of ways:

The SMODE[1:0] pins configure the serial I/O port
into one of four possible ways: a passive SIO config-
uration, an active SIO configuration, and two multi-
processor SIO configurations.

A fixed most significant bit (MSB) first data format.

A fixed 16-bit data mode.

The serial address (SADD) is an input during the
passive and active SIO configurations to select
between data and control SIO transfers. It is
intended to be connected to the DSP's SADD pin,
which is an output during passive and active SIO.
Note that the DSP's SADD output is inverted and is
composed of two 8-bit fields that are shifted out least
significant bit (LSB) first.

The multiprocessor mode time slots and serial
addresses are restricted to two sets, one of which is
selected based on the state of SMODE0.

The SMODE2 pin should always be tied low for the
serial I/O port to operate as described.

The frequency of the serial I/O interface clock input
IOCK (F

IOCK

) must be greater than the frequency of

the internal oversampling clock (F

IOCK

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

Figure 22. Passive Communication and Connections

4.6.4 Passive I/O Configuration (SMODE[1:0] = 00)

The passive SIO configuration allows the user maxi-
mum flexibility in interfacing the CSP1027 to a variety
of system hardware configurations. It requires that the
user supply a serial input/output clock (IOCK) and per-
form data transfers at the sampling rate, f

. Serial data

transfers can be made to occur at the sampling rate by
applying a clock that is synchronous with the codec
clock, ICLK, to the SYNC pin or by polling the codec
STATUS flag, which indicates that the cdx(A/D) regis-
ter is full and the cdx(D/A) register is empty. The STA-
TUS flag appears on DO when the SADD pin selects a
control word.

Passive SIO is selected by setting both SMODE1 and
SMODE0 low. The input/output clock (IOCK) is an input
and the common input/output load, SYNC, (equivalent
to a DSP16A's ILD and OLD tied together) is also an
input. Serial data input (DI) is an input and serial data
output, DO, is an output. The serial address (SADD) is
an input, which determines if the transfer is to the con-
trol registers, cioc[0:3], or the data register, cdx(D/A).

A high-to-low transition of SYNC pin signal, latched by
the next rising edge of IOCK, initiates the start of an
input and output transaction. If the CSP1027's output

buffer, cdx(A/D), is full, it will be loaded into the output
shift register (osr) and shifted out on the DO pin. The
CSP1027 shifts in the data from the DI pin into its input
shift register (isr). A serial transmit address on the
SADD line is received simultaneously with data on the
DI line. If SADD is high for the first 15 bits, correspond-
ing to a zero serial transmit address, this causes the isr
to be latched into cdx(D/A) after 16 bits have been
shifted in. If SADD is low for any of the first 15 bits, cor-
responding to a nonzero transmit address, this causes
the isr to be latched into cioc[0:3] and also changes
the output data stream on DO to display the internal
codec STATUS flag. If SADD is low for any clock cycle,
while not involved in a serial transaction, the codec
STATUS flag is displayed on the DO pin until the next
data transfer.

An example of the passive SIO configuration is shown
in Figure 22. The DSP supplies both the serial clock
(IOCK) and the sampling synchronization signal to
SYNC, or polls the internal codec STATUS flag to
determine when a data transmission is needed. This
configuration allows the user maximum flexibility in
interfacing the CSP1027 to a variety of other system
hardware configurations.

A/D DATA

CLOCK

CONTROL/DATA ADDRESS

INPUT/OUTPUT LOAD

OLD

OCK

DSP

CSP1027

SMODE0
SMODE1
SMODE2

D/A DATA

5-7590 (F)

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

Figure 23. Active Communication and Connections

4.6.5 Active I/O Configuration (SMODE[1:0] = 01)

The active SIO configuration causes the CSP1027 to
generate an active input/output load (SYNC) to perform
input/output transmissions when needed. The user
supplies only a serial input/output clock (IOCK).

The active SIO is selected by setting SMODE1 low and
SMODE0 high. The input/output clock (IOCK) is an
input and the input/output load (SYNC) is an output.
While the codec is inactive, ACTIVE = 0 (see Table 7
on page 26), SYNC generates serial I/O transfers at an
IOCK ÷ 16 rate to allow loading the codec control regis-
ters, cioc[0:3]. While the codec is active, ACTIVE = 1
(see Table 7 on page 26), SYNC generates serial I/O
transfers at the sampling rate, synchronized to the
codec's emptying of the cdx(D/A) and loading of the
cdx(A/D). The serial address (SADD) functions as
described previously for the passive SIO configuration,
but with the SYNC pin being active and determining

data transfers, the need for polling the codec STATUS
flag is eliminated. The serial address during the data
stream still is used to determine whether data in the
input shift register is latched into cdx(D/A) or cioc[0:3]
at the end of the transaction. Note that cioc0, with
ACTIVE = 1, should be written last since this will
change the rate of serial I/O transfers from IOCK ÷ 16
to the sampling rate.

An example of the active SIO configuration is shown in
Figure 23. The DSP supplies the serial clock (IOCK)
while the CSP1027 supplies the input/output load,
SYNC. The serial address (SADD) is connected so that
writing the DSP's srta register addresses the cdx(D/A)
when srta = 0x0, or the cioc0, cioc1, cioc2, cioc3
when srta = 0x1. The DSP can activate the codec by
writing the cioc0 register in the CSP1027, and then let-
ting its input buffer full flag (IBF) indicate when the
CSP1027 has transferred data. This is the preferred
interface for a single DSP and a CSP1027.

A/D DATA

CLOCK

CONTROL/DATA ADDRESS

INPUT/OUTPUT LOAD

SAD

SYN

DSP

CSP1027

SMODE0
SMODE1
SMODE2

D/A DATA

SAD

5-7591 (F)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

4 Architectural Information

(continued)

Figure 24. Multiprocessor Communication and Connections

4.6.6 Multiprocessor Configuration (SMODE[1:0] = 1X)

The CSP1027 serial I/O supports a multiprocessor mode
that allows multiple devices to be connected together to
provide data transmission between any of the individual
devices. This mode requires no external hardware and
uses a time-division multiplex (TDM) interface with eight
time slots per frame.

Figure 24 shows an example of a multiprocessor system
with multiple DSPs and a CSP1027. The following pins
are connected together to form a four-wire bus:

DI and DO from the CSP1027s and DSPs form a data
line referred to as DATA.

IOCK from the CSP1027s and ICK and OCK from the
DSPs form a clock line referred to as CK.

SADD from the CSP1027s and DSPs form an address
line referred to as ADD.

SYNC from the CSP1027s and DSPs form a synchro-
nization line referred to as SYN.

Figure 25 on page 25 shows the time-slot allocation tim-
ing used in multiprocessor mode. One frame is defined
as the time between SYN high-to-low transitions. Each
frame is divided into eight time slots of 16 bits each. A

high-to-low transition of SYN defines the beginning of
time slot 0 and also resynchronizes any devices which
are operating on the multiprocessor bus with SYN as an
input. Note that the DSP device which drives the multi-
processor bus during time slot 0 also drives the SYN line.

Each CSP1027 sends data in a time slot determined by
its SMODE0 pin. Each DSP sends data in a time slot or
time slots determined by its tdms register. Only one
device can be assigned a particular time slot, and each
of the eight time slots must be assigned to a device (note
that one device can be assigned more than one time
slot). These requirements must be met by the user's
choice of CSP1027's SMODE0 connections and the
DSP's tdms register contents. A CSP1027 is assigned to
time slot 2 if SMODE0 is low or to time slot 5 if SMODE0
is high. This allows up to two CSP1027s to be placed on
a multiprocessor bus.

The DSP input/output format can be configured to either
most significant bit (MSB) first or least significant bit
(LSB) first. The CSP1027 only supports MSB first format;
hence, the DSPs connected on a multiprocessor bus
must be using MSB first data format, configured in the
sioc register, when transferring and receiving data and
control words with the CSP1027.

5-4181 (F).

DEV7

DATA

ADD

SYN

OCK

DEV1

OCK

DEV0

(DSP1616)

(DSP1610)

(CSP1027)

SMODE1
SMODE0
SMODE2

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

4 Architectural Information

(continued)

During each time slot (see Figure 26 on page 25), the
device that is assigned to that time slot drives the ADD
and DATA lines. If the assigned device's output buffer is
full, it loads its output shift register and shifts the 16 bits
of data, MSB first, out DO onto the DATA line. The 8-bit
transmit address is inverted and shifted out, LSB first,
onto the ADD line at the same time as the first 8 bits of
data. The inverted 8-bit protocol information is then
shifted out, LSB first, on the ADD line at the same time
as the last 8 bits of data. The CSP1027's transmit
address, AT[7:0], is determined by the SMODE0 pin
(see Table 6 on page 25). The DSP's transmit address
is determined by the srta register. The CSP1027's pro-
tocol information is always all zeros, which is inverted
to appear as all ones on the ADD line. The DSP's pro-
tocol information is determined by the saddx register. If
during a time slot the assigned device's output buffer is
empty, then zeros are shifted out on the DATA line and
zeros are shifted and inverted to become ones on the
ADD line.

During each time slot, each device receives the data on
the DATA line and inverts and receives the address and
protocol information on the ADD line. Each device
compares the transmitted 8-bit address with its receive
address. If the transmitted address and the device's
receive address have at least one occurrence of a one
in the same bit location, the address matches and the
device transfers the data from the input shift register to
its input buffer. If the transmitted address and the
receive address do not match, the data remains in the
input shift register and is overwritten during the next
time slot. The DSP's receive address is determined by
its srta register. Each CSP1027 has two receive
addresses, one for data and another for control, the
values of these two addresses are determined by the
SMODE0 pin (see Table 7 on page 26). When the data
receive address matches, the input shift register is

loaded into the cdx(D/A) register. When the control
receive address matches, the input shift register is
loaded into one of the four cioc registers, based upon
the two most significant bits of the 16-bit word. The
CSP1027 ignores the protocol information.

Multiprocessor communication with a CSP1027 is
intended to follow the sequence:

The DSP writes the control registers, cioc[0:3], in the
CSP1027 to configure the clock dividers and codec.
The codec is also activated.

The CSP1027's A/D fills the output buffer, cdx(A/D),
and empties the input buffer, cdx(D/A), at the same
time. This causes the A/D data to be transmitted by
the CSP1027 to the DSP during the next CSP1027
time slot. The DSP's receive address is set to match
the CSP1027's transmit address.

When the DSP receives A/D data from the CSP1027,
it responds by sending D/A data to the CSP1027 dur-
ing the DSP's next time slot. The DSP's transmit
address is set to match the CSP1027's data receive
address. The new data is loaded into the CSP1027's
cdx(D/A) register to be used as the next D/A sample.

If the DSP wants to send a control word to the
CSP1027 to change the configuration or inactivate
the codec, this can be done by setting the DSP's
transmit address to match the codec's control
receive address.

Note that since the CSP1027 sends all zeros for the
protocol information, this will have to be used to identify
the A/D data from the CSP1027. If two CSP1027s are
connected to the multiprocessor bus and the DSP's
receive address is set to match both CSP1027's trans-
mit addresses, the DSP will have to identify which A/D
data came from which CSP1027 by the order in which
the data arrives, since both CSP1027s will be sending
the same protocol information.

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

6 Signal Descriptions

Figure 27. CSP1027 Pinout by Interface

Figure 27 shows the pinout by interface for the
CSP1027. The signals can be separated into five inter-
faces as shown. These interfaces and the signals that
comprise them are described below.

6.1 Clock Interface

The clock interface consists of the clock input, crystal
oscillator, and clock outputs for the codec.

6.1.1 CLK

Clock Input: The input clock for the CSP1027 when
the XOSCEN is a logic low. Codec operation restricts
CLK to integer frequencies from 1 MHz to 40 MHz or
specific multiples of 8 kHz. When XOSCEN is tied to
logic high, the CLK input is not selected but should be
tied low or high to minimize input buffer power.

6.1.2 XLO

Crystal Input: The crystal for CSP1027 voice band
codec is connected between XLO and XHI. When the
crystal is not being used, this pin is to be left floating
and the CMOS clock applied to CLK.

6.1.3 XHI

Crystal Output: The crystal for CSP1027 handset
codec is connected between XLO and XHI. When the
crystal is not being used, this pin is to be left floating
and the CMOS clock applied to CLK.

6.1.4 XOSCEN

Crystal Oscillator Enable: When a logic high, the
crystal oscillator is selected for XLO and XHI pins.
When a logic low, the input buffer is selected for CLK
pin and the crystal oscillator is powered down.

Note: The XOSCEN pin does not have a pull-up or

pull-down device. Make sure that it is tied to V

tied to V

, or driven by valid logic levels.

CSP1027

CLK

XOSCEN

XLO

XHI

CKO1
CKO2

RSTB

PORCAP

PORB

IOCK

SYNC

SADD

SMODE0
SMODE1
SMODE2

EIGS

MICIN
AUXIN
V

REG

AOUTP
AOUTN

REFC

CLOCK

INTERFACE

RESET

INTERFACE

SERIAL

INTERFACE

EXTERNAL

GAIN SELECT

INTERFACE

ANALOG

INTERFACE

5-7606 (F)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

6 Signal Descriptions

(continued)

6.1.5 CKO1

Clock Out 1: ICLK ÷ CDIV1 (see Table 8 on page 27).
General-purpose output clock that can be used by a
baseband codec, such as the CSP1084.

6.1.6 CKO2

Clock Out 2: CLK ÷ CDIV2 (see Table 7 on page 26).
General-purpose output clock that can be used by a
processor, such as the DSP1616.

6.2 Reset Interface

The reset interface consists of the reset input, power-
on reset input, and power-on reset output for the codec.

6.2.1 RSTB

Reset: A high-to-low transition causes entry into the
reset state. The cioc[0:3] register bits are set to their
default states.

6.2.2 PORB

Power-On Reset: A high-to-low transition indicates
entry into the power-on reset state.

6.2.3 PORCAP

Power-On Reset Capacitor: A capacitor is to be
attached to this pin for the power-on reset circuit. POR-
CAP has an internal resistor (nominal value of 155 k

)

connected to digital power, V

6.3 Serial I/O Interface

The serial I/O interface consists of the serial clock
input, synchronizing signal, data input, data output,
serial address, and serial modes for the codec.

6.3.1 SMODE 0

Serial Mode 0: Configures the CSP1027 serial I/O
interface. When in active/passive mode (SMODE1 low),
SYNC is an output when SMODE0 is high, and SYNC
is an input when SMODE0 is low. In multiprocessor
mode (SMODE1 high), SMODE0 selects between two
possible time slots, and between two possible transmit
and receive address combinations. See Table 6 on
page 25 in the architectural information for the
addresses.

6.3.2 SMODE1

Serial Mode 1: Configures the CSP1027 serial I/O
interface to multiprocessor mode when active-high; oth-
erwise, active/passive mode is selected when low.

6.3.3 SMODE2

Serial Mode 2: Must be tied low to configure the
CSP1027 serial I/O interface as described.

6.3.4 DI

Serial Data Input: Serial data input is latched on rising
edge of IOCK, MSB first. DI and DO should be con-
nected together when in multiprocessor mode.

6.3.5 DO

Serial Data Output: Serial data output from the output
shift register (osr), MSB first, when the data register,
cdx(A/D), is selected or codec status flag when the
control registers, cioc[0:3], are selected. When an out-
put, DO changes on the rising edges of IOCK. DI and
DO should be connected together when in multiproces-
sor mode, SMODE1 high.

6.3.6 IOCK

Serial Input/Output Clock: Input clock for serial PCM
input and output data.

Note: The frequency of the serial I/O interface clock

input IOCK (F

IOCK

) must be greater than the fre-

quency of the internal oversampling clock CK

CKOS

6.3.7 SYNC

Serial Input/Output Load Strobe and Sync: When
not in multiprocessor mode, the falling edge of SYNC
indicates the beginning of a serial input and a serial
output word. The falling edge of SYNC loads the output
shift register (osr) from the codec data register
(cdx(A/D)). Sixteen IOCK clock cycles after the falling
edge of SYNC, the codec data (cdx(D/A)) or control
register (cioc) is loaded from the input shift register
(isr). SYNC is an input when the SMODE0 pin is low
and an output when the SMODE0 pin is high.

In multiprocessor mode, SYNC is the multiprocessor
synchronization input signal. A falling edge of SYNC
indicates the first word of a TDM I/O stream and
causes the resynchronization of the internal input and
output load generators.

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

6 Signal Descriptions

(continued)

6.3.8 SADD

Serial Address: When not in multiprocessor mode,
SADD is an input that selects between the codec data
registers, cdx(D/A) and cdx(A/D), and codec control
registers, cioc[0:3]. SADD is inverted and latched on the
rising edge of IOCK and compared against a zero for
data and a one for control, to determine if input data on
DI is loaded from the input shift register (isr) into
cdx(D/A) or one of cioc[0:3]. Once SADD indicates a
control word, the internal codec status flag appears on
DO, replacing cdx(A/D). While not performing a serial
transmission, SADD low causes the internal codec sta-
tus flag to be output on DO.

In multiprocessor mode, SADD is an output when the
tdms time slot dictates a serial output transmission; oth-
erwise, it is an input. While an output, SADD is the
inverted 8-bit serial transmit address output, LSB first.
SADD changes on the rising edges of IOCK. While an
input, SADD is inverted and latched on the rising edge
of IOCK and compared against the cdx(D/A) and
cioc[0:3] serial receive addresses to determine if input
data on DI is loaded from the input shift register (isr) into
cdx(D/A) or cioc.

6.4 External Gain Control Interface

The external gain control interface consists of one input.

6.4.1 EIGS

External Input Gain Select: A logic low or no connect
selects the microphone preamplifier. A logic high selects
the single op amp input mode where external resistors
set the A/D input range. Note that EIGS is a digital pin
whose input levels are relative to digital power and
ground (V

and V

6.5 Digital Power and Ground

Digital Power Supply: 3.0 V to 5.0 V supply.

Digital Ground: 0 V.

6.6 Analog Interface

The analog interface consists of the two inputs, two out-
puts, a regulated output voltage reference, and a capac-
itor connection for the codec.

6.6.1 MICIN

Analog Input from Microphone: Low-level analog sig-
nal from electret condenser microphone selected by
INSEL bit in codec control register, cioc0 (see Table 7
on page 26).

6.6.2 AUXIN

Analog Input from Auxiliary: When used in preampli-
fier mode (EIGS = 0), AUXIN is a low-level analog signal
selected by INSEL bit in codec control register, cioc0
(see Table 7 on page 26). The characteristics of AUXIN
are identical to MICIN.

When used in external gain select mode (EIGS = 1),
AUXIN is the output of the inverting amplifier. The INSEL
bit has no effect in this mode.

6.6.3 AOUTP

Noninverting Analog Output: In conjunction with
AOUTN, this output can drive a 2 k

load in differential

mode or a 1 k

load ac-coupled to analog ground.

6.6.4 AOUTN

Inverting Analog Output: In conjunction with AOUTP,
this output can drive a 2 k

load in differential mode or a

1 k

load ac-coupled to ground.

6.6.5 V

REG

Regulated Output Voltage: For electret condenser
microphone. Vout = 3 V ± 10%, Iout = 250 µA max. A
1 µF and 0.1 µF ceramic type X7R capacitor to ground
must be provided at this pin (see Figure 28 on page 34).

6.6.6 REFC

External Capacitor Connection: Internal voltage regu-
lator bypassing. A 0.22 µF ceramic type X7R capacitor
to ground must be provided at this pin.

6.7 Analog Power and Ground

DDA

Analog Power Supply: 5.0 V supply.

SSA

Analog Ground: 0 V.

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

7 Application Information

This section begins with application information for the
analog section, followed by power distribution, crystal
oscillator, and codec clock generation programming
examples.

7.1 Analog Information

The A/D input block is covered first, followed by the
D/A, and the microphone voltage regulator.

7.1.1 A/D in the Preamplifier Mode

Figure 28 on page 34 shows a typical telephone hand-
set application. The codec is shown with the preamp
mode (EIGS = V

) selected and connected to a micro-

phone. The analog-to-digital conversion path begins
with an on-chip preamplifier front end having two
single-ended inputs. The preamp inputs are MICIN and
AUXIN. Selection of MICIN or AUXIN is made via the
INSEL field in the cioc0 register (see Table 8 on page
27) and can be dynamically changed, as desired. The
electrical specifications for both inputs are the same.

An off-chip ac-coupling capacitor, Cin, is required
before each input. However, if either input is unused, it
may be left unconnected (floating). The input resis-
tance (Rin) to either MICIN or AUXIN is approximately
40 k

. The recommended value of Cin is 0.15 µF. This

creates a high-pass filter pole at approximately 26 Hz.
A larger capacitor value may be used if desired, in
order to allow lower frequencies to pass to the A/D con-
verter, but smaller capacitor values are not recom-
mended.

7.1.2 A/D in the External Input Gain Select Mode

The external input gain select (EIGS = V

) is used

when the input range is set by the user (see Section
4.3 on page 13). The A/D input circuitry of Figure 28 on
page 34 is modified as shown in Figure 5 on page 7.
When EIGS = V

, the following notes apply.

1. The recommended range of values for the feedback

resistor and capacitor are the following:
10 k

Rfb

45 k

and 150 pF

Cfb

680 pF.

2. The

external resistor ratio accuracy directly

impacts the absolute accuracy of the A/D path. A 1%
ratio error adds 86 mdB of absolute gain error.

3. The A/D input sampling switches have an effective

bandwidth on the order of 15 MHz. The amplifier
unity gain frequency is on the order of 3 MHz. High-
frequency noise in the 1 MHz to 50 MHz range that
couples to the AUXIN pin will be somewhat attenu-
ated by the amplifier output impedance, but a signifi-
cant portion will be sampled by the A/D and aliased
down to the baseband. Special care in circuit board
layout is required to keep noise sources from cou-
pling into the AUXIN or MICIN pins so that the noise
and distortion performance shown in Table 16 on
page 52 can be achieved.

The codec is not as sensitive to wideband noise
when the preamplifier is used (EIGS = V

) because

the A/D inputs are driven from an on-chip low-pass
filter.

4. The external gain mode input circuitry of Figure 5 on

page 7 is an integrator with a great deal of loss. The
frequency response of Table 16 on page 52
assumes that the Rfb

Cfb corner frequency is

25 kHz so the 3 kHz droop is less than 65 dBm. Sim-
ilarly, the Cin

Rin corner frequency is set to 7 Hz.

These RC combinations create a bandpass anti-
aliasing filter with corner frequencies given by:

When selecting component values, verify that the
A/D frequency response will still meet the application
requirements.

7.1.3 D/A Analog Output

The CSP1027 D/A has two analog outputs, AOUTP
and AOUTN, capable of operating as two single-ended
drivers, or a single fully differential driver. The output
impedance of each is no more than 6

(12

if config-

ured as fully differential) over the dc to 4 kHz frequency
range.

The maximum open-circuit output levels are 2.1 Vp
(4.2 Vp-p) if fully differential, and one-half of these lev-
els if single-ended. These levels correspond to a full-
scale 16-bit two's complement PCM input into the D/A
converter, with the output gain setting (OGSEL) at
0 dB. For any given PCM input, the output levels will be
reduced by a voltage division of the D/A output and the
load impedance:

The driver linearity is only guaranteed for the single-
ended output load resistance (R

) of at least 1000

and the differential output load resistance (R

) of at

least 2000

Rfb
Rin

----------

Rin

Cin

---------------------------------------

Rfb

Cfb

---------------------------------------

OU T

----------------------

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

7 Application Information

(continued)

Figure 29 (A--E) on page 35 illustrates five different analog output configurations. In Figure 29A, the output load is
driven in a fully differential manner. It is assumed that the load is floating, having no path to ground. Figure 29B is
similar, but ac-coupling capacitors are used. A low-pass pole is created, whose frequency should be less than
30 Hz in order to not interfere with the voice band frequency response.

In Figure 29C--E, different variations of single-ended loads are shown. In any single-ended configuration, ac-
coupling capacitors are required.

Notes:
Analog (V

SSA

) and digital (V

) ground pins are tied together to prevent substrate currents from ground bounce.

Capacitors C

, C

REF

, C

REG

1, and C

REG

2 should be type X7R ceramic.

Capacitors C

and C

should be tantalum or low ESR aluminum.

All capacitors should be located as close to the chip pins as possible.

Keep analog and digital grounds separate, and then join at the V

SSA

pin.

Keep the regulator as close to the chip as possible.

The power connections are shown when the analog and digital are run from 5.0 V. When the digital is run from a 3.3 V power supply, C

and

go from the 3.3 V regulator to ground, and Ra goes to the 5.0 V regulator.

Figure 28. Analog External Configurations in Preamplifier Mode (EIGS = 0)

5-7607 (F)

AOUTP

AOUTN

CSP1027

TO 5.0 V
REGULATOR

DIGITAL

GND

PLANE

DDA

SSA

REFC

)

(0.1

(10

SSA

REG

(0.1

ANALOG

GND PLANE

EIGS

AUXIN

MICIN

REG

SSA

Cin

(0.15

REF

(0.22

Vin2

MICROPHONE

SHIELDING

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

7 Application Information

(continued)

7.1.4 Microphone Regulator

REG

is a 3.0 V regulated supply that provides up to

250 µA to an external microphone or other device (see
Figure 28 on page 34). The regulator uses the external
capacitors C

REG

1 and C

REG

2 to band limit its noise

and for frequency compensation. The C

REG

1 off-chip

capacitor (1 µF) is required in order to meet the noise
specification of 100 µV on V

REG

. C

REG

2 (0.1 µF)

should be placed in parallel with C

REG

1 to improve

high-frequency noise filtering. The minimum value of
the C

REG

1 and C

REG

2 combination is 0.1 µF for V

REG

to be stable. If V

REG

is not used, this pin should be

either connected through a 0.1 µF capacitor to analog
ground (V

SSA

) or tied directly to ground. Connecting

REG

to ground will produce a dc current of 250 µA to

400 µA out the V

REG

pin, but will not change the total

supply current. Do not leave the V

REG

pin unconnected

because it will oscillate.

7.2 Power Supply Configuration

Figure 28 on page 34 illustrates the recommended
configuration for the analog and digital power and
grounds.

An external supply feeds an off-chip voltage regulator.
Capacitor C

(10 µF) and C

(0.1 µF) are used for

decoupling the noise on the V

digital power bus.

Capacitors C

(10 µF) and C

(0.1 µF) are for de-

coupling the noise on the analog power bus (V

DDA

The Ra resistor (3

) decouples the analog and digital

power buses when a common 5.0 V power supply is
used. The analog and digital circuits share the same
substrate since this codec is a monolithic device. In the
technology used to fabricate the device, the substrate
is connected to ground. To avoid large substrate cur-
rents caused by digital ground-bounce, it is recom-
mended that the analog and digital grounds be tied
together at the package, as shown in Figure 28 on
page 34. It is recommended that the analog and digital
ground planes also meet at this point. In a typical appli-
cation where the CSP1027 is interfaced to a DSP, it is
advisable to place the DSP as close to the codec as
possible, with the DSP's digital ground plane extending
to the points where the SIO lines meet the CSP1027.

7.2.1 REFC Capacitor

An off-chip capacitor, C

REF

(0.22 µF), is required on pin

REFC in order to meet the noise requirements for the
internal signal paths.

7.2.2 Capacitor Proximity to Pins

In all cases, the external capacitors should be placed
as closely as possible to the CSP1027 pins, in order to
meet the noise specifications.

7.3 The Need for Fully Synchronous

Operation

7.3.1 Introduction to Sampled Data Systems

The analog circuits in the A/D and D/A converters are
sampled data circuits. This means that there are
switches that close to sample the signal and then open
to hold the signal. An example of this kind of discrete
time analog circuit is the well-known switched capacitor
technique used to implement A/D and D/A converter
circuits as well as filters.

A fundamental property of any sampled data system is
that any noise or signal that is in the signal path when
the sampling switches open is sampled. The sampling
process modulates the noise and signal about multi-
ples of the sample clock rate. For a sample rate of f

and a noise tone at a frequency of f

, this modulation

process produces new tones at

(k x f

) ± f

where k = 1, 2, 3 . . . . For noise near a multiple of f

the difference term can modulate all the way down to
baseband and be heard as a tone.

A typical source of noise is that generated by the nor-
mal operation of the digital circuits. The digital circuits
tend to have fast edge transitions (large dv/dt and
di/dt). The dv/dt changes couple into the analog signal
path through parasitic capacitance on-chip and in the
circuit board. The di/dt changes cause voltages to be
generated across parasitic inductance and cause
ground-bounce on-chip. The ground-bounce can turn
on intrinsic parasitic diodes to the substrate of the
CSP1027, and the subsequent substrate currents can
couple the noise into the analog circuits. The di/dt tran-
sients are also inductively coupled into the off-chip
analog routing and thus added to the analog signals.
Layout techniques help reduce the dv/dt and di/dt cou-
pling, but it is very difficult to eliminate it.

To gauge the magnitude of the problem, consider the
numbers from the CSP1027. The digital logic swing is
ground to V

, which can be as large as 5.5 V. The

full-scale preamplifier input level (when IRSEL = 1) is
160 mVp, and the A/D path has a noise floor that is
guaranteed to be 70 dB below full scale. For the digital
noise to raise the noise floor by less than 3 dB, the

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

7 Application Information

(continued)

noise must be less than 36 µVrms referred to the
preamplifier input. As a worst-case analysis, assume
that all the digital noise is in the baseband (noise
source of 2.5 Vrms for a 5.0 V digital signal). The atten-
uation (isolation) between the digital signal and the
preamplifier for this case must be

isolation

= 70,000:1 (97 dB)

Usually, only a small portion of a particular digital signal
ac-couples into the signal path, but this is tempered by
having many digital signals. It is the sum of these noise
sources that must be held to less than 36 µVrms. In
audio applications, the needed isolation is actually
greater than calculated above because the human ear
can detect tones that are 10 dB below the noise floor.

7.3.2 Typical Ways Digital Noise Couples into

Analog Circuits

1. Digital signals have overshoot or undershoot

because of circuit board impedance mismatches.
These reflections can turn on internal I/O protection
diodes. The diodes then inject the noise current into
the substrate as well as the chip power rails, and the
noise is distributed throughout the chip.

2. Fine-line CMOS (like that used to fabricate the

CSP1027) generates hot electron currents when the
logic gates change state. This current is injected into
the substrate and adds to the supply current. The
substrate current can couple directly into the analog
circuits, and the supply current transients can couple
through common supply impedance and by inductive
coupling.

3. Coupling off-chip, such as the package bond wires

and package pins, and coupling into the analog sig-
nals on the circuit board.

4. The classic coupling method: common power and

ground impedance.

7.3.3 The Problem with an Asynchronous Codec

Clock

When the I/O and sample clocks are not derived from
the same time base, their edges will drift with time.
Even if the I/O and sample time bases use master
oscillators that are stated to be the same frequency (or
one being a multiple of the other), they will differ by
some amount from their intended values. This differ-
ence in frequency will cause the digital circuit clock
edges to slide past the analog sample clock edges in a
periodic way, causing the sampled digital noise to also
vary in the same periodic way (noise tones in the base-
band).

7.3.4 The Advantage of Fully Synchronous

Operation

When all the clocks that are used in or connected to the
codec are generated from the same master time base,
the sampling switches sample in the quiet time before
the digital circuits change state, or at least sample the
same portion of the ground bounce transient. The por-
tion of the noise that does not change from one sample
to the next will alias to the signal path as a dc offset.
The portion that is signal dependent (like a data line
coupling into the analog signal path) can show up as a
tone even though its transitions occur synchronously
with the sample clock unless care is taken to ensure
that the analog sampling switches only open during a
quiet time (usually before the digital circuits change
state).

2.5 V

----------------

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

7 Application Information

(continued)

7.4.3 Printed-Circuit Board Layout Considerations

The following guidelines should be followed when designing the printed-circuit board layout for a crystal-based
application:

1. Keep crystal and external capacitors as close to XLO and XHI pins as possible to minimize board stray capaci-

tance.

2. Keep high-frequency digital signals such as CKO1 and CKO2 away from XLO and XHI traces to avoid coupling

into the oscillator.

7.4.4 LC Network Design for Third Overtone Crystal Circuits

For operating frequencies of greater than 30 MHz, it is usually cost advantageous to use a third overtone crystal as
opposed to a fundamental mode crystal. When using third overtone crystals, it is necessary, however, to filter out
the fundamental frequency so that the circuit will oscillate only at the third overtone. There are several techniques
that will accomplish this; one of these is described below.

Figure 35 shows the basic setup for third overtone operation.

Figure 35. Third Overtone Crystal Configuration

The parallel combination of L

and C

forms a resonant circuit with a resonant frequency between the first and third

harmonic of the crystal such that the LC network appears inductive at the fundamental frequency and capacitive at
the third harmonic. This ensures that a 360

phase shift around the oscillator loop will occur at the third overtone

frequency but not at the fundamental. The blocking capacitor, C

, provides dc isolation for the trap circuit and

should be chosen to be large compared to C

For example, suppose it is desired to operate with a 40 MHz, third overtone, crystal.

Let:

= operating frequency of third overtone crystal (40 MHz in this example)

= fundamental frequency of third overtone crystal, or f

/3 (13.3 MHz in this example)

= resonant frequency of trap =

= external load capacitor (10 pF in this example)

= dc blocking capacitor (0.1 µF in this example)

XLO

XHI

XTAL

5-7614 (F)

L1C1

--------------------------

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

7 Application Information

(continued)

7.4.5 Frequency Accuracy Considerations

For most applications, clock frequency errors in the hundreds of parts per million (ppm) can be tolerated with no
adverse effect. However, for applications where precise frequency tolerance on the order 100 ppm is required, care
must be taken in the choice of external components (crystal and capacitors) as well as in the layout of the printed-
circuit board. Several factors determine the frequency accuracy of a crystal-based oscillator circuit. Some of these
factors are determined by the properties of the crystal itself. Generally, a low-cost, standard crystal will not be suffi-
cient for a high-accuracy application, and a custom crystal must be specified. Most crystal manufacturers provide
extensive information concerning the accuracy of their crystals, and an applications engineer from the crystal ven-
dor should be consulted prior to specifying a crystal for a given application.

In addition to absolute, temperature, and aging tolerances of a crystal, the operating frequency of a crystal is also
determined by the total load capacitance seen by the crystal. When ordering a crystal from a vendor, it is neces-
sary to specify a load capacitance at which the operating frequency of the crystal will be measured. Variations in
this load capacitance due to temperature and manufacturing variations will cause variations in the operating fre-
quency of the oscillator. Figure 36 illustrates some of the sources of this variation.

Notes:
C

EXT

External load capacitor (one each required for XLO and XHI).

Parasitic capacitance of the CSP1027 itself.

Parasitic capacitance of the printed-wiring board.

Parasitic capacitance of crystal (not part of C

, but still a source of frequency variation).

Figure 36. Components of Load Capacitance for Crystal Oscillator

The load capacitance, C

, must be specified to the crystal vendor. The crystal manufacturer will cut the crystal so

that the frequency of oscillation will be correct when the crystal sees this load capacitance. Note that C

refers to a

capacitance seen across the crystal leads, meaning that for the circuit shown in Figure 36, C

is the series combi-

nation of the two external capacitors (C

EXT

/2) plus the equivalent board and device strays (C

/2 + C

/2). For exam-

ple, if 10 pF external capacitors were used and parasitic capacitance is neglected, then the crystal should be
specified for a load capacitance of 5 pF. If the load capacitance deviates from this value due to the tolerance on the
external capacitors or the presence of strays, then the frequency will also deviate.

XLO

XHI

XTAL

EXT

5-7615 (F)

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

7 Application Information

(continued)

This change in frequency as function of load capacitance is known as pullability and is expressed in units of ppm/
pF. For small deviations of a few pF, pullability can be determined by the equation below.

pullability (ppm/pF) =

where C

= parasitic capacitance of crystal.

= motional capacitance of crystal (usually around 1 fF--25 fF, value can be obtained from

crystal vendor).

= total load capacitance seen by crystal.

Note that for a given crystal, the pullability can be reduced, and hence, the frequency stability improved, by making
C

as large as possible while still maintaining sufficient negative resistance to ensure start-up per the curves

shown in Figures 31 and 32 on page 39.

Since it is not possible to know the exact values of the parasitic capacitance in a crystal-based oscillator system,
the external capacitors are usually selected empirically to null out the frequency offset on a typical prototype board.
Thus, if a crystal is specified to operate with a load capacitance of 10 pF, the external capacitors would have to be
made slightly less than 20 pF each in order to account for strays. Suppose, for instance, that a crystal for which
C

= 10 pF is specified is plugged into the system and it is determined empirically that the best frequency accuracy

occurs with C

EXT

= 18 pF. This would mean that the equivalent board and device strays from each lead to ground

would be 2 pF.

As an example, suppose it is desired to design a 26 MHz, 3.3 V system with ±100 ppm frequency accuracy. The
parameters for a typical high-accuracy, custom, 26 MHz fundamental mode crystal are as follows:

Initial Tolerance

10 ppm

Temperature Tolerance

25 ppm

Aging Tolerance

6 ppm

Series Resistance

max

Motional Capacitance (C

)

15 pF max

Parasitic Capacitance (C

)

7 pF max

In order to ensure oscillator start-up, the negative resistance of the oscillator with load and parasitic capacitance
must be at least twice the series resistance of the crystal, or 40

. Interpolating from Figure 32 on page 39, exter-

nal capacitors plus strays can be made as large as 30 pF while still achieving 40

of negative resistance. Assume

for this example that external capacitors are chosen so that the total load capacitance including strays is 30 pF per
lead, or 15 pF total. Thus, a load capacitance, C

= 15 pF would be specified to the crystal manufacturer.

From the above equation, the pullability would be calculated as follows:

pullability =

= 15.5 ppm/pF

If 2% external capacitors are used, the frequency deviation due to this variation is equal to

(0.02)(15 pF)(15.5 ppm/pF) = 4.7 ppm.

Note: To simplify analysis, C

EXT

is considered to be 30 pF. In practice, it would be slightly less than this value to

account for strays. Also, temperature and aging tolerance on the capacitors have been neglected.

Typical capacitance variation of oscillator circuit in the CSP1027 itself across process, temperature, and supply
voltage is ±1 pF. Thus, the expected frequency variation due to the CSP1027 is as follows:

(1 pF)(15.5 ppm/pF) = 15.5 ppm.

Approximate variation in parasitic capacitance of crystal = ±0.5 pF.

Frequency shift due to variation in C

= (0.5 pF)(15.5 ppm/pF) = 7.75 ppm.

Approximate variation in parasitic capacitance of printed-circuit board = ±1.5 pF.

Frequency shift due to variation in board capacitance = (1.5 pF)(15.5 ppm/pF) = 23.25 ppm.

(

)

(

)

2 C

(

)

--------------------------------

(

)

(

)

2 C

(

)

-------------------------------

0.015

(

)

(

)

2 7

(

)

----------------------------------

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

7 Application Information

(continued)

Thus, the contributions to frequency variation add up
as follows:

Initial Tolerance of Crystal

10.0 ppm

Temperature Tolerance of Crystal

25.0 ppm

Aging Tolerance of Crystal

6.0 ppm

Load Capacitor Variation

4.7 ppm

CSP1027 Circuit Variation

15.5 ppm

Variation

7.8 ppm

Board Variation

23.3 ppm

Total

92.3 ppm

This type of detailed analysis should be performed for
any crystal-based application where frequency accu-
racy is critical.

7.5 Programmable Clock Generation

Refer to Figure 17 on page 15 for the following discus-
sion.

The programmable clock divider is set by writing the
6-bit CDIV3 field of the cioc2 register (see Table 9 on
page 28). The user can select an appropriate integer
value which sets the ratio of the CLK input clock to the
oversampling rate of the codec. The following examples
illustrate this feature.

7.5.1 Application Example 1

GSM application.

Input clock, CLK, rate: 26 MHz (38.46 ns).

Codec PCM rate required: 8 kHz (oversampling
rate = 1 MHz).

Solution:

CLK/CK

= 26, so set CLK/ICLK0 = 1 and ICLK0/

to 26.

Set CDIV0 = 0 and CDIV3 = 26 (011010).

7.5.2 Application Example 2

IS-54 application.

Input clock, CLK, rate: 40 MHz (25.0 ns).

Codec PCM rate required: 8 kHz (oversampling
rate = 1 MHz).

Solution:

CLK/CK

= 40, so set CLK/ICLK0 = 1 and ICLK0/

to 40.

Set CDIV0 = 0 and CDIV3 = 40 (101000).

7.5.3 Application Example 3

Modem data pump.

Codec sampling frequency required = 9.6 kHz
(instead of 8 kHz).

Need highest possible input clock, CLK, rate (allow-
able by the DSP).

Solution:

Codec oversampling rate = 9.6 kHz * 125 = 1.2 MHz.

Assuming a DSP16A or DSP1616 with
maximum rate of 40 MHz,
CLK = 39.6 MHz = 1.2 MHz x 33, so
CLK/ICLK0 = 1 and ICLK0/CK

= 33.

Set CDIV0 = 0 and CDIV3 = 33 (100001).

Disable the high-pass filters (HPFE = 1) because the
3 dB corner frequency is now too high
(270 Hz x 1.2 = 324 Hz).

Low-pass filter 3 dB corner frequency is now
4.08 kHz (= 3.4 kHz x 1.2). (Note that external DSP
software can provide additional postfiltering, if
desired.)

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

7 Application Information

(continued)

7.5.4 Enhanced Oversampling Clock Generation

If system constraints make the requirement of integer
multiples of 125 x the sampling rate (typically integer
multiples of 1.0 MHz) difficult to provide, the CSP1027
can also operate with the ICLK0 internal clock rate at
integer multiples of the sampling rate (typically integer
multiples of 8 kHz). See Section 4.5 on page 16 for
more information.

The following two examples illustrate the usage:

Application Example 4

Standard codec application.

CLK input clock rate: 2.048 MHz.

Codec sampling rate, f

: 8 kHz.

Solution:

= 125 x 8 kHz = 1.0 MHz, so CLK/CK

2.048 or CLK/CK

= 2.048 x 125 = 256. Values of M

and N must be found to satisfy this requirement, as
shown below.

Set CDIV0 for ÷1 (CDIV0 = 0); hence, f

ICLK

= f

CLK.

Using the equations from Section 4.5 on page 16,

Hence, M = 2, S = 1, and N = 6. (Note that if CDIV0
set for ÷ 2, then M = 1, S = +1, and N = 3, which is
not allowed.)

Using Tables 2 through 4 on page 18:
CDIV3 = 00 0010.
CDIFS = 0.
CDIF0 = 01 0101.
CDIF1 = 10 0110.
CDIF2 = 0 0000.

Application Example 5

IS-54 application.

Codec sampling rate, f

: 8 kHz.

CLK needs to be a common multiple of 8 kHz and
48 x 48.6 kHz.

Need phase adjustment.

Solution:

48 x 48.6 = 2.3328E6
2.3328E6 ÷ 8E3 = 291.6
To get a common multiple, 291.6 must be multiplied
by a factor to become an integer: 291.6 x 10 = 2916
(an integer). So, CLK/CK

= 2916 and CLK/ICLK0 =

2 CLK = 2916 x 8E3 = 23.328E6.

CLK input clock rate: 23.328 MHz.

Set CDIV0 for ÷2 (CDIV0 = 1) to allow advance/
retard for phase adjustment.

ICLK0 internal clock rate is 11.664 MHz.

Using the equations from Section 4.5 on page 16,

Hence, M = 12, S = 1, and N = 42.

Using Tables 2 through 4 on page 18:
CDIV3 = 00 1100.
CDIFS = 1.
CDIF0 = 00 0011.
CDIF1 = 11 0110.
CDIF2 = 0 0000.

ICLK0

-----------------

125

(

)

(

)

ICL K0

-----------------

2.048 MHz

8 kHz

------------------------------

256

125

(

)

(

)

ICLK0

-----------------

11.664 MHz

8 kHz

---------------------------------

1458

125

(

)

(

)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

8 Device Characteristics

8.1 Absolute Maximum Ratings

Stresses in excess of the absolute maximum ratings can cause permanent damage to the device. These are abso-
lute stress ratings only. Functional operation of the device is not implied at these or any other conditions in excess
of those given in the operational sections of the data sheet. Exposure to absolute maximum ratings for extended
periods can adversely affect device reliability.

External leads can be bonded and soldered safely at temperatures of up to 300 °C.

Voltage Range on any Pin with Respect to Ground................................................................. 0.5 V to +6 V
Power Dissipation ................................................................................................................................. 0.3 W
Ambient Temperature Range...............................................................................................40 °C to +85 °C
Storage Temperature Range

.......................................................................................................

65 °C to +150 °C

8.2 Handling Precautions

All MOS devices must be handled with certain precautions to avoid damage due to the accumulation of static
charge. Although input protection circuitry has been incorporated into the devices to minimize the effect of this
static buildup, proper precautions should be taken to avoid exposure to electrostatic discharge during handling and
mounting. Lucent Technologies employs a human-body model for ESD-susceptibility testing. Since the failure volt-
age of electronic devices is dependent on the current, voltage, and, hence, the resistance and capacitance, it is
important that standard values be employed to establish a reference by which to compare test data. Values of
100 pF and 1500

are the most common and are the values used in the Lucent human-body model test circuit.

The breakdown voltage for the CSP1027 is greater than 1000 V.

8.3 Recommended Operating Conditions

8.3.1 Package Thermal Considerations

The recommended operating temperature specified above is based on the maximum power, package type, and
maximum junction temperature. The following equation describes the relationship between these parameters. Cer-
tain applications' maximum power may be less than the worst-case value and can use this relationship to determine
the maximum ambient temperature allowed.

= T

P x

Maximum Junction Temperature (T

) in 44-Pin QFP ...........................................................................125 °C

44-pin QFP Maximum Thermal Resistance in Still-Air-Ambient (

).............................................. 39 °C/W

Maximum Junction Temperature (T

) in 48-Pin TQFP......................................................................... 125 °C

48-pin TQFP Maximum Thermal Resistance in Still-Air-Ambient (

) ........................................... 90 °C/W

Table 11. Recommended Operating Conditions

Parameter

Symbol

Min

Max

Unit

Analog Supply Voltage

DDA

4.5

5.5

Digital Supply Voltage

2.7

5.5

Ambient Temperature

°C

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

9 Electrical Characteristics and Requirements

The following electrical characteristics are preliminary and are subject to change. Electrical characteristics refer to
the behavior of the device under specified conditions. Electrical requirements refer to conditions imposed on the
user for proper operation of the device. The parameters below are valid for the following conditions:

= 5 V ± 10% (See Section 8.3 on page 47 for exceptions.)

Table 12. Digital Electrical Characteristics and Requirements

Parameter

Symbol

Min

Max

Unit

Input Voltage (except PORCAP):

Low
High

0.7 x V

0.3 x V

V
V

PORCAP Input Voltage:

Low
High

0.7 x V

0.5 x V

V
V

Input Current (except EIGS, IOCK, PORCAP):

Low (V

= 0 V)

High (V

= 5.5 V)

µA
µA

Input Current (EIGS):

Low (V

= 0 V)

High (V

= 5.5 V)

100

µA
µA

Input Current (IOCK):

Low (V

= 0 V)

High (V

= 5.5 V)

100

µA
µA

Input Current (PORCAP):

Low (V

= 0 V)

High (V

= 5.5 V)

100

µA
µA

Output Low Voltage:

Low (I

= 2.0 mA)

Low (I

= 50 µA)

--
--

0.4
0.2

V
V

Output High Voltage:

High (I

= 2.0 mA)

High (I

= 50 µA)

0.7

0.2

--
--

V
V

Output 3-State Current:

Low (V

= 5.5 V, V

= 0 V)

High (V

= 5.5 V, V

= 5.5 V)

OZL

OZH

µA
µA

Input Capacitance

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

9 Electrical Characteristics and Requirements

(continued)

9.1 Power Dissipation

Power dissipation is highly dependent on the frequency of operation. The typical power dissipation listed is for a
selected application. The following electrical characteristics are preliminary and are subject to change.

The power dissipation listed is for internal power dissipation only. Total power dissipation can be calculated on the
basis of the application by adding C x V

x f for each output, where C is the additional load capacitance and f is

the effective output frequency.

Power dissipation due to the input and I/O buffers is highly dependent on the input voltage level. At full CMOS lev-
els, essentially no dc current is drawn. However, for levels near the threshold of 0.5 x V

, high and unstable levels

can flow. Therefore, all unused input pins should be tied inactive to V

or V

, and all unused I/O pins should be

tied inactive through a 10 k

resistor to V

or V

. Table 13 shows the input buffer power dissipation for inputs at

dc levels, V

DD or

Table 13. Power Dissipation

Operating Mode

Analog Supply

DDA

)

Digital Supply

)

5.0 V

3.3 V

3.0 V

Typ

5.0 V

Max

5.5 V

Typ

5.0 V

Max

5.5 V

Typ

3.3 V

Max

3.6 V

Typ

3.0 V

Max

3.3 V

Codec Active,
Crystal Osc. Disabled

(cioc0: ACTIVE = 1,
XOSCEN = 0,
CLK at 25 MHz,
IOCK at 6.25 MHz,
CK

at 1 MHz)

11.0 mA

55.0 mW

12.2 mA

67.1 mW

4.6 mA

23.0 mW

5.8 mA

31.9 mW

3.0 mA

9.9 mW

3.8 mA

13.7 mW

2.8 mA

8.4 mW

3.5 mA

11.5 mW

Codec Inactive,
Crystal Osc. Disabled

(cioc0: ACTIVE = 0,
XOSCEN = 0,
CLK at 25 MHz,
IOCK at 6.25 MHz)

0.01 mA

0.05 mW

0.02 mA

0.11 mW

3.7 mA

18.5 mW

4.9 mA

26.9 mW

2.5 mA

8.2 mW

3.2 mA

11.5 mW

2.2 mA

6.6 mW

2.9 mA

9.6 mW

Codec Active,
Crystal Osc. Enabled

(cioc0: ACTIVE = 1,
XOSCEN = 1,
25 MHz crystal,
IOCK at 6.25 MHz,
CK

at 1 MHz)

11.0 mA

55.0 mW

12.2 mA

67.1 mW

10.2 mA

51.0 mW

11.4 mA

62.7 mW

4.1 mA

13.5 mW

5.0 mA

18.0 mW

3.9 mA

11.7 mW

4.7 mA

15.5 mW

Codec Inactive,
Crystal Osc. Enabled

(cioc0: ACTIVE = 0,
XOSCEN = 1,
25 MHz crystal,
IOCK at 6.25 MHz)

0.01 mA

0.05 mW

0.02 mA

0.11 mW

9.3 mA

46.5 mW

10.5 mA

57.8 mW

3.6 mA

11.9 mW

4.4 mA

15.8 mW

3.3 mA

9.9 mW

4.1 mA

13.5 mW

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

10 Analog Characteristics and Requirements

The following analog characteristics and requirements are preliminary information and are subject to change. Ana-
log characteristics refer to the behavior of the device under specified conditions. Analog requirements refer to con-
ditions imposed on the user for proper operation of the device. All analog data is valid for the following conditions
unless otherwise specified:

= 40 °C to +85 °C.

DDA

= 5 V ± 10% (See Section 8.2 on page 47.)

Sampling frequency = 8 kHz, oversampling clock (CK

) = 1.0 MHz, input clock (CLK) = 25 MHz.

0.22 µF capacitors connected to the REFC pin.

0.15 µF coupling capacitors connected to the MICIN and AUXIN pins when EIGS = 0.

Rfb = Rin = 24 k

, Cfb = 270 pF, and Cin = 1 µF when in the external input gain select mode (EIGS = 1).

2 k

differential output load connected between AOUTP and AOUTN pins.

1 µF and 0.1 µF bypass capacitors connected between the V

REG

and V

SSA

0 dBm0 is the level that corresponds to a sine wave that is 3.14 dB below the clipping (overload) level at the out-
put of the A/D or input to the D/A.

All noise and distortion measurements are flat weighted and integrated over the 300 Hz to 4 kHz frequency band.

10.1 Analog Input and Microphone Regulator

Note: The input clipping level corresponds to an A/D path output of 3.14 dBm0.

Note: V

REG

must be bypassed to analog ground through a 1 µF low ESR capacitor to meet this noise specification. The minimum bypass capac-

itance for stable V

REG

operation is 0.1 µF, with a maximum noise of 200 µVrms.

Table 14. Analog Input Characteristics and Requirements

Symbol

Parameter

Min

Typ

Max

Unit

Rin

A/D Input Resistance of AUXIN and MICIN:

EIGS = 0
EIGS = 1

1000

--
--

Cin

A/D Input Capacitance on AUXIN and MICIN

A/D Input Clipping Level at AUXIN or MICIN:

EIGS = 0, cioc0: IRSEL = 0
EIGS = 0, cioc0: IRSEL = 1

0.490
0.143

0.500
0.160

0.510
0.180

Vp
Vp

A/D Input Clipping Level at AUXIN:

EIGS = 1

1.49

1.578

1.67

Gain Referred to Nominal Clipping Level:

EIGS = 0, cioc0: IRSEL = 0
EIGS = 0, cioc0: IRSEL = 1
EIGS = 1

0.18

1.0
0.5

0
0
0

0.18

1.0
0.5

dB
dB
dB

Table 15. Microphone Regulator Characteristics

Parameter

Min

Typ

Max

Unit

REG

Output Voltage

2.7

3.0

3.3

Vrms

REG

Output Current

250

µA

REG

Output Noise

100

µVrms

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

10 Analog Characteristics and Requirements

(continued)

10.2 Analog-to-Digital Path

Notes:
The signal to distortion plus noise ratio is from the MICIN or AUXIN input to the PCM output when EIGS = 0, and from Vin (of Figure 5 on page
7) to PCM output when EIGS = 1.

The A/D signal to distortion plus noise ratio is valid for sampling rates up to 16 kHz. For sampling rates between 16 kHz and 24 kHz, the signal
to distortion plus noise ratio is no better than 60 dB.

* Gain is relative to the 0 dBm0 signal level with EIGS = 0 and IRSEL = 0.

Note: The frequency response scales linearly with codec oversampling clock rate. Frequencies greater than 4000 Hz are affected by antialias

filtering attenuation.

Table 16. A/D Signal to Distortion Plus Noise Ratio

Output Signal Level

Preamp

(EIGS = 0)

0.5 Vp Range

(cioc0: IRSEL = 0)

Preamp

(EIGS = 0)

0.16 Vp Range

(cioc0: IRSEL = 1)

Ext. Gain Select

(EIGS = 1)

1.578 Vp Range

Unit

Min

Max

Min

Max

Min

Max

0 dBm0

10 dBm0

30 dBm0

40 dBm0

45 dBm0

55 dBm0

Table 17. A/D Relative Gain Accuracy*

Output Signal Level

Min

Max

Unit

+3 dBm0 to 40 dBm0

0.1

40 dBm0 to 50 dBm0

0.4

50 dBm0 to 55 dBm0

1.2

Table 18. A/D Frequency Response Relative to 1 kHz Output Level (f

= 1 MHz and f

= 8 kHz)

Frequency

High-Pass Filter Enabled

(HPFE = 0)

High-Pass Filter Disabled

(HPFE = 1)

Unit

Min

Max

Min

Max

50 Hz

0.25

60 Hz

0.25

100 Hz

0.25

200 Hz

0.25

300 Hz

0.25

3000 Hz

0.25

0.4

0.25

3400 Hz

0.9

0.25

0.9

0.25

4000 Hz

4600 Hz

8000 Hz

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

10 Analog Characteristics and Requirements

(continued)

10.3 Digital-to-Analog Path

Notes:
The D/A signal to distortion plus noise ratio decreases by 3 dB for each 3 dB gain step below 0 dB. The D/A SDNR is specified with a differential
load. For a single-ended load, the D/A SDNR is degraded by about 6 dB.

The D/A signal to distortion plus noise ratio is valid for sampling rates up to 16 kHz. For sampling rates between 16 kHz and 24 kHz, the signal
to distortion plus noise ratio is no better than 60 dB.

Table 19. D/A Signal to Distortion Plus Noise Ratio (0 dB Output Setting)

Output Signal Level

Min

Max

Unit

0 dBm0

10 dBm0

30 dBm0

40 dBm0

45 dBm0

55 dBm0

Table 20. D/A Relative Gain Accuracy*

* Gain is relative to the 0 dBm0 signal level. Absolute gain accuracy at the 0 dBm0 signal level is ±0.18 dB. Digital-to-analog path gain is spec-

ified with a differential load; a single-ended load adds ±0.1 dB to absolute gain.

Output Signal Level

Min

Max

Unit

+3 dBm0 to 40 dBm0

0.1

40 dBm0 to 50 dBm0

0.4

50 dBm0 to 55 dBm0

1.2

Table 21. D/A Output Gain Adjustment

Gain Setting

Min

Typ

Max

Unit

0 dB

0.2

3 dB

3.21

3.01

2.81

6 dB

6.22

6.02

5.82

9 dB

9.23

9.03

8.83

12 dB

12.24

12.04

11.84

.
.
.

42 dB

42.34

42.14

41.94

45 dB

45.35

45.15

44.95

CSP1027 Voice Band Codec for

Data Sheet

Cellular Handset and Modem Applications

December 1999

Lucent Technologies Inc.

10 Analog Characteristics and Requirements

(continued)

10.4 Miscellaneous

* The codec is intended to drive a floating 2 k

load, such as a telephone handset speaker, or 1 k

loads ac-coupled to ground on both ana-

log outputs. Since the codec outputs AOUTP and AOUTN have common-mode dc voltage, ac coupling must be used if there is a dc path to
V

or V

(ground) through the load.

Table 22. D/A Frequency Response Relative to 1 kHz Output Level (f

= 1 MHz and f

= 8 kHz)

Frequency

High-Pass Filter Enabled

(HPFE = 0)

High-Pass Filter Disabled

(HPFE = 1)

Unit

Min

Max

Min

Max

50 Hz

0.25

60 Hz

0.25

100 Hz

0.25

200 Hz

0.25

300 Hz

0.25

3000 Hz

0.25

3400 Hz

0.9

0.25

0.9

0.25

4000 Hz

4600 Hz

8000 Hz

Table 23. Other Analog Characteristics and Requirements*

Parameter

Min

Max

Unit

D/A Differential Output Resistance (0 kHz to 4 kHz)

D/A Single-ended Output Resistance (0 kHz to 4 kHz)

Analog-to-Digital Power Supply Rejection Ratio at 3 kHz

Digital-to-Analog Power Supply Rejection Ratio at 3 kHz

Analog Input Coupling Capacitor Input Leakage Current

Idle Channel Noise at Analog-to-Digital Output with Input Gain Setting of

500 mVp or 160 mVp

dBm0

Idle Channel Noise at Digital-to-Analog Output

300

µVrms

A/D to D/A and D/A to A/D Crosstalk

Digital-to-Analog Image Frequency Attenuation Above 4600 Hz

Digital-to-Analog Output Amplifier Differential Swing for 2 k

Load

1.5

Vrms

Codec Filter Group Delay for Frequencies Less than 800 Hz

2.8

Codec Filter Group Delay for Frequencies Greater than 800 Hz

0.8

Recovery Time of Digital-to-Analog Output Due to a Change from Inac-

tive Mode to Active Mode, Muted to Not Muted, or Change in Output
Gain (See cioc0 register, ACTIVE, MUTE, OGSEL.)

100

Recovery Time of Analog-to-Digital PCM Output and V

REG

Due to a

Change from Inactive Mode to Active Mode (See cioc0 register,
ACTIVE.)

600

Recovery Time of Analog Circuits Due to a Change in Input Select or

Input Range (See cioc0 register, INSEL and IRSEL.)

100

Allowable CLK Input Jitter

Allowable CLK Frequency Error

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

11 Timing Characteristics and Requirements

The following timing characteristics and requirements are preliminary information and are subject to change. Tim-
ing characteristics refer to the behavior of the device under specified conditions. Timing requirements refer to con-
ditions imposed on the user for proper operation of the device. All timing data is valid for the following conditions:

= 40 °C to +85 °C or 0 °C to 70 °C (See Section 8.2 on page 47.)

= 5 V ± 0.5 V, 3.3 V ± 0.3 V, or 3.0 V ± 0.3 V, V

= 0 V (See Section 8.2 on page 47.)

Capacitance load on outputs (C

) = 50 pF

Output characteristics can be derated as a function of load capacitance (C

All outputs: dt/dC

0.06 ns/pF for 0

100 pF at V

for rising edge

dt/dC

0.05 ns/pF for 0

100 pF at V

for falling edge

For example, if the actual load capacitance is 30 pF instead of 50 pF, the derating for a rising edge is
(30 pF 50 pF) x 0.06 ns/pF = 1.2 ns less than the specified rise time or delay which includes a rise time.

Test conditions for inputs:

Rise and fall times of 4 ns or less

Timing reference levels for delays = V

, V

Test conditions for outputs:

LOAD

= 50 pF

Timing reference levels for delays = V

, V

3-state delays measured to the high-impedance state of the output driver

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

11 Timing Characteristics and Requirements

(continued)

11.4 Serial I/O Communication

Figure 42. Serial Input/Output Timing Diagram

Table 29. Timing Requirements for Serial Input/Output

Abbreviated

Reference

Parameter

5.0 V

3.3 V

3.0 V

Unit

Min

Max

Min

Max

Min

Max

t11

Clock Period (high to high)

--*

* Device is fully static; t11 is tested at 100 ns. The frequency of IOCK must be greater than the frequency of the internal oversampling clock

CKOS (F

CKOS

--*

120

--*

t12

Clock Low Time (low to change)

t13

Clock High Time (high to change)

t14

Sync High Setup (high to high)

t15

Sync Low Setup (low to high)

t16

Sync Hold (high to invalid)

t17

Data Setup (valid to high)

t18

Data Hold (high to invalid)

t19

Address Setup (valid to high)

t20

Address Hold (high to invalid)

Table 30. Timing Characteristics for Serial Input/Output

Abbreviated

Reference

Parameter

5.0 V

3.3 V

3.0 V

Unit

Min

Max

Min

Max

Min

Max

t21

Data/Status Delay (high to valid)

t22

Data/Status Hold (high to invalid)

t11

t13

t12

t14

t16

t15

t16

t17

t18

t19

t20

t21

t22

t20

t18

SYNC

SADD

IOCK

B15

B14

AD0

AD1

AS0

AS7

B15

B14

5-7621 (F)

Lucent Technologies Inc.

Data Sheet

CSP1027 Voice Band Codec for

December 1999

Cellular Handset and Modem Applications

11 Timing Characteristics and Requirements

(continued)

11.5 Serial Multiprocessor Communication

Figure 44. SIO Multiprocessor Timing Diagram

Note: Capacitance load on DO, SYNC, and SADD = 100 pF.

Table 32. Timing Requirements for Multiprocessor Communication

Abbreviated

Reference

Parameter

5.0 V

3.3 V

3.0 V

Unit

Min

Max

Min

Max

Min

Max

t25

Sync High Setup (high to high)

t26

Sync Low Setup (low to high)

t27

Sync Hold (high to invalid)

Table 33. Timing Characteristics for Multiprocessor Communication

Abbreviated

Reference

Parameter

5.0 V

3.3 V

3.0 V

Unit

Min

Max

Min

Max

Min

Max

t28

Data Delay (bit 0 only) (low to valid)

t29

Data Disable Delay (high to 3-state)

t30

Address Delay (bit 0 only) (low to valid)

t31

Address Delay (high to valid)

t32

Address Hold (low to valid)

t33

Address Disable Delay (high to 3-state)

B15

B14

B15

AD0

AD1

AD7

AS0

AD0

AS7

IOCK

SYNC

DO/DI

SADD

t27

t26

t27

t25

t28

t31

t30

t32

t33

t29

5-7623 (F)

Lucent Technologies Inc. reserves the right to make changes to the product(s) or information contained herein without notice. No liability is assumed as a result of their use or application. No
rights under any patent accompany the sale of any such product(s) or information.

December 1999
DS00-063AUTO (Replaces DS99-081WDSP)

For additional information, contact your Microelectronics Group Account Manager or the following:
INTERNET:

http://www.lucent.com/micro

E-MAIL:

docmaster@micro.lucent.com

N. AMERICA: Microelectronics Group, Lucent Technologies Inc., 555 Union Boulevard, Room 30L-15P-BA, Allentown, PA 18103

1-800-372-2447, FAX 610-712-4106 (In CANADA: 1-800-553-2448, FAX 610-712-4106)

ASIA PACIFIC: Microelectronics Group, Lucent Technologies Singapore Pte. Ltd., 77 Science Park Drive, #03-18 Cintech III, Singapore 118256

Tel. (65) 778 8833, FAX (65) 777 7495

CHINA:

Microelectronics Group, Lucent Technologies (China) Co., Ltd., A-F2, 23/F, Zao Fong Universe Building, 1800 Zhong Shan Xi Road, Shanghai
200233 P. R. China Tel. (86) 21 6440 0468, ext. 316, FAX (86) 21 6440 0652

JAPAN:

Microelectronics Group, Lucent Technologies Japan Ltd., 7-18, Higashi-Gotanda 2-chome, Shinagawa-ku, Tokyo 141, Japan
Tel. (81) 3 5421 1600, FAX (81) 3 5421 1700

EUROPE:

Data Requests: MICROELECTRONICS GROUP DATALINE: Tel. (44) 7000 582 368, FAX (44) 1189 328 148
Technical Inquiries:GERMANY: (49) 89 95086 0 (Munich), UNITED KINGDOM: (44) 1344 865 900 (Ascot),

FRANCE: (33) 1 40 83 68 00 (Paris), SWEDEN: (46) 8 594 607 00 (Stockholm), FINLAND: (358) 9 4354 2800 (Helsinki),
ITALY: (39) 02 6608131 (Milan), SPAIN: (34) 1 807 1441 (Madrid)

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: CSP1027

Document Outline

Ð­Ð»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: CSP1027

Document Outline

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: CSP1027