ADS1210, 1211

ADS1210

ADS1211

24-Bit ANALOG-TO-DIGITAL CONVERTER

FEATURES

DELTA-SIGMA A/D CONVERTER

23 BITS EFFECTIVE RESOLUTION AT 10Hz
AND 20 BITS AT 1000Hz

DIFFERENTIAL INPUTS

PROGRAMMABLE GAIN AMPLIFIER

FLEXIBLE SPI COMPATIBLE SSI
INTERFACE WITH 2-WIRE MODE

PROGRAMMABLE CUT-OFF FREQUENCY
UP TO 15.6kHz

INTERNAL/EXTERNAL REFERENCE

ON CHIP SELF-CALIBRATION

ADS1211 INCLUDES 4 CHANNEL MUX

DESCRIPTION

The ADS1210 and ADS1211 are precision, wide
dynamic range, delta-sigma analog-to-digital converters
with 24-bit resolution operating from a single +5V
supply. The differential inputs are ideal for direct
connection to transducers or low level voltage sig-
nals. The delta-sigma architecture is used for wide
dynamic range and to guarantee 22 bits of no missing
code performance. An effective resolution of 23 bits
is achieved through the use of a very low-noise input
amplifier at conversion rates up to 10Hz. Effective
resolutions of 20 bits can be maintained up to a
sample rate of 1kHz through the use of the unique
Turbo modulator mode of operation. The dynamic
range of the converters is further increased by provid-
ing a low-noise programmable gain amplifier with a
gain range of 1 to 16 in binary steps.

The ADS1210 and ADS1211 are designed for high
resolution measurement applications in smart trans-
mitters, industrial process control, weigh scales, chro-
matography and portable instrumentation. Both con-
verters include a flexible synchronous serial interface
which is SPI compatible and also offers a two-wire
control mode for low cost isolation.

The ADS1210 is a single channel converter and is
offered in both 18-pin DIP and 18-lead SOIC pack-
ages. The ADS1211 includes a 4 channel input multi-
plexer and is available in 24-pin DIP, 24-lead SOIC,
and 28-lead SSOP packages.

APPLICATIONS

INDUSTRIAL PROCESS CONTROL

INSTRUMENTATION

BLOOD ANALYSIS

SMART TRANSMITTERS

PORTABLE INSTRUMENTS

WEIGH SCALES

PRESSURE TRANSDUCERS

ADS1211 Only

ADS1210/11

ADS1210

ADS1211

ADS1210

ADS1211

PGA

+2.5V

Reference

+3.3V Bias

Generator

Clock Generator

Serial Interface

Second-Order

Modulator

Instruction Register
Command Register

Data Output Register

Offset Register

Full-Scale Register

Third-Order

Digital Filter

Micro Controller

Modulator Control

AGND AV

REF

OUT

REF

BIAS

OUT

MODE

DSYNC

DRDY

SCLK

DGND

SDIO

SDOUT

MUX

1996 Burr-Brown Corporation

PDS-1284E

Printed in U.S.A. May, 2000

International Airport Industrial Park · Mailing Address: PO Box 11400, Tucson, AZ 85734 · Street Address: 6730 S. Tucson Blvd., Tucson, AZ 85706 · Tel: (520) 746-1111

Twx: 910-952-1111 · Internet: http://www.burr-brown.com/ · Cable: BBRCORP · Telex: 066-6491 · FAX: (520) 889-1510 · Immediate Product Info: (800) 548-6132

ADS1210, 1211

All specifications T

MIN

to T

MAX

, AV

= DV

= +5V, f

XIN

= 10MHz, programmable gain amplifier setting of 1, Turbo Mode Rate of 1, REF

OUT

disabled,V

BIAS

disabled,

and external 2.5V reference, unless otherwise specified.

The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes
no responsibility for the use of this information, and all use of such information shall be entirely at the user's own risk. Prices and specifications are subject to change
without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant
any BURR-BROWN product for use in life support devices and/or systems.

SPECIFICATIONS

ADS1210U, P/ADS1211U, P, E

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

ANALOG INPUT
Input Voltage Range

(1)

With V

BIAS

(2)

+10

Input Impedance

G = Gain, TMR = Turbo Mode Rate

4/(G · TMR)

(3)

Programmable Gain Amplifier

User Programmable: 1, 2, 4, 8, or 16

Input Capacitance

Input Leakage Current

At +25

At T

MIN

to T

MAX

SYSTEMS PERFORMANCE
Resolution

Bits

No Missing Codes

DATA

= 60Hz

Bits

Integral Linearity

DATA

= 60Hz

0.0015

%FSR

DATA

= 1000Hz, TMR of 16

0.0015

%FSR

Unipolar Offset Error

(4)

See Note 5

Unipolar Offset Drift

(6)

Gain Error

(4)

See Note 5

Gain Error Drift

(6)

Common-Mode Rejection

(9)

At DC, +25

100

115

At DC, T

MIN

to T

MAX

115

50Hz, f

DATA

= 50Hz

(7)

160

60Hz, f

DATA

= 60Hz

(7)

160

Normal-Mode Rejection

50Hz, f

DATA

= 50Hz

(7)

100

60Hz, f

DATA

= 60Hz

(7)

100

Output Noise

See Typical Performance Curves

Power Supply Rejection

DC, 50Hz, and 60Hz

VOLTAGE REFERENCE
Internal Reference (REF

OUT

)

2.4

2.5

2.6

Drift

ppm/

Noise

Vp-p

Load Current

Source or Sink

Output Impedance

External Reference (REF

)

2.0

3.0

Load Current

2.5

BIAS

Output

Using Internal Reference

3.15

3.3

3.45

Drift

ppm/

Load Current

Source or Sink

10mA

DIGITAL INPUT/OUTPUT
Logic Family

TTL Compatible CMOS

Logic Level: (all except X

)

= +5

2.0

+0.3

= +5

0.3

0.8

= 2 TTL Loads

2.4

= 2 TTL Loads

0.4

Input Levels: V

3.5

+0.3

0.3

0.8

Frequency Range (f

XIN

)

0.5

MHz

Output Data Rate (f

DATA

)

User Programmable

2.4

15,625

XIN

= 500kHz

0.12

781

Data Format

User Programmable

Two's Complement

or Offset Binary

SYSTEM CALIBRATION
Offset and Full-Scale Limits

= Full-Scale Differential Voltage

(8)

0.7 · (2 · REF

)/G

| V

= Offset Differential Voltage

(8)

1.3 · (2 · REF

)/G

ADS1210, 1211

POWER SUPPLY REQUIREMENTS
Power Supply Voltage

4.75

5.25

Power Supply Current:

Analog Current

Digital Current

3.5

Additional Analog Current with

REF

OUT

Enabled

1.6

BIAS

Enabled

No Load

Power Dissipation

TMR of 16

XIN

= 2.5MHz

XIN

= 2.5MHz, TMR of 16

Sleep Mode

TEMPERATURE RANGE
Specified

+85

Storage

+125

NOTES: (1) In order to achieve the converter's full-scale range, the input must be fully differential (A

N = 2 · REF

P). If the input is single-ended (A

N or

P is fixed), then the full scale range is one-half that of the differential range. (2) This range is set with external resistors and V

BIAS

(as described in the text).

Other ranges are possible. (3) Input impedance is higher with lower f

XIN

. (4) Applies after calibration. (5) After system calibration, these errors will be of the order

of the effective resolution of the converter. Refer to the Typical Performance Curves which apply to the desired mode of operation. (6) Recalibration can remove
these errors. (7) The specification also applies at f

DATA

/i, where i is 2, 3, 4, etc. (8) Voltages at the analog inputs must remain within AGND to AV

. (9) The common-

mode rejection test is performed with a 100mV differential input.

SPECIFICATIONS

(CONT)

ADS1210U, P/ADS1211U, P, E

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

PACKAGE/ORDERING INFORMATION

PACKAGE

DRAWING

TEMPERATURE

PRODUCT

PACKAGE

NUMBER

(1)

RANGE

ADS1210P

18-Pin Plastic DIP

218

C to +85

ADS1210U

18-Lead SOIC

219

C to +85

ADS1211P

24-Pin Plastic DIP

243

C to +85

ADS1211U

24-Lead SOIC

239

C to +85

ADS1211E

28-Lead SSOP

324

C to +85

NOTE: (1) For detailed drawing and dimension table, please see end of data
sheet, or Appendix C of Burr-Brown IC Data Book.

ELECTROSTATIC
DISCHARGE SENSITIVITY

This integrated circuit can be damaged by ESD. Burr-Brown
recommends that all integrated circuits be handled with ap-
propriate precautions. Failure to observe proper handling and
installation procedures can cause damage.

Electrostatic discharge can cause damage ranging from
performance degradation to complete device failure. Burr-
Brown Corporation recommends that all integrated circuits be
handled and stored using appropriate ESD protection
methods.

Analog Input: Current ................................................

100mA, Momentary

10mA, Continuous

Voltage ................................... AGND 0.3V to AV

+0.3V

to DV

........................................................................... 0.3V to 6V

to AGND ......................................................................... 0.3V to 6V

to DGND ......................................................................... 0.3V to 6V

AGND to DGND ................................................................................

0.3V

REF

Voltage to AGND ............................................ 0.3V to AV

+0.3V

Digital Input Voltage to DGND .................................. 0.3V to DV

+0.3V

Digital Output Voltage to DGND ............................... 0.3V to DV

+0.3V

Lead Temperature (soldering, 10s) .............................................. +300

Power Dissipation (Any package) .................................................. 500mW

ABSOLUTE MAXIMUM RATINGS

All specifications T

MIN

to T

MAX

, AV

= DV

= +5V, f

XIN

= 10MHz, programmable gain amplifier setting of 1, Turbo Mode Rate of 1, REF

OUT

disabled,V

BIAS

disabled,

and external 2.5V reference, unless otherwise specified.

ADS1210, 1211

PGA

+2.5V

Reference

+3.3V Bias

Generator

Clock Generator

Serial Interface

Second-Order

Modulator

Instruction Register
Command Register

Data Output Register

Offset Register

Full-Scale Register

Third-Order

Digital Filter

Micro Controller

Modulator Control

AGND

REF

OUT

REF

BIAS

OUT

DSYNC

DRDY

MODE

SCLK

DGND

SDIO

SDOUT

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

ADS1210 SIMPLIFIED BLOCK DIAGRAM

ADS1210 PIN DEFINITIONS

PIN NO

NAME

DESCRIPTION

Noninverting Input.

Inverting Input.

AGND

Analog Ground.

BIAS

Bias Voltage Output, +3.3V nominal.

Chip Select Input.

DSYNC

Control Input to Synchronize Serial Output Data.

System Clock Input.

OUT

System Clock Output (for Crystal or Resonator).

DGND

Digital Ground.

Digital Supply, +5V nominal.

SCLK

Clock Input/Output for serial data transfer.

SDIO

Serial Data Input (can also function as Serial Data
Output).

SDOUT

Serial Data Output.

DRDY

Data Ready.

MODE

SCLK Control Input (Master = 1, Slave = 0).

Analog Supply, +5V nominal.

REF

OUT

Reference Output, +2.5V nominal.

REF

Reference Input.

TOP VIEW

DIP/SOIC

ADS1210 PIN CONFIGURATION

ADS1210, 1211

PGA

+2.5V

Reference

+3.3V Bias

Generator

Clock Generator

Serial Interface

Second-Order

Modulator

Third-Order

Digital Filter

Modulator Control

AGND

REF

OUT

REF

BIAS

OUT

DSYNC

DRDY

MODE

SCLK

DGND

SDIO

SDOUT

MUX

Instruction Register
Command Register

Data Output Register

Offset Register

Full-Scale Register

Micro Controller

ADS1211 SIMPLIFIED BLOCK DIAGRAM

ADS1211P AND ADS1211U PIN DEFINITIONS

TOP VIEW

DIP/SOIC

ADS1211P AND ADS1211U PIN CONFIGURATION

PIN NO

NAME

DESCRIPTION

Inverting Input Channel 3.

Noninverting Input Channel 2.

Inverting Input Channel 2.

Noninverting Input Channel 1.

Inverting Input Channel 1.

AGND

Analog Ground.

BIAS

Bias Voltage Output, +3.3V nominal.

Chip Select Input.

DSYNC

Control Input to Synchronize Serial Output Data.

System Clock Input.

OUT

System Clock Output (for Crystal or Resonator).

DGND

Digital Ground.

Digital Supply, +5V nominal.

SCLK

Clock Input/Output for serial data transfer.

SDIO

Serial Data Input (can also function as Serial Data

Output).

SDOUT

Serial Data Output.

DRDY

Data Ready.

MODE

SCLK Control Input (Master = 1, Slave = 0).

Analog Supply, +5V nominal.

REF

OUT

Reference Output: +2.5V nominal.

REF

Reference Input.

Noninverting Input Channel 4.

Inverting Input Channel 4.

Noninverting Input Channel 3.

ADS1211P
ADS1211U

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

ADS1210, 1211

ADS1211E

AGND

BIAS

NIC

DSYNC

OUT

DGND

REF

OUT

MODE

NIC

DRDY

SDOUT

SDIO

SCLK

ADS1211E PIN DEFINITIONS

PIN NO

NAME

DESCRIPTION

Inverting Input Channel 3.

Noninverting Input Channel 2.

Inverting Input Channel 2.

Noninverting Input Channel 1.

Inverting Input Channel 1.

AGND

Analog Ground.

BIAS

Bias Voltage Output, +3.3V nominal.

NIC

Not Internally Connected.

NIC

Not Internally Connected.

Chip Select Input.

DSYNC

Control Input to Synchronize Serial Output Data.

System Clock Input.

OUT

System Clock Output (for Crystal or Resonator).

DGND

Digital Ground.

Digital Supply, +5V nominal.

SCLK

Clock Input/Output for serial data transfer.

SDIO

Serial Data Input (can also function as Serial Data

Output).

SDOUT

Serial Data Output.

DRDY

Data Ready.

NIC

Not Internally Connected.

NIC

Not Internally Connected.

MODE

SCLK Control Input (Master = 1, Slave = 0).

Analog Supply, +5V nominal.

REF

OUT

Reference Output: +2.5V nominal.

REF

Reference Input.

Noninverting Input Channel 4.

Inverting Input Channel 4.

Noninverting Input Channel 3.

ADS1211E PIN CONFIGURATION

TOP VIEW

SSOP

ADS1210, 1211

TYPICAL PERFORMANCE CURVES

At T

= +25

C, AV

= DV

DD =

+5V, f

XIN

= 10MHz, programmable gain amplifier setting of 1, Turbo Mode Rate of one, REF

OUT

disabled, V

BIAS

disabled, and external

2.5V reference, unless otherwise noted.

RMS NOISE vs INPUT VOLTAGE LEVEL

(60Hz Data Rate)

Analog Input Differential Voltage (V)

5.0 4.0 3.0 2.0 1.0

1.0

2.0

3.0

4.0

5.0

RMS Noise (ppm)

2.5

2.0

1.5

1.0

0.5

EFFECTIVE RESOLUTION vs DATA RATE

(2.5MHz Clock)

Data Rate (Hz)

100

Effective Resolution in Bits (rms)

Turbo 1

Turbo 2

Turbo 4

Turbo 8

Turbo 16

EFFECTIVE RESOLUTION vs DATA RATE

(5MHz Clock)

Data Rate (Hz)

100

Effective Resolution in Bits (rms)

Turbo 1

Turbo 2

Turbo 4

Turbo 8

Turbo 16

EFFECTIVE RESOLUTION vs DATA RATE

(10MHz Clock)

Data Rate (Hz)

100

Effective Resolution in Bits (rms)

Turbo 1

Turbo 2

Turbo 4

Turbo 8

Turbo 16

EFFECTIVE RESOLUTION vs DATA RATE

Data Rate (Hz)

100

Effective Resolution in Bits (rms)

PGA 1

PGA 2

PGA 4

PGA 16

PGA 8

EFFECTIVE RESOLUTION vs DATA RATE

(1MHz Clock)

Data Rate (Hz)

100

Effective Resolution in Bits (rms)

Turbo 1

Turbo 2

Turbo 4

Turbo 8

Turbo 16

ADS1210, 1211

ANALOG

ANALOG INPUT

INPUT

(1)

UTILIZING V

BIAS

(1,2)

FULL-

EXAMPLE

FULL-

EXAMPLE

SCALE

VOLTAGE

SCALE

VOLTAGE

GAIN

RANGE

(3)

RANGE

(3)

SETTING

(V)

0 to 5

1.25 to 3.75

2.5

1.88 to 3.13

2.5

1.25

2.19 to 2.81

1.25

0.625

2.34 to 2.66

2.5

0.625

NOTE: (1) With a 2.5V reference, such as the internal reference. (2) This
example utilizes the circuit in Figure 12. Other input ranges are possible. (3)
The ADS1210/11 allows common-mode voltage as long as the absolute
input voltage on A

P or A

N does not go below AGND or above AV

THEORY OF OPERATION

The ADS1210 and ADS1211 are precision, high dynamic
range, self-calibrating, 24-bit, delta-sigma A/D converters
capable of achieving very high resolution digital results.
Each contains a programmable gain amplifier (PGA); a
second-order delta-sigma modulator; a programmable digi-
tal filter; a microcontroller including the Instruction, Com-
mand and Calibration registers; a serial interface; a clock
generator circuit; and an internal 2.5V reference. The
ADS1211 includes a 4-channel input multiplexer.

In order to provide low system noise, common-mode rejec-
tion of 115dB and excellent power supply rejection, the
design topology is based on a fully differential switched
capacitor architecture. Turbo Mode, a unique feature of the
ADS1210/11, can be used to boost the sampling rate of the
input capacitor, which is normally 19.5kHz with a 10MHz
clock. By programming the Command Register, the sam-
pling rate can be increased to 39kHz, 78kHz, 156kHz, or
312kHz. Each increase in sample rate results in an increase
in performance when maintaining the same output data rate.

The programmable gain amplifier (PGA) of the ADS1210/
11 can be set to a gain of 1, 2, 4, 8 or 16--substantially
increasing the dynamic range of the converter and simplify-
ing the interface to the more common transducers (see Table
I). This gain is implemented by increasing the number of
samples taken by the input capacitor from 19.5kHz for a
gain of 1 to 312kHz for a gain of 16. Since the Turbo Mode
and PGA functions are both implemented by varying the
sampling frequency of the input capacitor, the combination
of PGA gain and Turbo Mode Rate is limited to 16 (see
Table II). For example, when using a Turbo Mode Rate of
8 (156kHz at 10MHz), the maximum PGA gain setting is 2.

TABLE I. Full-Scale Range vs PGA Setting.

TURBO MODE RATE

AVAILABLE PGA SETTINGS

1, 2, 4, 8, 16

1, 2, 4, 8

1, 2, 4

1, 2

TABLE II. Available PGA Settings vs Turbo Mode Rate.

The output data rate of the ADS1210/11 can be varied from
a few hertz to as much as 15,625kHz, trading off lower
resolution results for higher data rates. In addition, the data
rate determines the first null of the digital filter and sets the
3dB point of the input bandwidth (see the Digital Filter
section). Changing the data rate of the ADS1210/11 does not
result in a change in the sampling rate of the input capacitor.
The data rate effectively sets the number of samples which
are used by the digital filter to obtain each conversion result.
A lower data rate results in higher resolution, lower input
bandwidth, and different notch frequencies than a higher
data rate. It does not result in any change in input impedance
or modulator frequency, or any appreciable change in power
consumption.

The ADS1210/11 also includes complete on-board calibra-
tion that can correct for internal offset and gain errors or
limited external system errors. Internal calibration can be
run when needed, or automatically and continuously in the
background. System calibration can be run as needed and the
appropriate input voltages must be provided to the ADS1210/
11. For this reason, there is no continuous System Calibra-
tion Mode. The calibration registers are fully readable and
writable. This feature allows for switching between various
configurations--different data rates, Turbo Mode Rates, and
gain settings--without re-calibrating.

The various settings, rates, modes, and registers of the
ADS1210/11 are read or written via a synchronous serial
interface. This interface can operate in either a self-clocked
mode (Master Mode) or an externally clocked mode (Slave
Mode). In the Master Mode, the serial clock (SCLK) fre-
quency is one-half of the ADS1210/11 X

clock frequency.

This is an important consideration for many systems and
may determine the maximum ADS1210/11 clock that can be
used.

The high resolution and flexibility of the ADS1210/11 allow
these converters to fill a wide variety of A/D conversion
tasks. In order to ensure that a particular configuration will
meet the design goals, there are several important items
which must be considered. These include (but are certainly
not limited to) the needed resolution, required linearity,
desired input bandwidth, power consumption goal, and sen-
sor output voltage.

The remainder of this data sheet discusses the operation of
the ADS1210/11 in detail. In order to allow for easier
comparison of different configurations, "effective resolu-
tion" is used as the figure of merit for most tables and
graphs. For example, Table III shows a comparison between
data rate (and 3dB input bandwidth) versus PGA setting at
a Turbo Mode Rate of 1 and a clock rate of 10MHz. See the
Definition of Terms section for a definition of effective
resolution.

ADS1210, 1211

DATA

-3DB

RATE

FREQUENCY

(HZ)

G = 1

G = 2

G = 4

G = 8

G = 16

2.62

21.5

21.0

20.0

6.55

20.5

20.0

19.5

7.86

20.5

20.0

19.5

13.1

20.0

19.5

19.0

15.7

19.5

19.0

100

26.2

18.0

250

65.5

15.0

500

131

12.5

1000

262

10.0

10.5

10.0

EFFECTIVE RESOLUTION (BITS RMS)

TABLE III. Effective Resolution vs Data Rate and Gain

Setting. (Turbo Mode Rate of 1 and a 10MHz
clock.)

DEFINITION OF TERMS

An attempt has been made to be consistent with the termi-
nology used in this data sheet. In that regard, the definition
of each term is given as follows:

Analog Input Differential Voltage--For an analog signal
that is fully differential, the voltage range can be compared
to that of an instrumentation amplifier. For example, if both
analog inputs of the ADS1210 are at 2.5V, then the differ-
ential voltage is 0V. If one is at 0V and the other at 5V, then
the differential voltage magnitude is 5V. But, this is the case
regardless of which input is at 0V and which is at 5V, while
the digital output result is quite different.

The analog input differential voltage is given by the follow-
ing equation: A

P A

N. Thus, a positive digital output is

produced whenever the analog input differential voltage is
positive, while a negative digital output is produced when-
ever the differential is negative.

For example, when the converter is configured with a 2.5V
reference and placed in a gain setting of 2, the positive full-
scale output is produced when the analog input differential
is 2.5V. The negative full-scale output is produced when the
differential is 2.5V. In each case, the actual input voltages
must remain within the AGND to AV

range (see Table I).

Actual Analog Input Voltage--The voltage at any one
analog input relative to AGND.

Full-Scale Range (FSR)--As with most A/D converters,
the full-scale range of the ADS1210/11 is defined as the
"input" which produces the positive full-scale digital output
minus the "input" which produces the negative full-scale
digital output.

For example, when the converter is configured with a 2.5V
reference and is placed in a gain setting of 2, the full-scale
range is: [2.5V (positive full scale) minus 2.5V (negative
full scale)] = 5V.

Typical Analog Input Voltage Range--This term de-
scribes the actual voltage range of the analog inputs which
will cover the converter's full-scale range, assuming that
each input has a common-mode voltage that is greater than
REF

/PGA and smaller than (AV

REF

/PGA).

LSB Weight

Full

Scale Range

For example, when the converter is configured with a 2.5V
reference and placed in a gain setting of 2, the typical input
voltage range is 1.25V to 3.75V. However, an input range of
0V to 2.5V or 2.5V to 5V would also cover the converter's
full-scale range.

Voltage Span--This is simply the magnitude of the typical
analog input voltage range. For example, when the converter
is configured with a 2.5V reference and placed in a gain
setting of 2, the input voltage span is 2.5V.

Least Significant Bit (LSB) Weight--This is the theoreti-
cal amount of voltage that the differential voltage at the
analog input would have to change in order to observe a
change in the output data of one least significant bit. It is
computed as follows:

where N is the number of bits in the digital output.

Effective Resolution--The effective resolution of the
ADS1210/11 in a particular configuration can be expressed
in two different units: bits rms (referenced to output) and
microvolts rms (referenced to input). Computed directly
from the converter's output data, each is a statistical calcu-
lation based on a given number of results. Knowing one, the
other can be computed as follows:

The 10V figure in each calculation represents the full-scale
range of the ADS1210/11 in a gain setting of 1. This means
that both units are absolute expressions of resolution--the
performance in different configurations can be directly com-
pared regardless of the units. Comparing the resolution of
different gain settings expressed in bits rms requires ac-
counting for the PGA setting.

Main Controller--A generic term for the external
microcontroller, microprocessor, or digital signal processor
which is controlling the operation of the ADS1210/11 and
receiving the output data.

ER in bits rms

20 ·

log

PGA

ER in Vrms

- 1

ER in Vrms

PGA

02 ·

ER in bits rms

+ 1

ADS1210, 1211

DATA

XIN

Turbo Mode

512 ·

Decimation Ratio

+ 1

(

)

DATA

XIN

Turbo Mode

512 ·

Decimation Ratio

+ 1

(

)

DATA

FILTER RESPONSE

Frequency (Hz)

100

120

140

160

FILTER RESPONSE

Frequency (Hz)

100

120

140

160

100

150

200

250

300

Gain (dB)

NORMALIZED DIGITAL FILTER RESPONSE

Frequency (Hz)

100

120

140

160

Gain (dB)

FIGURE 3. Digital Filter Response at a Data Rate of 60Hz.

MOD

XIN

Turbo Mode

512

SAMP

XIN

Turbo Mode

Gain Setting

512

FIGURE 1. Normalized Digital Filter Response.

FIGURE 2. Digital Filter Response at a Data Rate of 50Hz.

If the effective resolution at a 50Hz or 60Hz data rate is not
adequate for the particular application, then power line fre-
quencies could still be rejected by operating the ADS1210/11
at 25/30Hz, 16.7/20Hz, 12.5/15Hz, etc. If a higher data rate
is needed, then power line frequencies must either be rejected
before conversion (with an analog notch filter) or after
conversion (with a digital notch filter running on the main
controller).

XIN

--The frequency of the crystal oscillator or CMOS

compatible input signal at the X

input of the ADS1210/11.

MOD

--The frequency or speed at which the modulator of the

ADS1210/11 is running, given by the following equation:

SAMP

--The frequency or switching speed of the input

sampling capacitor. The value is given by the following
equation:

DATA

, t

DATA

--The frequency of the digital output data

produced by the ADS1210/11 or the inverse of this (the
period), respectively, f

DATA

is also referred to as the data rate.

Conversion Cycle--The term "conversion cycle" usually
refers to a discrete A/D conversion operation, such as that
performed by a successive approximation converter. As
used here, a conversion cycle refers to the t

DATA

time period.

However, each digital output is actually based on the modu-
lator results from the last three t

DATA

time periods.

DIGITAL FILTER

The digital filter of the ADS1210/11 computes the output
result based on the most recent results from the delta-sigma
modulator. The number of modulator results that are used
depend on the decimation ratio set in the Command Regis-
ter. At the most basic level, the digital filter can be thought
of as simply averaging the modulator results and presenting
this average as the digital output.

While the decimation ratio determines the number of modu-
lator results to use, the modulator runs faster at higher Turbo
Modes. These two items, together with the ADS1210/11
clock frequency, determine the output data rate:

Also, since the conversion result is essentially an average,
the data rate determines where the resulting notches are in
the digital filter. For example, if the output data rate is 1kHz,
then a 1kHz input frequency will average to zero during the
1ms conversion cycle. Likewise, a 2kHz input frequency
will average to zero, etc.

In this manner, the data rate can be used to set specific notch
frequencies in the digital filter response (see Figure 1 for the
normalized response of the digital filter). For example, if the
rejection of power line frequencies is desired, then the data
rate can simply be set to the power line frequency. Figures
2 and 3 show the digital filter response for a data rate of
50Hz and 60Hz, respectively.

FILTER RESPONSE

Frequency (Hz)

100

120

140

160

FILTER RESPONSE

Frequency (Hz)

100

120

140

160

100

150

200

250

300

Gain (dB)

ADS1210, 1211

Filter Equation

The digital filter is described by the following transfer
function:

where N is the Decimation Ratio.

This filter has a (sin(x)/x)

response and is referred to a sinc

filter. For the ADS1210/11, this type of filter allows the data
rate to be changed over a very wide range (nearly four orders
of magnitude). However, the 3dB point of the filter is 0.262
times the data rate. And, as can be seen in Figures 1 and 2,
the rejection in the stopband (frequencies higher than the
first notch frequency) may only be 40dB.

These factors must be considered in the overall system
design. For example, with a 50Hz data rate, a significant
signal at 75Hz may alias back into the passband at 25Hz.
The analog front end can be designed to provide the needed
attenuation to prevent aliasing, or the system may simply
provide this inherently. Another possibility is increasing the
data rate and then post filtering with a digital filter on the
main controller.

Filter Settling

The number of modulator results used to compute each
conversion result is three times the Decimation Ratio. This
means that any step change (or any channel change for the
ADS1211) will require at least three conversions to fully
settle. However, if the change occurs asynchronously, then at
least four conversions are required to ensure complete set-
tling. For example, on the ADS1211, the fourth conversion
result after a channel change will be valid (see Figure 4).

| H (f ) |

sin

MOD

sin

MOD

the effective resolution of the output data at a given data rate,
but there is also an increase in power dissipation. For Turbo
Mode Rates 2 and 4, the increase is slight. For rates 8 and
16, the increase is more substantial. See the Typical Perfor-
mance Curves for more information.

In a Turbo Mode Rate of 16, the ADS1210/11 can offer 20
bits of effective resolution at a 1kHz data rate. A comparison
of effective resolution versus Turbo Mode Rates and output
data rates is shown in Table IV while Table V shows the
corresponding noise level in

Vrms.

TURBO MODE

The ADS1210/11 offers a unique Turbo Mode feature which
can be used to increase the modulator sampling rate by 2, 4,
8, or 16 times normal. With the increase of modulator
sampling frequency, there can be a substantial increase in

Data

Turbo

Rate

Mode

(Hz)

Rate 1

Rate 2

Rate 4

Rate 8

Rate 16

21.5

22.0

22.5

21.0

22.0

22.5

20.0

21.5

22.0

22.5

23.0

20.0

21.5

22.0

23.0

19.5

21.0

21.5

22.0

23.0

100

18.0

20.0

21.0

21.5

22.5

1000

10.0

12.5

15.0

17.5

20.0

Effective Resolution (Bits rms)

TABLE IV. Effective Resolution vs Data Rate and Turbo Mode

Rate. (Gain setting of 1 and 10MHz clock.)

DATA

TURBO

RATE

MODE

(Hz)

RATE 1

RATE 2

RATE 4

RATE 8

RATE 16

2.9

1.7

1.3

4.3

2.1

1.7

1.3

6.9

3.0

2.3

1.6

1.0

8.1

3.2

2.4

1.8

1.0

10.5

3.9

2.6

1.9

1.0

100

26.9

6.9

3.5

2.7

1.4

1000

6909.7

1354.5

238.4

46.6

7.8

TABLE V. Noise Level vs Data Rate and Turbo Mode Rate.

(Gain setting of 1 and 10MHz clock.)

NOISE LEVEL (

Vrms)

The Turbo Mode feature allows trade-offs to be made
between the ADS1210/11 X

clock frequency, power dissi-

pation, and effective resolution. If a 5MHz clock is available
but a 10MHz clock is needed to achieve the desired perfor-
mance, a Turbo Mode Rate of 2X will result in the same
effective resolution. Table VI provides a comparison of
effective resolution at various clock frequencies, data rates,
and Turbo Mode Rates.

DATA

CLOCK

TURBO

EFFECTIVE

RATE

FREQUENCY

MODE

RESOLUTION

(Hz)

(MHz)

RATE

(Bits rms)

19.5

2.5

19.5

1.25

19.5

0.625

19.5

100

18.0

100

18.0

100

2.5

18.0

100

1.25

18.0

100

0.625

18.0

TABLE VI. Effective Resolution vs Data Rate, Clock

Frequency, and Turbo Mode Rate. (Gain set-
ting of 1.)

FIGURE 4. Asynchronous ADS1210/11 Analog Input Volt-

age Step or ADS1211 Channel Change to Fully
Settled Output Data.

DRDY

Serial

I/O

Valid

Data

Valid

Data

Valid

Data

Valid

Data

not

Valid

Data

not

Valid

Data

not

Valid

Significant Analog Input Change

ADS1211 Channel Change

DATA

ADS1210, 1211

The Turbo Mode Rate (TMR) is programmed via the Sam-
pling Frequency bits of the Command Register. Due to the
increase in input capacitor sampling frequency, higher Turbo
Mode settings result in lower analog input impedance;

Impedance (

) = (10MHz/f

XIN

)·4.3E6/(G·TMR)

where G is the gain setting. Because the modulator rate also
changes in direct relation to the Turbo Mode setting, higher
values result in a lower impedance for the REF

input:

REF

Impedance (

) = (10MHz/f

XIN

)·1E6/TMR

The Turbo Mode Rate can be set to 1, 2, 4, 8, or 16. Consult
the graphs shown in the Typical Performance Curves for full
details on the performance of the ADS1210/11 operating in
different Turbo Mode Rates. Keep in mind that higher Turbo
Mode Rates result in fewer available gain settings as shown
in Table II.

PROGRAMMABLE GAIN AMPLIFIER

The programmable gain amplifier gain setting is programmed
via the PGA Gain bits of the Command Register. Changes
in the gain setting (G) of the programmable gain amplifier
results in an increase in the input capacitor sampling fre-
quency. Thus, higher gain settings result in a lower analog
input impedance:

Impedance (

) = (10MHz/f

XIN

)·4.3E6/(G·TMR)

where TMR is the Turbo Mode Rate. Because the modulator
speed does not depend on the gain setting, the input imped-
ance seen at REF

does not change.

The PGA can be set to gains of 1, 2, 4, 8, or 16. These gain
settings with their resulting full-scale range and typical
voltage range are shown in Table I. Keep in mind that higher
Turbo Mode Rates result in fewer available gain settings as
shown in Table II.

SOFTWARE GAIN

The excellent performance, flexibility, and low cost of the
ADS1210/11 allow the converter to be considered for de-
signs which would not normally need a 24-bit ADC. For
example, many designs utilize a 12-bit converter and a high-
gain INA or PGA for digitizing low amplitude signals. For
some of these cases, the ADS1210/11 by itself may be a
solution, even though the maximum gain is limited to 16.

To get around the gain limitation, the digital result can
simply be shifted up by "n" bits in the main controller--
resulting in a gain of "n" times G, where G is the gain
setting. While this type of manipulation of the output data
is obvious, it is easy to miss how much the gain can be
increased in this manner on a 24-bit converter.

For example, shifting the result up by three bits when the
ADS1210/11 is set to a gain of 16 results in an effective gain
of 128. At lower data rates, the converter can easily provide
more than 12 bits of resolution. Even higher gains are
possible. The limitation is a combination of the needed data
rate, desired noise performance, and desired linearity.

CALIBRATION

The ADS1210/11 offers several different types of calibra-
tion, and the particular calibration desired is programmed
via the Command Register. In the case of Background
Calibration, the calibration will repeat at regular intervals
indefinitely. For all others, the calibration is performed once
and then normal operation is resumed.

Each type of calibration is covered in detail in their respec-
tive section. In general, calibration is recommended imme-
diately after power-on and whenever there is a "significant"
change in the operating environment. The amount of change
which should cause a re-calibration is dependent on the
application, effective resolution, etc. Where high accuracy is
important, re-calibration should be done on changes in
temperature and power supply. In all cases, re-calibration
should be done when the gain, Turbo Mode, or data rate is
changed.

After a calibration has been accomplished, the Offset Cali-
bration Register and the Full-Scale Calibration Register
contain the results of the calibration. The data in these
registers are accurate to the effective resolution of the
ADS1210/11's mode of operation during the calibration.
Thus, these values will show a variation (or noise) equiva-
lent to a regular conversion result.

For those cases where this error must be reduced, it is
tempting to consider running the calibration at a slower data
rate and then increasing the converter's data rate after the
calibration is complete. Unfortunately, this will not work as
expected. The reason is that the results calculated at the
slower data rate would not be valid for the higher data rate.
Instead, the calibration should be done repeatedly. After
each calibration, the results can be read and stored. After the
desired number of calibrations, the main controller can
compute an average and write this value into the calibration
registers. The resulting error in the calibration values will be
reduced by the square root of the number of calibrations
which were averaged.

The calibration registers can also be used to provide system
offset and gain corrections separate from those computed by
the ADS1210/11. For example, these might be burned into
E

PROM during final product testing. On power-on, the

main controller would load these values into the calibration
registers. A further possibility is a look-up table based on the
current temperature.

Note that the values in the calibration registers will vary from
configuration to configuration and from part to part. There is
no method of reliably computing what a particular calibration
register should be to correct for a given amount of system
error. It is possible to present the ADS1210/11 with a known
amount of error, perform a calibration, read the desired
calibration register, change the error value, perform another
calibration, read the new value and use these values to
interpolate an intermediate value.

ADS1210, 1211

Valid

Data

DRDY

Serial

I/O

Valid

Data

SOC

(1)

DATA

Normal

Mode

Offset

Calibration on

System Offset

(2)

Analog

Input

Conversion

System Offset

Calibration Mode

Possibly

Valid

Data

Possibly

Valid

Data

Normal

Mode

NOTES: (1) SOC = System Offset Calibration instruction.
(2) In Slave Mode, this function requires 4 cycles.

Valid

Data

DRDY

Serial

I/O

Valid

Data

(1)

DATA

Normal

Mode

Valid

Data

Valid

Data

Normal

Mode

Offset

Calibration on

Internal Offset

(2)

Self-Calibration

Mode

Full-Scale

Calibration on

Internal Full-Scale

Analog

Input

Conversion

NOTES: (1) SC = Self-Calibration instruction. (2) In Slave Mode, this function requires 4 cycles.

FIGURE 5. Self-Calibration Timing.

Mode bits are reset to 000 (Normal Mode). A single conver-
sion is done with DRDY HIGH. After this conversion, the
DRDY signal goes LOW indicating resumption of normal
operation.

Normal operation returns within a single conversion cycle
because it is assumed that the input voltage at the converter's
input is not removed immediately after the offset calibration
is performed. In this case, the digital filter already contains
a valid result.

For full system calibration, offset calibration must be per-
formed first and then full-scale calibration. In addition, the
offset calibration error will be the rms sum of the conversion
error and the noise on the system offset voltage. See the
System Calibration Limits section for information regarding
the limits on the magnitude of the system offset voltage.

System Full-Scale Calibration

A system full-scale calibration is performed after the bits
011 have been written to the Command Register Operation
Mode bits (MD2 through MD0). This initiates the following
sequence (see Figure 7). At the start of the next conversion
cycle, the DRDY signal will not go LOW but will remain
HIGH and will continue to remain HIGH throughout the
calibration sequence. The full-scale calibration will be per-
formed on the differential input voltage (2 · REF

/G)

present at the converter's input over the next three conver-
sion periods (four in Slave Mode). When this is done, the
Operation Mode bits are reset to 000 (Normal Mode). A
single conversion is done with DRDY HIGH. After this
conversion, the DRDY signal goes LOW indicating resump-
tion of normal operation.

FIGURE 7. System Full-Scale Calibration Timing.

Self-Calibration

A self-calibration is performed after the bits 001 have been
written to the Command Register Operation Mode bits
(MD2 through MD0). This initiates the following sequence
at the start of the next conversion cycle (see Figure 5). The
DRDY signal will not go LOW but will remain HIGH and
will continue to remain HIGH throughout the calibration
sequence. The inputs to the sampling capacitor are discon-
nected from the converter's analog inputs and are shorted
together. An offset calibration is performed over the next
three conversion periods (four in Slave Mode). Then, the
input to the sampling capacitor is connected across REF

and a full-scale calibration is performed over the next three
conversions.

After this, the Operation Mode bits are reset to 000 (normal
mode) and the input capacitor is reconnected to the input.
Conversions proceed as usual over the next three cycles in
order to fill the digital filter. DRDY remains HIGH during
this time. On the start of the fourth cycle, DRDY goes LOW
indicating valid data and resumption of normal operation.

System Offset Calibration

A system offset calibration is performed after the bits 010
have been written to the Command Register Operation
Mode bits (MD2 through MD0). This initiates the following
sequence (see Figure 6). At the start of the next conversion
cycle, the DRDY signal will not go LOW but will remain
HIGH and will continue to remain HIGH throughout the
calibration sequence. The offset calibration will be per-
formed on the differential input voltage present at the
converter's input over the next three conversion periods
(four in Slave Mode). When this is done, the Operation

FIGURE 6. System Offset Calibration Timing.

Valid

Data

DRDY

Serial

I/O

Valid

Data

SFSC

(1)

DATA

Normal

Mode

Full-Scale

Calibration on

System Full-Scale

(2)

Analog

Input

Conversion

System Full-Scale

Calibration Mode

Possibly

Valid

Data

Possibly

Valid

Data

Normal

Mode

NOTES: (1) SFSC = System Full-Scale Calibration instruction.
(2) In Slave Mode, this function requires 4 cycles.

ADS1210, 1211

Normal

Mode

Background Calibration
Mode

Valid

Data

DRDY

Serial

I/O

Valid

Data

(1)

DATA

Offset

Calibration on

Internal Offset

(2)

Analog

Input

Conversion

Analog

Input

Conversion

Cycle Repeats

with Offset
Calibration

Full-Scale

Calibration on

Internal Full-Scale

NOTES: (1) BC = Background Calibration instruction. (2) In Slave Mode, the very first offset
calibration will require 4 cycles. All subsequent offset calibrations will require 3 cycles.

Normal operation returns within a single conversion cycle
because it is assumed that the input voltage at the converter's
input is not removed immediately after the full-scale calibra-
tion is performed. In this case, the digital filter already
contains a valid result.

For full system calibration, offset calibration must be per-
formed first and then full-scale calibration. The calibration
error will be a sum of the rms noise on the conversion result
and the input signal noise. See the System Calibration Limits
section for information regarding the limits on the magni-
tude of the system full-scale voltage.

Pseudo System Calibration

The Pseudo System Calibration is performed after the bits
100 have been written to the Command Register Operation
Mode bits (MD2 through MD0). This initiates the following
sequence (see Figure 8). At the start of the next conversion
cycle, the DRDY signal will not go LOW but will remain
HIGH and will continue to remain HIGH throughout the
calibration sequence. The offset calibration will be performed
on the differential input voltage present at the converter's
input over the next three conversion periods (four in Slave
Mode). Then, the input to the sampling capacitor is discon-
nected from the converter's analog input and connected
across REF

. A gain calibration is performed over the next

three conversions.

After this, the Operation Mode bits are reset to 000 (normal
mode) and the input capacitor is then reconnected to the

FIGURE 8. Pseudo System Calibration Timing.

FIGURE 9. Background Calibration Timing.

Valid

Data

DRDY

Serial

I/O

Valid

Data

PSC

(1)

DATA

Normal

Mode

Valid

Data

Valid

Data

Normal

Mode

Offset

Calibration on

System Offset

(2)

Pseudo System

Calibration Mode

Full-Scale

Calibration on

Internal Full-Scale

Analog

Input

Conversion

NOTES: (1) PSC = Pseudo System Calibration instruction. (2) In Slave Mode, this function requires 4 cycles.

input. Conversions proceed as usual over the next three
cycles in order to fill the digital filter. DRDY remains
HIGH during this time. On the next cycle, the DRDY signal
goes LOW indicating valid data and resumption of normal
operation.

The system offset calibration range of the ADS1210/11
is limited and is listed in the Specifications Table. For
more information on how to use these specifications, see
the System Calibration Limits section. To calculate V

use 2 · REF

/ GAIN for V

Background Calibration

The Background Calibration Mode is entered after the bits
101 have been written to the Command Register Operation
Mode bits (MD2 through MD0). This initiates the following
continuous sequence (see Figure 9). At the start of the next
conversion cycle, the DRDY signal will not go LOW but
will remain HIGH. The inputs to the sampling capacitor are
disconnected from the converter's analog input and shorted
together. An offset calibration is performed over the next
three conversion periods (in Slave Mode, the very first offset
calibration requires four periods and all subsequent offset
calibrations require three periods). Then, the input capacitor
is reconnected to the input. Conversions proceed as usual
over the next three cycles in order to fill the digital filter.
DRDY remains HIGH during this time. On the next cycle,
the DRDY signal goes LOW indicating valid data.

ADS1210, 1211

Also, during this cycle, the sampling capacitor is discon-
nected from the converter's analog input and is connected
across REF

. A gain calibration is initiated and proceeds

over the next three conversions. After this, the input capaci-
tor is once again connected to the analog input. Conversions
proceed as usual over the next three cycles in order to fill the
digital filter. DRDY remains HIGH during this time. On the
next cycle, the DRDY signal goes LOW indicating valid
data, the input to the sampling capacitor is shorted, and an
offset calibration is initiated. At this point, the Background
Calibration sequence repeats.

In essence, the Background Calibration Mode performs
continuous self-calibration where the offset and gain cali-
brations are interleaved with regular conversions. Thus, the
data rate is reduced by a factor of 6. The advantage is that
the converter is continuously adjusting to environmental
changes such as ambient or component temperature (due to
airflow variations).

The ADS1210/11 will remain in the Background Calibra-
tion Mode indefinitely. To move to any other mode, the
Command Register Operation Mode bits (MD2 through
MD0) must be set to the appropriate values.

System Calibration Offset and Full-Scale
Calibration Limits

The System Offset and Full-Scale Calibration range of the
ADS1210/11 is limited and is listed in the Specifications
Table. The range is specified as:

| V

|) < 1.3 · (2 · REF

)/GAIN

| V

|) > 0.7 · (2 · REF

)/GAIN

where V

is the system full-scale voltage and | V

| is the

absolute value of the system offset voltage. In the following
discussion, keep in mind that these voltages are differential
voltages.

For example, with the internal reference (2.5V) and a gain of
two, the previous equations become (after some manipulation):

3.25 < V

< V

1.75

If V

is perfect at 2.5V (positive full-scale), then V

must

be greater than 0.75V and less than 0.75V. Thus, when offset
calibration is performed, the positive input can be no more
than 0.75V below or above the negative input. If this range is
exceeded, the ADS1210/11 may not calibrate properly.

This calculation method works for all gains other than one.
For a gain of one and the internal reference (2.5V), the
equation becomes:

6.5 < V

< V

3.5

With a 5V positive full-scale input, V

must be greater than

1.5V and less than 1.5V. Since the offset represents a
common-mode voltage and the input voltage range in a gain
of one is 0V to 5V, a common-mode voltage will cause the
actual input voltage to possibly go below 0V or above 5V.
The specifications also show that for the specifications to be
valid, the input voltage must not go below AGND by more
than 30mV or above AV

by more than 30mV.

This will be an important consideration in many systems
which use a 2.5V or greater reference, as the input range is
constrained by the expected power supply variations. In
addition, the expected full-scale voltage will impact the
allowable offset voltage (and vice-versa) as the combination
of the two must remain within the power supply and ground
potentials, regardless of the results obtained via the range
calculation shown previously.

There are only two solutions to this constraint: either the
system design must ensure that the full-scale and offset
voltage variations will remain within the power supply and
ground potentials, or the part must be used in a gain of 2 or
greater.

SLEEP MODE

The Sleep Mode is entered after the bits 110 have been
written to the Command Register Operation Mode bits
(MD2 through MD0). This mode is exited by entering a new
mode into the MD2-MD0 bits.

The Sleep Mode causes the analog section and a good deal
of the digital section to power down. For full analog power
down, the V

BIAS

generator and the internal reference must

also be powered down by setting the BIAS and REFO bits
in the Command Register accordingly. The power dissipa-
tion shown in the Specifications Table is with the internal
reference and the V

BIAS

generator disabled.

To initiate serial communication with the converter while it
is in Sleep Mode, one of the following procedures must be
used: If CS is being used, simply taking CS LOW will
enable serial communication to proceed normally. If CS is
not being used (tied LOW) and the ADS1210/11 is in the
Master Mode, then a falling edge must be produced on the
SDIO line. If SDIO is LOW, the SDIO line must be taken
HIGH for 2 · t

XIN

periods (minimum) and then taken LOW.

Alternatively, SDIO can be forced HIGH after putting the
ADS1210/11 to "sleep" and then taken LOW when the
Sleep Mode is to be exited. Finally, if CS is not being used
(tied LOW) and the ADS1210/11 is in the Slave Mode, then
simply sending a normal Instruction Register command will
re-establish communication.

Once serial communication is resumed, the Sleep Mode is
exited by changing the MD2-MD0 bits to any other mode.
When a new mode (other than Sleep) has been entered, the
ADS1210/11 will execute a very brief internal power-up
sequence of the analog and digital circuitry. Once this has
been done, one normal conversion cycle is performed before
the new mode is actually entered. At the end of this conversion
cycle, the new mode takes effect and the converter will
respond accordingly. The DRDY signal will remain HIGH
through the first conversion cycle. It will also remain HIGH
through the second, even if the new mode is the Normal Mode.

If the V

BIAS

generator and/or the internal reference have

been disabled, then they must be manually re-enabled via the
appropriate bits in the Command Register. In addition, the
internal reference will have to charge the external bypass
capacitor(s) and possibly other circuitry. There may also be

ADS1210, 1211

considerations associated with V

BIAS

and the settling of

external circuitry. All of these must be taken into account
when determining the amount of time required to resume
normal operation. The timing diagram shown in Figure 10
does not take into account the settling of external circuitry.

FIGURE 10. Sleep Mode to Normal Mode Timing.

DRDY

Serial

I/O

NOTE: (1) Assuming that the external circuitry has
been stable for the previous three t

DATA

periods.

DATA

One

Normal

Conversion

(Other

Modes

Start Here)

Data

Not

Valid

Data

(1)

Valid

Data

(1)

Sleep Mode

Change to Normal Mode Occurs Here

ANALOG OPERATION

ANALOG INPUT

The input impedance of the analog input changes with
ADS1210/11 clock frequency (f

XIN

), gain (G), and Turbo

Mode Rate (TMR). The relationship is:

Impedance (

) = (10MHz/f

XIN

)·4.3E6/(G·TMR)

Figure 11 shows the basic input structure of the ADS1210.
The ADS1211 includes an input multiplexer, but this has
little impact on the analysis of the input structure. The
impedance is directly related to the sampling frequency of
the input capacitor. The X

clock rate sets the basic sam-

pling rate in a gain of 1 and Turbo Mode Rate of 1. Higher
gains and higher Turbo Mode Rates result in an increase of
the sampling rate, while slower clock (X

) frequencies

result in a decrease.

FIGURE 11. Analog Input Structure.

(8k

typical)

Switching Frequency

= f

SAMP

High

Impedance

> 1G

INT

8pF Typical

This input impedance can become a major point of consid-
eration in some designs. If the source impedance of the input
signal is significant or if there is passive filtering prior to the
ADS1210/11, then a significant portion of the signal can be
lost across this external impedance. How significant this
effect is depends on the desired system performance.

There are two restrictions on the analog input signal to the
ADS1210/11. Under no conditions should the current into
or out of the analog inputs exceed 10mA. In addition, while

the analog signal must reside within this range, the linearity
of the ADS1210/11 is only guaranteed when the actual
analog input voltage resides within a range defined by
AGND 30mV and AV

+30mV. This is due to leakage

paths which occur within the part when AGND and AV

are exceeded.

For this reason, the 0V to 5V input range (gain of 1 with a 2.5V
reference) must be used with caution. Should AV

be 4.75V,

the analog input signal would swing outside of the guaranteed
specifications of the device. Designs utilizing this mode of
operation should consider limiting the span to a slightly smaller
range. Common-mode voltages are also a significant concern
in this mode and must be carefully analyzed.

An input voltage range of 0.75V to 4.25V is the smallest
span that is allowed if a full system calibration will be
performed (see the Calibration section for more details).
This also assumes an offset error of zero. A better choice
would be 0.5V to 4.5V (a full-scale range of 9V). This span
would allow some offset error, gain error, power supply
drift, and common-mode voltage while still providing full
system calibration over reasonable variation in each of these
parameters.

The actual input voltage exceeding AGND or AV

should not

be a concern in higher gain settings as the input voltage range
will reside well within 0V to 5V. This is true unless the
common-mode voltage is large enough to place positive full-
scale or negative full-scale outside of the AGND to AV

range.

REFERENCE INPUT

The input impedance of the REF

input changes with clock

frequency (f

XIN

) and Turbo Mode Rate (TMR). The relationship

is:

REF

Impedance (

) = (10MHz/f

XIN

)·1E6/TMR

Unlike the analog input, the reference input impedance has
a negligible dependency on the PGA gain setting.

The reference input voltage can vary between 2V and 3V. A
nominal voltage of 2.5V appears at REF

OUT

, and this can be

directly connected to REF

. Higher reference voltages will

cause the full-scale range to increase while the internal
circuit noise of the converter remains approximately the
same. This will increase the LSB weight but not the internal
noise, resulting in increased signal-to-noise ratio and effec-
tive resolution. Likewise, lower reference voltages will de-
crease the signal-to-noise ratio and effective resolution.

REFERENCE OUTPUT

The ADS1210/11 contains an internal +2.5V reference.
Tolerances, drift, noise, and other specifications for this
reference are given in the Specification Table. Note that it is
not designed to sink or to source more than 1mA of current.
In addition, loading the reference with a dynamic or variable
load is not recommended. This can result in small changes
in reference voltage as the load changes. Finally, for designs
approaching or exceeding 20 bits of effective resolution, a
low-noise external reference is recommended as the internal
reference may not provide adequate performance.

ADS1210, 1211

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

XTAL

12pF

GND

DGND

12pF

±10V

AGND

1.0µF

FIGURE 12.

10V Input Configuration Using V

BIAS

The circuitry which generates the +2.5V reference can be
disabled via the Command Register and will result in a lower
power dissipation. The reference circuitry consumes a little over
1.6mA of current with no external load. When the ADS1210/11
is in its default state, the internal reference is enabled.

BIAS

The V

BIAS

output voltage is dependent on the reference input

(REF

) voltage and is approximately 1.33 times as great.

This output is used to bias input signals such that bipolar
signals with spans of greater than 5V can be scaled to match
the input range of the ADS1210/11. Figure 12 shows a
connection diagram which will allow the ADS1210/11 to
accept a

10V input signal (40V full-scale range).

This method of scaling and offsetting the

20V differential

input signal will be a concern for those requiring minimum
power dissipation. V

BIAS

will supply 1.68mA for every chan-

nel connected as shown. For the ADS1211, the current draw
is within the specifications for V

BIAS

, but, at 12mW, the

power dissipation is significant. If this is a concern, resistors
R

and R

can be set to 9k

and R

to 3k

. This will

reduce power dissipation by one-third. In addition, these
resistors can also be set to values which will provide any
arbitrary input range. In all cases, the maximum current into
or out of V

BIAS

should not exceed its specification of 10mA.

Note that the connection diagram shown in Figure 12 causes
a constant amount of current to be sourced by V

BIAS

. This

will be very important in higher resolution designs as the
voltage at V

BIAS

will not change with loading, as the load is

constant. However, if the input signal is single-ended and one
side of the input is grounded, the load will not be constant and
V

BIAS

will change slightly with the input signal. Also, in all

cases, note that noise on V

BIAS

introduces a common-mode

error signal which is rejected by the converter.

The 3k resistors should not be used as part of an anti-alias
filter with a capacitor across the inputs. The ADS1210
samples charge from the capacitor which has the effect of
introducing an offset in the measurement. This might be
acceptable for relative differential measurements.

The circuitry to generate V

BIAS

is disabled when the

ADS1210/11 is in its default state, and it must be enabled,
via the Command Register, in order for the V

BIAS

voltage to

be present. When enabled, the V

BIAS

circuitry consumes

approximately 1mA with no external load.

On power-up, external signals may be present before V

BIAS

is enabled. This can create a situation in which a negative
voltage is applied to the analog inputs (2.5V for the circuit
shown in Figure 12), reverse biasing the negative input
protection diode. This situation should not be a problem as
long as the resistors R

and R

limit the current being

sourced by each analog input to under 10mA (a potential of
0V at the analog input pin should be used in the calculation).

DIGITAL OPERATION

SYSTEM CONFIGURATION

The Micro Controller (MC) consists of an ALU and a
register bank. The MC has two states: power-on reset and
convert. In the power-on reset state, the MC resets all the
registers to their default state, sets up the modulator to a
stable state, and performs self-calibration at a 850Hz data
rate. After this, it enters the convert mode, which is the
normal mode of operation for the ADS1210/11.

The ADS1210/11 has 5 internal registers, as shown in Table
VII. Two of these, the Instruction Register and the Com-
mand Register, control the operation of the converter. The
Data Output Register (DOR) contains the result from the
most recent conversion. The Offset and Full-Scale Calibra-
tion Registers (OCR and FCR) contain data used for correct-
ing the internal conversion result before it is placed into the
DOR. The data in these two registers may be the result of a
calibration routine, or they may be values which have been
written directly via the serial interface.

TABLE VII. ADS1210/11 Registers.

INSR

Instruction Register

8 Bits

DOR

Data Output Register

24 Bits

CMR

Command Register

32 Bits

OCR

Offset Calibration Register

24 Bits

FCR

Full-Scale Calibration Register

24 Bits

Communication with the ADS1210/11 is controlled via the
Instruction Register (INSR). Under normal operation, the INSR
is written as the first part of each serial communication. The
instruction that is sent determines what type of communication
will occur next. It is not possible to read the INSR.

ADS1210, 1211

The Command Register (CMR) controls all of the ADS1210/
11's options and operating modes. These include the PGA
gain setting, the Turbo Mode Rate, the output data rate
(decimation ratio), etc. The CMR is the only 32-bit register
within the ADS1210/11. It, and all the remaining registers,
may be read from or written to.

Instruction Register (INSR)

The INSR is an 8-bit register which commands the serial
interface either to read or to write "n" bytes beginning at the
specified register location. Table VIII shows the format for
the INSR.

Each serial communication starts with the 8-bits of the INSR
being sent to the ADS1210/11. This directs the remainder of
the communication cycle, which consists of n bytes being
read from or written to the ADS1210/11. The read/write bit,
the number of bytes n, and the starting register address are
defined, as shown in Table VIII. When the n bytes have been
transferred, the INSR is complete. A new communication
cycle is initiated by sending a new INSR (under restrictions
outlined in the Interfacing section).

Command Register (CMR)

The CMR controls all of the functionality of the ADS1210/
11. The new configuration takes effect on the negative
transition of SCLK for the last bit in each byte of data being
written to the command register. The organization of the
CMR is shown in Table X.

The internal reference circuitry consumes approximately
1.6mA of steady state current with no external load. See the
Reference Output section for full details on the internal
reference.

TABLE VIII. Instruction Register.

R/W (Read/Write) Bit--For a write operation to occur, this
bit of the INSR must be 0. For a read, this bit must be 1, as
follows:

MB1, MB0 (Multiple Bytes) Bits--These two bits are used
to control the word length (number of bytes) of the read or
write operation, as follows:

A3-A0 (Address) Bits--These four bits select the begin-
ning register location which will be read from or written to,
as shown in Table IX. Each subsequent byte will be read
from or written to the next higher location. (If the BD bit in
the Command Register is set, each subsequent byte will be
read from the next lower location. This bit does not affect the
write operation.) If the next location is not defined in Table
IX, then the results are unknown. Reading or writing contin-
ues until the number of bytes specified by MB1 and MB0
have been transferred.

REGISTER BYTE

Data Output Register Byte 2 (MSB)

Data Output Register Byte 1

Data Output Register Byte 0 (LSB)

Command Register Byte 3 (MSB)

Command Register Byte 2

Command Register Byte 1

Command Register Byte 0 (LSB)

Offset Cal Register Byte 2 (MSB)

Offset Cal Register Byte 1

Offset Cal Register Byte 0 (LSB)

Full-Scale Cal Register Byte 2 (MSB)

Full-Scale Cal Register Byte 1

Full-Scale Cal Register Byte 0 (LSB)

Note: MSB = Most Significant Byte, LSB = Least Significant Byte

R/W

Write

Read

MB1

MB0

1 Byte

2 Bytes

3 Bytes

4 Bytes

TABLE IX. A3-A0 Addressing.

MSB

LSB

R/W

MB1

MB0

The V

BIAS

circuitry consumes approximately 1mA of steady

state current with no external load. See the V

BIAS

section for

full details. When the internal reference (REF

OUT

) is con-

nected to the reference input (REF

), V

BIAS

is 3.3V, nominal.

REFO (Reference Output) Bit--The REFO bit controls
the internal reference (REF

OUT

) state, either on (2.5V) or off

(disabled), as follows:

REFO

INTERNAL REFERENCE

REF

OUT

STATUS

Off

High Impedance

2.5V

Default

BIAS (Bias Voltage) Bit--The BIAS bit controls the V

BIAS

output state--either on (1.33 · REF

) or off (disabled), as

follows:

BIAS

GENERATOR

BIAS

STATUS

Off

Disabled

Default

1.33·REF

Most Significant Bit

Byte 3

DSYNC

(1)

BIAS REFO

U/B

MSB

SDL

DRDY

0 Off

1 On

0 Two's 0 Biplr 0 MSByte 0 MSB 0 SDIO

Defaults

NOTE: (1) DSYNC is Write only, DRDY is Read only.

Byte 2

MD2

MD1

MD0

CH1

CH0

000 Normal Mode

000 Gain 1

00 Channel 1

Defaults

Byte 1

SF2

SF1

SF0

DR12

DR11

DR10

DR9

DR8

000 Turbo Mode Rate of 1

00000

Defaults

Byte 0

Least Significant Bit

DR7

DR6

DR5

DR4

DR3

DR2

DR1

DR0

(00000) 0001 0111 (23) Data Rate of 814Hz

Defaults

TABLE X. Organization of the Command Register and

Default Status.

ADS1210, 1211

DF (Data Format) Bit--The DF bit controls the format of
the output data, either Two's Complement or Offset Binary,
as follows:

FORMAT

ANALOG INPUT

DIGITAL OUTPUT

Two's

+Full-Scale

7FFFFF

Default

Complement

Zero

000000

Full Scale

800000

Offset Binary

+Full-Scale

FFFFFF

Zero

800000

Full-scale

000000

These two formats are the same for all bits except the most
significant, which is simply inverted in one format vs the
other. This bit only applies to the Data Output Register--it
has no effect on the other registers.

U/B (Unipolar) Bit--The U/B bit controls the limits im-
posed on the output data, as follows:

U/B

MODE

LIMITS

Bipolar

None

Default

Unipolar

Zero to +Full-Scale only

The particular mode has no effect on the actual full-scale
range of the ADS1210/11, data format, or data format vs
input voltage. In the bipolar mode, the ADS1210/11 oper-
ates normally. In the unipolar mode, the conversion result is
limited to positive values only (zero included).

This bit only controls what is placed in the Data Output
Register. It has no effect on internal data. When cleared, the
very next conversion will produce a valid bipolar result.

BD (Byte Order) Bit--The BD bit controls the order in
which bytes of data are read, either most significant byte
first or least significant byte, as follows:

SDL (Serial Data Line) Bit--The SDL bit controls which
pin on the ADS1210/11 will be used as the serial data output
pin, either SDIO or SDOUT, as follows:

The MSB bit only affects read operations, it has no affect on
write operations.

SDL

SERIAL DATA OUTPUT PIN

SDIO

Default

SDOUT

If SDL is LOW, then SDIO will be used for both input and
output of serial data--see the Timing section for more
details on how the SDIO pin transitions between these two
states. In addition, SDOUT will remain in a tri-state condi-
tion at all times.

Important Note: Since the default condition is SDL LOW,
SDIO has the potential of becoming an output once every
data output cycle if the ADS1210/11 is in the Master Mode.
This will occur until the Command Register can be written
and the SDL bit set HIGH. See the Interfacing section for
more information.

DRDY (Data Ready) Bit--The DRDY bit is a read only bit
which reflects the state of the ADS1210/11's DRDY output
pin, as follows:

DRDY

MEANING

Data Ready

Data Not Ready

DSYNC (Data Synchronization) Bit--The DSYNC bit is
a write only bit which occupies the same location as DRDY.
When a `one' is written to this location, the affect on the
ADS1210/11 is the same as if the DSYNC input pin had
been taken LOW and returned HIGH. That is, the modulator
count for the current conversion cycle will be reset to zero.

The DSYNC bit is provided in order to reduce the number of
interface signals that are needed between the ADS1210/11
and the main controller. Consult "Making Use of DSYNC"
in the Serial Interface section for more information.

MD2-MD0 (Operating Mode) Bits--The MD2-MD0 bits
initiate or enable the various calibration sequences, as follows:

DSYNC

MEANING

No Change in Modulator Count

Modulator Count Reset to Zero

MD2

MD1

MD0

OPERATING MODE

Normal Mode

Self-Calibration

System Offset Calibration

System Full-Scale Calibration

Pseudo System Calibration

Background Calibration

Sleep

Reserved

The Normal Mode, Background Calibration Mode, and
Sleep Mode are permanent modes and the ADS1210/11 will
remain in these modes indefinitely. All other modes are
temporary and will revert to Normal Mode once the appro-
priate actions are complete. See the Calibration and Sleep
Mode sections for more information.

BYTE ACCESS ORDER

Most Significant

Default

to Least Significant Byte

Least Significant

to Most Significant Byte

Note that when BD is clear and a multi-byte read is initiated,
A3-A0 of the Instruction Register is the address of the most
significant byte and subsequent bytes reside at higher ad-
dresses. If BD is set, then A3-A0 is the address of the least
significant byte and subsequent bytes reside at lower ad-
dresses. The BD bit only affects read operations, it has no
affect on write operations.

MSB (Bit Order) Bit--The MSB bit controls the order in
which bits within a byte of data are read, either most
significant bit first or least significant bit, as follows:

MSB

BIT ORDER

Most Significant Bit First

Default

Least Significant Bit First

ADS1210, 1211

Data

Deci-

Rate

mation

(Hz)

Ratio

DR12

DR11

DR10

DR9

DR8

DR7

DR6

DR5

DR4

DR3

DR2

DR1

DR0

1000

500

250

100

194

325

390

976

1952

Table XI. Decimation Ratios vs Data Rates (Turbo Mode rate of 1 and 10MHz clock).

CH1

CH0

ACTIVE INPUT

Channel 1

Default

Channel 2

Channel 3

Channel 4

TURBO

AVAILABLE

MODE

PGA

SF2

SF1

SF0

RATE

SETTINGS

1, 2, 4, 8, 16

Default

1, 2, 4, 8

1, 2, 4

1, 2

Most Significant Bit

Byte 2

DOR23

DOR22

DOR21

DOR20

DOR19

DOR18

DOR17

DOR16

Byte 1

DOR15

DOR14

DOR13

DOR12

DOR11

DOR10

DOR9

DOR8

Byte 0

Least Significant Bit

DOR7

DOR6

DOR5

DOR4

DOR3

DOR2

DOR1

DOR0

TABLE XII. Data Output Register.

G2-G0 (PGA Control) Bits--The G2-G0 bits control the
gain setting of the PGA, as follows:

GAIN

AVAILABLE TURBO

SETTING

MODE RATES

1, 2, 4, 8, 16

Default

1, 2, 4, 8

1, 2, 4

1, 2

The gain is partially implemented by increasing the input
capacitor sampling frequency, which is given by the follow-
ing equation:

SAMP

= G · TMR · f

XIN

/ 512

where G is the gain setting and TMR is the Turbo Mode
Rate. The product of G and TMR cannot exceed 16. The
sampling frequency of the input capacitor directly relates to
the analog input impedance. See the Programmable Gain
Amplifier and Analog Input sections for more details.

CH1-CH0 (Channel Selection) Bits--The CH1 and CH0 bits
control the input multiplexer on the ADS1211, as follows:

(For the ADS1210, CH1 and CH0 must always be zero.) The
channel change takes effect when the last bit of byte 2 has
been written to the Command Register. Output data will not
be valid for the next three conversions despite the DRDY
signal indicating that data is ready. On the fourth time that
DRDY goes LOW after a channel change has been written
to the Command Register, valid data will be present in the
Data Output Register (see Figure 4).

SF2-SF0 (Turbo Mode Rate) Bits--The SF2-SF0 bits
control the input capacitor sampling frequency and modula-
tor rate, as follows:

The input capacitor sampling frequency and modulator rate
can be calculated from the following equations:

SAMP

= G · TMR · f

XIN

/ 512

MOD

= TMR · f

XIN

/ 512

where G is the gain setting and TMR is the Turbo Mode
Rate. The sampling frequency of the input capacitor directly
relates to the analog input impedance. The modulator rate
relates to the power consumption of the ADS1210/11 and
the output data rate. See the Turbo Mode, Analog Input, and
Reference Input sections for more details.

DR12-DR0 (Decimation Ratio) Bits--The DR12-DR0 bits
control the decimation ratio of the ADS1210/11. In essence,
these bits set the number of modulator results which are used in
the digital filter to compute each individual conversion result.
Since the modulator rate depends on both the ADS1210/11
clock frequency and the Turbo Mode Rate, the actual output
data rate is given by the following equation:

DATA

= f

XIN

· TMR / (512 · (Decimation Ratio + 1))

where TMR is the Turbo Mode Rate. Table XI shows
various data rates and corresponding decimation ratios (with
a 10MHz clock). Valid decimation ratios are from 19 to
8000. Outside of this range, the digital filter will compute
results incorrectly due to inadequate or too much data.

Data Output Register (DOR)

The DOR is a 24-bit register which contains the most recent
conversion result (see Table XII). This register is updated
with a new result just prior to DRDY going LOW. If the
contents of the DOR are not read within a period of time
defined by 1/f

DATA

12·(1/f

XIN

), then a new conversion

result will overwrite the old. (DRDY is forced HIGH prior
to the DOR update, unless a read is in progress).

The contents of the DOR can be in Two's Complement or
Offset Binary format. This is controlled by the DF bit of the
Command Register. In addition, the contents can be limited to
unipolar data only with the U/B bit of the Command Register.

ADS1210, 1211

Offset Calibration Register (OCR)

The OCR is a 24-bit register which contains the offset
correction factor that is applied to the conversion result before
it is placed in the Data Output Register (see Table XIII). In
most applications, the contents of this register will be the
result of either a self-calibration or a system calibration.

The OCR is both readable and writeable via the serial
interface. For applications requiring a more accurate offset
calibration, multiple calibrations can be performed, each
resulting OCR value read, the results averaged, and a more
precise offset calibration value written back to the OCR.

The actual OCR value will change from part-to-part and
with configuration, temperature, and power supply. Thus,
the actual OCR value for any arbitrary situation cannot be
accurately predicted. That is, a given system offset could not
be corrected simply by measuring the error externally, com-
puting a correction factor, and writing that value to the OCR.
In addition, be aware that the contents of the OCR are not
used to directly correct the conversion result. Rather, the
correction is a function of the OCR value. This function is
linear and two known points can be used as a basis for
interpolating intermediate values for the OCR. Consult the
Calibration section for more details.

Most Significant Bit

Byte 2

OCR23

OCR22

OCR21

OCR20

OCR19

OCR18

OCR17

OCR16

Byte 1

OCR15

OCR14

OCR13

OCR12

OCR11

OCR10

OCR9

OCR8

Byte 0

Least Significant Bit

OCR7

OCR6

OCR5

OCR4

OCR3

OCR2

OCR1

OCR0

TABLE XIII. Offset Calibration Register.

The contents of the OCR are in Two's Complement format.
This is not affected by the DF bit in the Command Register.

Full-Scale Calibration Register (FCR)

The FCR is a 24-bit register which contains the full-scale
correction factor that is applied to the conversion result before
it is placed in the Data Output Register (see Table XIV). In
most applications, the contents of this register will be the
result of either a self-calibration or a system calibration.

TABLE XIV. Full-Scale Calibration Register.

Most Significant Bit

Byte 2

FSR23

FSR22

FSR21

FSR20

FSR19

FSR18

FSR17

FSR16

Byte 1

FSR15

FSR14

FSR13

FSR12

FSR11

FSR10

FSR9

FSR8

Byte 0

Least Significant Bit

FSR7

FSR6

FSR5

FSR4

FSR3

FSR2

FSR1

FSR0

The FCR is both readable and writable via the serial inter-
face. For applications requiring a more accurate full-scale
calibration, multiple calibrations can be performed, each
resulting FCR value read, the results averaged, and a more
precise calibration value written back to the FCR.

The actual FCR value will change from part-to-part and with
configuration, temperature, and power supply. Thus, the
actual FCR value for any arbitrary situation cannot be
accurately predicted. That is, a given system full-scale error
cannot be corrected simply by measuring the error exter-
nally, computing a correction factor, and writing that value
to the FCR. In addition, be aware that the contents of the
FCR are not used to directly correct the conversion result.
Rather, the correction is a function of the FCR value. This
function is linear and two known points can be used as a
basis for interpolating intermediate values for the FCR.
Consult the Calibration section for more details. The con-
tents of the FCR are in unsigned binary format. This is not
affected by the DF bit in the Command Register.

TIMING

Table XV and Figures 13 through 21 define the basic digital
timing characteristics of the ADS1210/11. Figure 13 and the
associated timing symbols apply to the X

input signal.

Figures 14 through 20 and associated timing symbols apply
to the serial interface signals (SCLK, SDIO, SDOUT, and
CS) and their relationship to DRDY. The serial interface is
discussed in detail in the Serial Interface section. Figure 21
and the associated timing symbols apply to the maximum
DRDY rise and fall times.

FIGURE 13. X

Clock Timing.

XIN

FIGURE 14. Serial Input/Output Timing, Master Mode.

FIGURE 15. Serial Input/Output Timing, Slave Mode.

SCLK

(External)

SDIO

(as input)

SDOUT

(or SDlO

as output)

SCLK

(Internal)

SDIO

(as input)

SDOUT

(or SDlO

as output)

ADS1210, 1211

TABLE XV. Digital Timing Characteristics.

SYMBOL

DESCRIPTION

MIN

NOM

MAX

UNITS

XIN

Clock Frequency

0.5

MHz

XIN

Clock Period

100

2000

Clock High

0.4 · t

XIN

Clock LOW

0.4 · t

XIN

Internal Serial Clock HIGH

XIN

Internal Serial Clock LOW

XIN

Data In Valid to Internal SCLK Falling Edge (Setup)

Internal SCLK Falling Edge to Data In Not Valid (Hold)

Data Out Valid to Internal SCLK Falling Edge (Setup)

XIN

Internal SCLK Falling Edge to Data Out Not Valid (Hold)

XIN

External Serial Clock HIGH

2.5 · t

XIN

External Serial Clock LOW

2.5 · t

XIN

Data In Valid to External SCLK Falling Edge (Setup)

External SCLK Falling Edge to Data In Not Valid (Hold)

Data Out Valid to External SCLK Falling Edge (Setup)

XIN

External SCLK Falling Edge to Data Out Not Valid (Hold)

1.5 · t

XIN

Falling Edge of DRDY to First SCLK Rising Edge

6 · t

XIN

(Master Mode, CS Tied LOW)

Falling Edge of Last SCLK for INSR to Rising Edge of First

5 · t

XIN

SCLK for Register Data (Master Mode)

Falling Edge of Last SCLK for Register Data to Rising Edge

3 · t

XIN

of DRDY (Master Mode)

Falling Edge of Last SCLK for INSR to Rising Edge of First

5.5 · t

XIN

SCLK for Register Data (Slave Mode)

Falling Edge of Last SCLK for Register Data to Rising Edge

4 · t

XIN

5 · t

XIN

of DRDY (Slave Mode)

Falling Edge of DRDY to Falling Edge of CS (Master and

0.5 · t

XIN

Slave Mode)

Falling Edge of CS to Rising Edge of SCLK (Master Mode)

5 · t

XIN

6 · t

XIN

Rising Edge of DRDY to Rising Edge of CS (Master and

Slave Mode)

Falling Edge of CS to Rising Edge of SCLK (Slave Mode)

5.5 · t

XIN

Falling Edge of Last SCLK for INSR to SDIO Tri-state

2 · t

XIN

(Master Mode)

SDIO as Output to Rising Edge of First SCLK for Register

2 · t

XIN

Data (Master and Slave Modes)

Falling Edge of Last SCLK for INSR to SDIO Tri-state

3 · t

XIN

4 · t

XIN

(Slave Mode)

SDIO Tri-state Time (Master and Slave Modes)

XIN

Falling Edge of Last SCLK for Register Data to SDIO Tri-State

XIN

(Master Mode)

Falling Edge of Last SCLK for Register Data to SDIO

2 · t

XIN

3 · t

XIN

Tri-state (Slave Mode)

DRDY Fall Time

DRDY Rise Time

Minimum DSYNC LOW Time

10.5 · t

XIN

DSYNC Valid HIGH to Falling Edge of X

(for Exact

Synchronization of Multiple Converters only)

Falling Edge of X

to DSYNC Not Valid LOW (for Exact

Synchronization of Multiple Converters only)

Falling Edge of Last SCLK for Register Data to Rising Edge

20.5 · t

XIN

of First SCLK of next INSR (Slave Mode, CS Tied LOW)

Rising Edge of CS to Falling Edge of CS (Slave Mode,

10.5 · t

XIN

Using CS)

Falling Edge of DRDY to First SCLK

5.5 · t

XIN

Rising Edge (Slave Mode, CS Tied LOW)

ADS1210, 1211

Synchronizing Multiple Converters

A negative going pulse on DSYNC can be used to synchro-
nize multiple ADS1210/11s. This assumes that each
ADS1210 is driven from the same master clock and is set to
the same Decimation Ratio and Turbo Mode Rate. The
affect that this signal has on data output timing in general is
discussed in the Serial Interface section.

The concern here is what happens if the DSYNC input is
completely asynchronous to this master clock. If the DSYNC
input rises at a critical point in relation to the master clock
input, then some ADS1210/11s may start-up one X

clock

cycle before the others. Thus, the output data will be syn-
chronized, but only to within one X

clock cycle.

For many applications, this will be more than adequate. In
these cases, the timing symbols which relate the DSYNC
signal to the X

signal can be ignored. For other multiple-

converter applications, this one X

clock cycle difference

could be a problem. These types of applications would
include using the DRDY and/or the SCLK output from one
ADS1210/11 as the "master" signal for all converters.

To ensure exact synchronization to the same X

edge, the

timing relationship between the DSYNC and X

signals,

as shown in Figure 22, must be observed. Figure 23 shows
a simple circuit which can be used to clock multiple
ADS1210/11s from one ADS1210/11, as well as to ensure
that an asynchronous DSYNC signal will exactly synchro-
nize all the converters.

FIGURE 22. DSYNC to X

Timing for Synchronizing

Mutliple ADS1210/11s.

SERIAL INTERFACE

The ADS1210/11 includes a flexible serial interface which
can be connected to microcontrollers and digital signal
processors in a variety of ways. Along with this flexibility,
there is also a good deal of complexity. This section de-
scribes the trade-offs between the different types of interfac-
ing methods in a top-down approach--starting with the
overall flow and control of serial data, moving to specific
interface examples, and then providing information on vari-
ous issues related to the serial interface.

Multiple Instructions

The general timing diagrams which appear throughout this
data sheet show serial communication to and from the
ADS1210/11 occurring during the DRDY LOW period (see
Figures 4 through 10 and Figure 36). This communication
represents one instruction that is executed by the ADS1210/
11, resulting in a single read or write of register data.

However, more than one instruction can be executed by the
ADS1210/11 during any given conversion period (see Fig-
ure 24). Note that DRDY remains HIGH during the subse-
quent instructions. There are several important restrictions
on how and when multiple instructions can be issued during
any one conversion period.

DRDY

Serial

I/O

12 · t

XIN

Internal
Update of DOR

FIGURE 24. Timing of Data Output Register Update.

The first restriction is that the converter must be in the Slave
Mode. There is no provision for multiple instructions when
the ADS1210/11 is operating in the Master Mode. The
second is that some instructions will produce invalid results
if started at the end of one conversion period and carried into
the start of the next conversion period.

FIGURE 23. Exactly Synchronizing Multiple ADS1210/11s to an Asynchronous DSYNC Signal.

DSYNC

12pF

XTAL

DSYNC

OUT

DGND

ADS1210/11

SDOUT

SDIO

SCLK

CLK

1/2 74AHC74

1/6 74AHC04

DSYNC

OUT

DGND

ADS1210/11

SDOUT

SDIO

SCLK

DSYNC

OUT

DGND

ADS1210/11

SDOUT

SDIO

SCLK

Asynchronous

DSYNC

Strobe

DGND

ADS1210, 1211

Start

Writing

ADS1210/11

drives DRDY LOW

state

ADS1210/11

generates 8

serial clock cycles

and receives

Instruction Register

data via SDIO

ADS1210/11

generates n

serial clock cycles

and receives

specified

via SDIO

ADS1210/11

drives DRDY HIGH

End

ADS1210/11

generates 8 serial clock

cycles and receives
Instruction Register

data via SDIO

ADS1210/11 generates n

serial clock cycles

and transmits specified

End

Start

Reading

ADS1210/11

drives DRDY LOW

state

Continuous

Read

Mode?

ADS1210/11

drives DRDY HIGH

SDOUT returns to
tri-state condition

SDOUT becomes

active from tri-state

Use

SDIO for

output?

HIGH

Yes

LOW

HIGH

Yes

ADS1210/11 generates n

serial clock cycles

and transmits specified

SDIO transitions to

tri-state condition

SDIO input to

output transition

state

LOW

HIGH

For example, Figure 24 shows that just prior to the DRDY
signal going LOW, the internal Data Output Register (DOR)
is updated. This update involves the Offset Calibration
Register (OCR) and the Full-Scale Register (FSR). If the
OCR or FSR are being written, their final value may not be
correct, and the result placed into the DOR will certainly not
be valid. Problems can also arise if certain bits of the
Command Register are being changed.

Note that reading the Data Output Register is an excep-
tion. If the DOR is being read when the internal update is

initiated, the update is blocked. The old output data will
remain in the DOR and the new data will be lost. The old
data will remain valid until the read operation has com-
pleted. In general, multiple instructions may be issued, but
the last one in any conversion period should be complete
within 12 · X

clock periods of the next DRDY LOW

time. In this usage, "complete" refers to the point where
DRDY rises in Figures 17 and 19 (in the Timing Section).
Consult Figures 25 and 26 for the flow of serial data
during any one conversion period.

FIGURE 25. Flowchart for Writing and Reading Register Data, Master Mode.

ADS1210, 1211

Start

Writing

ADS1210/11

drives DRDY LOW

state

External device

generates 8

serial clock cycles

and transmits

instruction register

data via SDIO

External device

generates n

serial clock cycles

and transmits

specified

via SDIO

ADS1210/11

drives DRDY HIGH

End

External device generates

8 serial clock cycles

and receives

instruction register

data via SDIO

External device generates

n serial clock cycles

and transmits

specified register

data via SDOUT

End

Start

Reading

ADS1210/11

drives DRDY LOW

state

Continuous

Read

Mode?

ADS1210/11

drives DRDY HIGH

SDOUT returns to
tri-state condition

SDOUT becomes

active

Use

SDIO for

output?

LOW

HIGH

Yes

LOW

Yes

External device generates

n serial clock cycles

and receives

specified register

data via SDIO

SDIO transitions to

tri-state condition

SDIO input to

output transition

Instructions?

Yes

Is Next

Instruction

a Read?

To Read

Flowchart

From Read

Flowchart

CS taken HIGH

for 10.5 t

periods

minimum (see text

if CS tied LOW).

state

HIGH

Instructions?

Is Next

Instruction

a Write?

To Write

Flowchart

Yes

See text

for restrictions

Yes

To Write

Flowchart

CS taken HIGH

for 10.5 t

periods

minimum (see text

if CS tied LOW).

state

LOW

HIGH

See text

for restrictions

Yes

LOW

HIGH

FIGURE 26. Flowchart for Writing and Reading Register Data, Slave Mode.

XIN

transmits

ADS1210, 1211

The recommended solution to this problem is to actively pull
SDIO LOW. If SDIO is LOW when the ADS1210/11 enters
the instruction byte, then the resulting instruction is a write
of one byte of data to the Data Output Register, which results
in no internal operation.

If the SDIO signal cannot be actively pulled LOW, then
another possibility is to time the initialization of the
controller's serial port such that it becomes active between
adjacent DRDY LOW periods. The default configuration for
the ADS1210/11 produces a data rate of 814Hz--a conver-
sion period of 1.2ms. This time should be more than ad-
equate for most microcontrollers and DSPs to monitor DRDY
and initialize the serial port at the appropriate time.

Master Mode

The Master Mode is active when the MODE input is HIGH.
All serial clock cycles will be produced by the ADS1210/11
in this mode, and the SCLK pin is configured as an output.
The frequency of the serial clock will be one-half of the X

frequency. Multiple instructions cannot be issued during a
single conversion period in this mode--only one instruction
per conversion cycle is possible.

The Master Mode will be difficult for some microcontrollers,
particularly when the X

input frequency is greater than a

few MHz, as the serial clock may exceed the microcontroller's
maximum serial clock frequency. For the majority of digital
signal processors, this will be much less of a concern. In
addition, if SDIO is being used as an input and an output,
then the transition time from input to output may be a
concern. This will be true for both microcontrollers and
DSPs. See Figure 20 in the Timing section.

Note that if CS is tied LOW, there are special considerations
regarding SDIO as outlined previously in this section. Also
note that if CS is being used to control the flow of data from
the ADS1210/11 and it remains HIGH for one or more
conversion periods, the ADS1210/11 will operate properly.
However, the result in the Data Output Register will be lost
when it is overwritten by each new result. Just prior to this
update, DRDY will be forced HIGH and will return LOW
after the update.

Slave Mode

Most systems will use the ADS1210/11 in the Slave Mode.
This mode allows multiple instructions to be issued per
conversion period as well as allowing the main controller to
set the serial clock frequency and pace the serial data
transfer. The ADS1210/11 is in the Slave Mode when the
MODE input is LOW.

There are several important items regarding the serial clock
for this mode of operation. The maximum serial clock
frequency cannot exceed the ADS1210/11 X

frequency

divided by 5 (see Figure 15 in the Timing section).

When using SDIO as the serial output, the falling edge of the
last serial clock cycle of the instruction byte will cause the
SDIO pin to begin its transition from input to output.
Between three and four X

cycles after this falling edge, the

SDIO pin will become an output. This transition may be too
fast for some microcontrollers and digital signal processors.

Using CS and Continuous Read Mode

The serial interface may make use of the CS signal, or this
input may simply be tied LOW. There are several issues
associated with choosing to do one or the other.

The CS signal does not directly control the tri-state condition
of the SDOUT or SDIO output. These signals are normally
in the tri-state condition. They only become active when
serial data is being transmitted from the ADS1210/11. If the
ADS1210/11 is in the middle of a serial transfer and SDOUT
or SDIO is an output, taking CS HIGH will not tri-state the
output signal.

If there are multiple serial peripherals utilizing the same
serial I/O lines and communication may occur with any
peripheral at any time, then the CS signal must be used. The
ADS1210/11 may be in the Master Mode or the Slave Mode.
In the Master Mode, the CS signal is used to hold-off serial
communication with a "ready" (DRDY LOW) ADS1210/11
until the main controller can accommodate the communica-
tion. In the Slave Mode, the CS signal is used to enable
communication with the ADS1210/11.

The CS input has another use. If the CS state is left LOW
after a read of the Data Output Register has been performed,
then the next time that DRDY goes LOW, the ADS1210/11
Instruction Register will not be entered. Instead, the Instruc-
tion Register contents will be re-used, and the new contents
of the Data Output Register, or some part thereof, will be
transmitted. This will occur as long as CS is LOW and not
toggled.

This mode of operation is called the Continuous Read Mode
and is shown in the read flowcharts of Figures 25 and 26. It
is also shown in the Timing Diagrams of Figures 18 and 19
in the Timing section. Note that once CS has been taken
HIGH, the Continuous Read Mode will be enabled (but not
entered) and can never be disabled. The mode is actually
entered and exited as described above.

Power-On Conditions for SDIO

Even if the SDIO connection will be used only for input,
there is one important item to consider regarding SDIO. This
only applies when the ADS1210/11 is in the Master Mode
and CS will be tied LOW. At power-up, the serial I/O lines
of most microcontrollers and digital signal processors will be
in a tri-state condition, or they will be configured as inputs.
When power is applied to the ADS1210/11, it will begin
operating as defined by the default condition of the Com-
mand Register (see Table X in the System Configuration
section). This condition defines SDIO as the data output pin.

Since the ADS1210/11 is in the Master Mode and CS is tied
LOW, the serial clock will run whenever DRDY is LOW and
an instruction will be entered and executed. If the SDIO line
is HIGH, as it might be with an active pull-up, then the
instruction is a read operation and SDIO will become an
output every DRDY LOW period--for 32 serial clock cycles.
When the serial port on the main controller is enabled, signal
contention could result.

ADS1210, 1211

If a serial communication does not occur during any conver-
sion period, the ADS1210/11 will continue to operate prop-
erly. However, the results in the Data Output Register will
be lost when they are overwritten by the new result at the
start of the next conversion period. Just prior to this update,
DRDY will be forced HIGH and will return LOW after the
update.

Making Use of DSYNC

The DSYNC input pin and the DSYNC write bit in the
Command Register reset the current modulator count to
zero. This causes the current conversion cycle to proceed as
normal, but all modulator outputs from the last data output
to the point where DSYNC is asserted are discarded. Note
that the previous two data outputs are still present in the
ADS1210/11 internal memory. Both will be used to com-
pute the next conversion result, and the most recent one will
be used to compute the result two conversions later. DSYNC
does not reset the internal data to zero.

There are two main uses of DSYNC. In the first case,
DSYNC allows for synchronization of multiple converters.
In regards to the DSYNC input pin, this case was discussed
under "Synchronizing Multiple Converters" in the Timing
section. In regards to the DSYNC bit, it will be difficult to
set all of the converter's DSYNC bits at the same time
unless all of the converters are in the Slave Mode and the
same instruction can be sent to all of the converters at the
same time.

The second use of DSYNC is to reset the modulator count
to zero in order to obtain valid data as quickly as possible.
For example, if the input channel is changed on the ADS1211,
the current conversion cycle will be a mix of the old channel
and the new channels. Thus, four conversions are needed in
order to ensure valid data. However, if the channel is
changed and then DSYNC is used to reset the modulator
count, the modulator data at the end of the current conver-
sion cycle will be entirely from the new channel. After two
additional conversion cycles, the output data will be com-
pletely valid. Note that the conversion cycle in which
DSYNC is used will be slightly longer than normal. Its
length will depend on when DSYNC was set.

Reset, Power-On Reset, and Brown-Out

The ADS1210/11 contains an internal power-on reset circuit.
If the power supply ramp rate is greater than 50mV/ms, this
circuit will be adequate to ensure that the device powers up
correctly. (Due to oscillator settling considerations, commu-

nication to and from the ADS1210/11 should not occur for at
least 25ms after power is stable.)

If this requirement cannot be met or if the circuit has
brown-out considerations, the timing diagram of Figure 27
can be used to reset the ADS1210/11. This timing applies
only when the ADS1210/11 is in the Slave Mode and
accomplishes the reset by controlling the duty cycle of the
SCLK input. In general, a reset is required after power-up,
after a brown-out has been detected, or when a watchdog
timer event has occured.

If the ADS1210/11 is in the Master Mode, a reset of the
device is not possible. If the power supply does not meet the
minimum ramp rate requirement, or brown-out is of concern,
low on-resistance MOSFETs or equivalent should be used to
control power to the ADS1210/11. When powered down, the
device should be left unpowered for at least 300ms before
power is reapplied. An alternate method would be to control
the MODE pin and temporarily place the ADS1210/11 in the
Slave Mode while a reset is initiated as shown in Figure 27.

Two-Wire Interface

For a two-wire interface, the Master Mode of operation may
be preferable. In this mode, serial communication occurs
only when data is ready, informing the main controller as to
the status of the ADS1210/11. The disadvantages are that the
ADS1210/11 must have a dedicated serial port on the main
controller, only one instruction can be issued per data ready
period, and the serial clock may define the maximum clock
frequency of the converter.

In the Slave Mode, the main controller must read and write
to the ADS1210/11 "blindly". Writes to the internal regis-
ters, such as the Command Register or Offset Calibration
Register, might occur during an update of the Data Output
Register. This can result in invalid data in the DOR. A two-
wire interface can be used if the main controller can read
and/or write to the converter, either much slower or much
faster that the data rate. For example, if much faster, the
main controller can use the DRDY bit to determine when
data is becoming valid (polling it multiple times during one
conversion cycle). Thus, the controller obtains some idea of
when to write to the internal register. If much slower, then
reads of the DOR might always return valid data (multiple
conversions have occurred since the last read of the DOR or
since any write of the internal registers).

FIGURE 27. Resetting the ADS1210/11 (Slave Mode only).

: > 256 · t

XIN

< 400 · t

XIN

: > 5 · t

XIN

: > 512 · t

XIN

< 900 · t

XIN

1024 · t

XIN

< 1200 · t

XIN

SCLK

Reset Occurs

at Falling Edge

ADS1210, 1211

Four-wire Interface

Figure 30 shows a four-wire interface with a 8xC32 micro-
processor. Again, the Slave Mode is being used.

Multi-wire Interface

Figures 31 and 32 show multi-wire interfaces with a 8xC51
or 68HC11 microprocessor. In these interfaces, the mode of
the ADS1210/11 is actually controlled dynamically. This
could be extremely useful when the ADS1210/11 is to be
used in a wide variety of ways. For example, it might be
desirable to have the ADS1210/11 produce data at a steady
rate and to have the converter operating in the Continuous
Read Mode. But for system calibration, the Slave Mode
might be preferred because multiple instructions can be
issued per conversion period.

Note that the MODE input should not be changed in the
middle of a serial transfer. This could result in misoperation
of the device. A Master/Slave Mode change will not affect
the output data.

Note that the X

input can also be controlled. It is possible

with some microcontrollers and digital signal processors to
produce a continuous serial clock, which could be connected
to the X

input. The frequency of the clock is often settable

over some range. Thus, the power dissipation of the
ADS1210/11 could be dynamically varied by changing both
the Turbo Mode and X

input-trading off conversion speed

and resolution for power consumption.

I/O Recovery

If serial communication stops during an instruction or data
transfer for longer than 4 · t

DATA

, the ADS1210/11 will reset

its serial interface. This will not affect the internal registers.
The main controller must not continue the transfer after this
event, but must restart the transfer from the beginning.

This feature is very useful if the main controller can be reset
at any point. After reset, simply wait 8 · t

DATA

before

starting serial communication.

FIGURE 31. Full Interface with a 8xC51 Microprocessor.

FIGURE 30. Four-Wire Interface with a 8xC32 Microprocessor.

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

P1.0

P1.1

P1.2

P1.3

P1.4

P1.5

P1.6

P1.7

P0.0

P0.1

P0.2

P0.3

P0.4

P0.5

P0.6

8xC51

12pF

XTAL

AGND

DGND

AGND

1.0µF

10k

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

P1.0

P1.1

P1.2

P1.3

P1.4

P1.5

P1.6

P1.7

RESET

RXD

TXD

INT0

INT1

8xC32

27pF

AGND

DGND

CLK

1/2 74HC74

XTAL

AGND

1.0µF

10k

ADS1210, 1211

FIGURE 32. Full Interface with a 68HC11 Microprocessor.

Isolation

The serial interface of the ADS1210/11 provides for simple
isolation methods. An example of an isolated four-wire
interface is shown in Figure 33. The ISO150 is used to
transmit the digital signals over the isolation barrier.

In addition, the digital outputs of the ADS1210/11 can, in
some cases, drive opto-isolators directly. Figures 34 and 35
show the voltage of the SDOUT pin versus source or sink
current under worst case conditions. Worst-case conditions
for source current occur when the analog input differential

FIGURE 33. Isolated Four-Wire Interface.

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

PB7

PB6

PB5

PB4

PB3

PB2

PB1

PB0

PE0

PE1

PE2

XIRQ

RESET

PC7

PC6

PC5

PC4

PC3

PC2

PC1

PC0

XTAL

68HC11

12pF

AGND

DGND

XTAL

AGND

1.0µF

10k

ADS1210

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

XTAL

12pF

DD1

AGND

DGND

DD1

ISO150

R/T

DD2

DD1

DD2

100

DGND

GND

DGND

GND

SCLK

DD2

DD1

DD2

DGND

GND

OUT

DRDY

ISO150

R/T

AGND

1.0µF

ADS1210, 1211

ADS1211U, P

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

+5V

49.9k

12pF

DGND

XTAL

DGND

REF1004

+2.5V

+5V

1.0µF

+5V

ADS1211U, P

AGND

BIAS

DSYNC

OUT

DGND

REF

OUT

MODE

DRDY

SDOUT

SDIO

SCLK

1.0µF

+5V

49.9k

12pF

XTAL

DGND

REF1004

+2.5V

+5V

DRDY A

DRDY B

DRDY C

DSYNC

DATA

voltage is 5V and the output format is Offset Binary
(FFFFFF

). For sink current, the worst-case condition oc-

curs when the analog input differential voltage is 0V and the
output format is Two's Complement (000000

Note that SDOUT is tri-stated for the majority of the
conversion period and the opto-isolator connection must
take this into account.

Synchronization of Multiple Converters

The DSYNC input is used to synchronize the output data of
multiple ADS1210/11s. Synchronization involves configur-
ing each ADS1210/11 to the same Decimation Ratio and
Turbo Mode setting, and providing a common signal to the
X

inputs. Then, the DSYNC signal is pulsed LOW (see

Figure 22 in the Timing section). This results in an internal
reset of the modulator count for the current conversion.
Thus, all the converters start counting from zero at the same
time, producing a DRDY LOW signal at approximately the
same point (see Figure 36).

Note that an asynchronous DSYNC input may cause mul-
tiple converters to be different from one another by one X

clock cycle. This should not be a concern for most applica-
tions. However, the Timing section contains information on
exactly synchronizing multiple converters to the same X

clock cycle.

FIGURE 36. Affect of Synchronization on Output Data

Timing.

FIGURE 35. Sink Current vs V

for SDOUT Under Worst-Case Conditions.

SINK CURRENT

(V)

OUT

(mA)

25°C

85°C

40°C

FIGURE 34. Source Current vs V

for SDOUT Under Worst-Case Conditions.

SOURCE CURRENT

(V)

OUT

(mA)

25°C

85°C

40°C

ADS1210, 1211

LAYOUT

POWER SUPPLIES

The ADS1210/11 requires the digital supply (DV

) to be

no greater than the analog supply (AV

) +0.3V. In the

majority of systems, this means that the analog supply must
come up first, followed by the digital supply. Failure to
observe this condition could cause permanent damage to the
ADS1210/11.

Inputs to the ADS1210/11, such as SDIO, A

, or REF

should not be present before the analog and digital supplies
are on. Violating this condition could cause latch-up. If these
signals are present before the supplies are on, series resistors
should be used to limit the input current (see the Analog
Input and V

BIAS

sections of this data sheet for more details

concerning these inputs).

The best scheme is to power the analog section of the design
and AV

of the ADS1210/11 from one +5V supply and the

digital section (and DV

) from a separate +5V supply. The

analog supply should come up first. This will ensure that A

and REF

do not exceed AV

and that the digital inputs

are present only after AV

has been established, and that

they do not exceed DV

The analog supply should be well regulated and low noise.
For designs requiring very high resolution from the ADS1210/
11, power supply rejection will be a concern. See the PSRR
vs Frequency curve in the Typical Performance Curves
section of this data sheet for more information.

The requirements for the digital supply are not as strict.
However, high frequency noise on DV

can capacitively

couple into the analog portion of the ADS1210/11. This
noise can originate from switching power supplies, very fast
microprocessors or digital signal processors.

For either supply, high frequency noise will generally be
rejected by the digital filter except at interger multiplies of
f

MOD

. Just below and above these frequencies, noise will

alias back into the passband of the digital filter, affecting the
conversion result.

If one supply must be used to power the ADS1210/11, the
AV

supply should be used to power DV

. This connec-

tion can be made via a 10

resistor which, along with the

decoupling capacitors, will provide some filtering between
DV

and AV

. In some systems, a direct connection can

be made. Experimentation may be the best way to determine
the appropriate connection between AV

and DV

GROUNDING

The analog and digital sections of the design should be care-
fully and cleanly partitioned. Each section should have its own
ground plane with no overlap between them. AGND should be
connected to the analog ground plane as well as all other analog
grounds. DGND should be connected to the digital ground
plane and all digital signals referenced to this plane.

The ADS1210/11 pinout is such that the converter is cleanly
separated into an analog and digital portion. This should allow
simple layout of the analog and digital sections of the design.

For a single converter system, AGND and DGND of the
ADS1210/11 should be connected together, underneath the
converter. Do not join the ground planes, but connect the
two with a moderate signal trace. For multiple converters,
connect the two ground planes at one location as central to
all of the converters as possible. In some cases, experimen-
tation may be required to find the best point to connect the
two planes together. The printed circuit board can be de-
signed to provide different analog/digital ground connec-
tions via short jumpers. The initial prototype can be used to
establish which connection works best.

DECOUPLING

Good decoupling practices should be used for the ADS1210/
11 and for all components in the design. All decoupling
capacitors, but specifically the 0.1

F ceramic capacitors,

should be placed as close as possible to the pin being
decoupled. A 1

F to 10

F capacitor, in parallel with a 0.1

ceramic capacitor, should be used to decouple AV

AGND. At a minimum, a 0.1

F ceramic capacitor should be

used to decouple DV

to DGND, as well as for the digital

supply on each digital component.

SYSTEM CONSIDERATIONS

The recommendations for power supplies and grounding
will change depending on the requirements and specific
design of the overall system. Achieving 20 bits or more of
effective resolution is a great deal more difficult than achiev-
ing 12 bits. In general, a system can be broken up into four
different stages:

Analog Processing
Analog Portion of the ADS1210/11
Digital Portion of the ADS1210/11
Digital Processing

For the simplest system consisting of minimal analog signal
processing (basic filtering and gain), a self-contained
microcontroller, and one clock source, high-resolution could
be achieved by powering all components by a common
power supply. In addition, all components could share a
common ground plane. Thus, there would be no distinctions
between "analog" and "digital" power and ground. The
layout should still include a power plane, a ground plane,
and careful decoupling.

In a more extreme case, the design could include: multiple
ADS1210/11s; extensive analog signal processing; one or
more microcontrollers, digital signal processors, or micro-
processors; many different clock sources; and interconnec-
tions to various other systems. High resolution will be very
difficult to achieve for this design. The approach would be
to break the system into as many different parts as possible.
For example, each ADS1210/11 may have its own "analog"
processing front end, its own analog power and ground
(possibly shared with the analog front end), and its own
"digital" power and ground. The converter's "digital" power
and ground would be separate from the power and ground
for the system's processors, RAM, ROM, and "glue" logic.

ADS1210, 1211

TOPIC INDEX

TOPIC

PAGE

TOPIC

PAGE

ANALOG OPERATION .................................................................

ANALOG INPUT ....................................................................................... 17

REFERENCE INPUT ................................................................................ 17

REFERENCE OUTPUT ............................................................................ 17

BIAS .....................................................................................................................................................

DIGITAL OPERATION ..................................................................

SYSTEM CONFIGURATION .................................................................... 18

Instruction Register (INSR) ................................................................... 19

Command Register (CMR) .................................................................... 19

Data Output Register (DOR) ................................................................. 21

Offset Calibration Register (OCR) ........................................................ 22

Full-Scale Calibration Register (FCR) ................................................... 22

TIMING ..................................................................................................... 22

Synchronizing Multiple Converters ........................................................ 26

SERIAL INTERFACE ............................................................................... 26

Multiple Instructions ............................................................................... 26

Using CS and Continuous Read Mode ................................................ 29

Power-On Conditions for SDIO ............................................................. 29

Master Mode .......................................................................................... 29

Slave Mode ............................................................................................ 29

Making Use of DSYNC ......................................................................... 30

Reset, Power-On Reset, and Brown-Out ............................................. 30

Two-Wire Interface ................................................................................ 30

Three-Wire Interface .............................................................................. 30

Four-Wire Interface ................................................................................ 30

Multi-Wire Interface ............................................................................... 32

I/O Recovery .......................................................................................... 32

Isolation ................................................................................................. 33

Synchronization of Multiple Converters ................................................ 34

LAYOUT ........................................................................................

POWER SUPPLIES ................................................................................. 35

GROUNDING ............................................................................................ 35

DECOUPLING .......................................................................................... 35

SYSTEM CONSIDERATIONS ................................................................. 35

APPLICATIONS ............................................................................

FEATURES .....................................................................................

APPLICATIONS .............................................................................

DESCRIPTION ...............................................................................

SPECIFICATIONS ..........................................................................

ABSOLUTE MAXIMUM RATINGS ............................................................. 3

ELECTROSTATIC DISCHARGE SENSITIVITY ........................................ 3

PACKAGE INFORMATION ........................................................................ 3

ORDERING INFORMATION ...................................................................... 3

ADS1210 SIMPLIFIED BLOCK DIAGRAM ................................................ 4

ADS1210 PIN CONFIGURATION .............................................................. 4

ADS1210 PIN DEFINITIONS ..................................................................... 4

ADS1211 SIMPLIFIED BLOCK DIAGRAM ................................................ 5

ADS1211P and ADS1211U PIN CONFIGURATION ................................. 5

ADS1211P and ADS1211U PIN DEFINITIONS ........................................ 5

ADS1211E PIN CONFIGURATION ........................................................... 6

ADS1211E PIN DEFINITIONS ................................................................... 6

TYPICAL PERFORMANCE CURVES ...........................................

THEORY OF OPERATION ............................................................

DEFINITION OF TERMS ......................................................................... 10

DIGITAL FILTER ...................................................................................... 11

Filter Equation ....................................................................................... 12

Filter Settling .......................................................................................... 12

TURBO MODE ......................................................................................... 12

PROGRAMMABLE GAIN AMPLIFIER ..................................................... 13

SOFTWARE GAIN ................................................................................... 13

CALIBRATION .......................................................................................... 13

Self-Calibration ...................................................................................... 14

System Offset Calibration ..................................................................... 14

System Full-Scale Calibration ............................................................... 14

Pseudo System Calibration ................................................................... 15

Background Calibration ......................................................................... 15

System Calibration Offset and Full-Scale Calibration Limits ................ 16

SLEEP MODE .......................................................................................... 16

ADS1210, 1211

FIGURE INDEX

TABLE INDEX

FIGURE

TITLE

PAGE

Figure 1

Normalized Digital Filter Response ......................................... 11

Figure 2

Digital Filter Response at a Data Rate of 50Hz ..................... 11

Figure 3

Digital Filter Response at a Data Rate of 60Hz ..................... 11

Figure 4

Asynchronous ADS1210/11 Analog Input Voltage Step or

ADS1211 Channel Change to Fully Settled Output Data ...... 12

Figure 5

Self-Calibration Timing ............................................................ 14

Figure 6

System Offset Calibration Timing ........................................... 14

Figure 7

System Full-Scale Calibration ................................................. 14

Figure 8

Pseudo System Calibration Timing ......................................... 15

Figure 9

Background Calibration ........................................................... 15

Figure 10

Sleep Mode to Normal Mode Timing ...................................... 17

Figure 11

Analog Input Structure ............................................................. 17

Figure 12

10V Input Configuration Using V

BIAS ....................................................

Figure 13

Clock Timing ...................................................................... 22

Figure 14

Serial Input/Output Timing, Master Mode ............................... 22

Figure 15

Serial Input/Output Timing, Slave Mode ................................. 22

Figure 16

Serial Interface Timing (CS LOW), Master Mode ................... 24

Figure 17

Serial Interface Timing (CS LOW), Slave Mode ..................... 24

Figure 18

Serial Interface Timing (Using CS), Master Mode .................. 24

Figure 19

Serial Interface Timing (Using CS), Slave Mode .................... 25

Figure 20

SDIO Input to Output Transition Timing ................................. 25

Figure 21

DRDY Rise and Fall Time ....................................................... 25

Figure 22

DSYNC to X

Timing for Synchronizing Multiple

ADS1210/11s ........................................................................... 26

Figure 23

Exactly Synchronizing Multiple ADS1210/11s

to Asynchronous DSYNC Signal ............................................. 26

Figure 24

Timing of Data Output Register Update ................................. 26

Figure 25

Flowchart for Writing and Reading Register Data, Master Mode 27

Figure 26

Flowchart for Writing and Reading Register Data, Slave Mode .. 28

Figure 27

Resetting the ADS1210/11 (Slave Mode Only) ...................... 30

Figure 28

Three-Wire Interface with an 8xC32 Microprocessor ............. 31

Figure 29

Three-Wire Interface with an 8xC51 Microprocessor ............. 31

Figure 30

Four-Wire Interface with an 8xC32 Microprocessor ............... 32

Figure 31

Full Interface with an 8xC51 Microprocessor ......................... 32

Figure 32

Full Interface with a 68HC11 Microprocessor ........................ 33

Figure 33

Isolated Four-Wire Interface .................................................... 33

Figure 34

Source Current vs V

for SDOUT Under

Worst-Case Conditions ............................................................ 34

Figure 35

Sink Current vs V

for SDOUT Under

Worst-Case Conditions ............................................................ 34

Figure 36

Affect of Synchronization on Output Data Timing .................. 34

Figure 37

Bridge Transducer Interface with Voltage Excitation .............. 36

Figure 38

Bridge Transducer Interface with Current Excitation .............. 36

Figure 39

PT100 Interface ....................................................................... 37

Figure 40

Complete 4-20mA Receiver .................................................... 37

Figure 41

Single Supply, High Accuracy Thermocouple ......................... 37

Figure 42

Dual Supply, High Accuracy Thermocouple ........................... 38

Figure 43

Single Supply, High Accuracy Thermocouple Interface

with Cold Junction Compensation ........................................... 38

Figure 44

Dual Supply, High Accuracy Thermocouple Interface

with Cold Junction Compensation ........................................... 39

Figure 45

Low Cost Bridge Transducer Interface with Current Excitation ..... 39

TABLE

TITLE

PAGE

Table I

Full-Scale Range vs PGA Setting ............................................. 9

Table II

Available PGA Settings vs Turbo Mode Rate .......................... 9

Table III

Effective Resolution vs Data Rate and Gain Setting ............. 10

Table IV

Effective Resolution cs Data Rate and Turbo Mode Rate ..... 12

Table V

Noise Level vs Data Rate and Turbo Mode Rate .................. 12

Table VI

Effective Resolution vs Data Rate, Clock Frequency, and

Turbo Mode Rate .................................................................... 12

Table VII

ADS1210/11 Registers ............................................................ 18

Table VIII

Instruction Register .................................................................. 19

Table IX

A3-A0 Addressing .................................................................... 19

Table X

Organization of the Command Register and Default Status .. 19

Table XI

Decimation Ratios vs Data Rates ........................................... 21

Table XII

Data Output Register ............................................................... 21

Table XIII

Offset Calibration Register ...................................................... 22

Table XIV

Full-Scale Calibration Register ................................................ 22

Table XV

Digital Timing Characteristics .................................................. 23

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