LT1370

500kHz High Efficiency

6A Switching Regulator

The LT

1370 is a monolithic high frequency current mode

switching regulator. It can be operated in all standard
switching configurations including boost, buck, flyback,
forward, inverting and "Cuk." A 6A high efficiency switch
is included on the die, along with all oscillator, control and
protection circuitry.

The LT1370 typically consumes only 4.5mA quiescent
current and has higher efficiency than previous parts.
High frequency switching allows for very small inductors
to be used.

New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an exter-
nal logic level source. A logic low on the Shutdown pin
reduces supply current to 12

A. Unique error amplifier

circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation tech-
niques. Nonlinear error amplifier transconductance
reduces output overshoot on start-up or overload recov-
ery. Oscillator frequency shifting protects external com-
ponents during overload conditions.

DESCRIPTIO

Faster Switching with Increased Efficiency

Uses Small Inductors: 4.7

All Surface Mount Components

Low Minimum Supply Voltage: 2.7V

Quiescent Current: 4.5mA Typ

Current Limited Power Switch: 6A

Regulates Positive or Negative Outputs

Shutdown Supply Current: 12

A Typ

Easy External Synchronization

Switch Resistance: 0.065

Typ

TYPICAL APPLICATIO

FEATURES

, LTC and LT are registered trademarks of Linear Technology Corporation.

Boost Regulators

Laptop Computer Supplies

Multiple Output Flyback Supplies

Inverting Supplies

APPLICATIO

5V to 12V Boost Converter

LT1370

GND

LT1370 · TA01

S/S

L1*

C1**
22

25V

C4**
22

25V

0.047

C3
0.0047

R2
6.19k
1%

R1
53.6k
1%

OUT

12V

MBRD835L

OFF

COILTRONICS
UP2-4R7 (4.7

UP4-100 (10

AVX TPSD226M025R0200

4.7

OUT

1.8A

MAX I

OUT

12V Output Efficiency

LOAD CURRENT (A)

EFFICIENCY (%)

1.0 1.2 1.4 1.6 1.8

0.2 0.4 0.6 0.8

2.0

LT1370 · TA02

= 5V

L = 10

LT1370

BSOLUTE

Supply Voltage ....................................................... 30V
Switch Voltage

LT1370 ............................................................... 35V
LT1370HV .......................................................... 42V

S/S, SHDN, SYNC Pin Voltage ................................ 30V
Feedback Pin Voltage (Transient, 10ms) ..............

10V

Feedback Pin Current ........................................... 10mA
Negative Feedback Pin Voltage

(Transient, 10ms) .............................................

10V

Operating Ambient Temperature Range ...... 0

C to 70

Operating Junction Temperature Range

Commercial .......................................... 0

C to 125

Industrial ......................................... 40

C to 125

Storage Temperature Range ................ 65

C to 150

Lead Temperature (Soldering, 10 sec) ................. 300

ELECTRICAL C

HARA TERISTICS

= 5V, V

= 0.6V, V

= V

REF

, V

, S/S and NFB pins open, T

= 25

C unless otherwise noted.

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

REF

Reference Voltage

Measured at Feedback Pin

1.230

1.245

1.260

= 0.8V

1.225

1.245

1.265

Feedback Input Current

= V

REF

250

550

900

Reference Voltage Line Regulation

2.7V

25V, V

= 0.8V

0.01

0.03

%/V

NFR

Negative Feedback Reference Voltage

Measured at Negative Feedback Pin

2.525

2.48

2.435

Feedback Pin Open, V

= 0.8V

2.560

2.48

2.400

NFB

Negative Feedback Input Current

NFB

= V

NFR

Negative Feedback Reference Voltage

2.7V

25V, V

= 0.8V

0.01

0.05

%/V

Line Regulation

Error Amplifier Transconductance

1100

1500

1900

mho

700

2300

mho

Error Amplifier Source Current

= V

REF

150mV, V

= 1.5V

120

200

350

Error Amplifier Sink Current

= V

REF

+ 150mV, V

= 1.5V

1400

2400

PACKAGE/ORDER I FOR ATIO

JMAX

= 125

= 30

C/W,

= 4

C/W

R PACKAGE

7-LEAD PLASTIC DD

FRONT VIEW

TAB

GND

NFB
V

GND
S/S
FB
V

7
6
5
4
3
2
1

ORDER PART

NUMBER

WITH PACKAGE SOLDERED TO 0.5 INCH

COPPER

AREA OVER BACKSIDE GROUND PLANE OR INTERNAL
POWER PLANE.

CAN VARY FROM 20

C/W TO

> 40

C/W DEPENDING ON MOUNTING TECHNIQUE

ORDER PART

NUMBER

T7 PACKAGE

7-LEAD TO-220

NFB
V

GND
S/S
FB
V

FRONT VIEW

7
6
5
4
3
2
1

TAB

GND

JMAX

= 125

= 50

C/W,

= 4

C/W

Consult factory for Military grade parts.

LT1370CR
LT1370HVCR
LT1370IR
LT1370HVIR

LT1370CT7
LT1370HVCT7
LT1370IT7
LT1370HVIT7

LT1370

SYMBOL

PARAMETER

CONDITIONS

MIN

TYP

MAX

UNITS

Error Amplifier Clamp Voltage

High Clamp, V

= 1V

1.5

1.8

2.30

Low Clamp, V

= 1.5V

0.2

0.3

0.52

Error Amplifier Voltage Gain

500

V/ V

Pin Threshold

Duty Cycle = 0%

0.9

1.1

1.35

Switching Frequency

2.7V

25V

460

500

550

kHz

125

440

500

580

kHz

C (I-Grade)

400

580

kHz

Maximum Switch Duty Cycle

Switch Current Limit Blanking Time

130

300

Output Switch Breakdown Voltage

LT1370

LT1370HVC, 0

125

LT1370HVI, 40

C (I-Grade)

SAT

Output Switch ON Resistance

= 6A

0.065

0.11

LIM

Switch Current Limit

Duty Cycle = 50%

Duty Cycle = 80% (Note 1)

Supply Current Increase During Switch ON Time

mA/A

Control Voltage to Switch Current

A/V

Transconductance

Minimum Input Voltage

2.4

2.7

Supply Current

2.7V

25V

4.5

Shutdown Supply Current

2.7V

25V, V

S/S

0.6V

Shutdown Threshold

2.7V

25V

0.6

1.3

Shutdown Delay

S/S Input Current

S/S

Synchronization Frequency Range

600

800

kHz

ELECTRICAL C

HARA TERISTICS

= 5V, V

= 0.6V, V

= V

REF

, V

, S/S and NFB pins open, T

= 25

C unless otherwise noted.

The

denotes specifications which apply over the full operating

temperature range.

Note 1: For duty cycles (DC) between 45% and 85%, minimum switch
current limit is given by I

LIM

= 2.65(2.7 DC).

LT1370

TYPICAL PERFOR

CE CHARACTERISTICS

Switching Frequency
vs Feedback Pin Voltage

VOLTAGE (V)

INPUT CURRENT (

LT1370 · G07

FEEDBACK PIN VOLTAGE (V)

SWITCHING FREQUENCY (% OF TYPICAL)

110

0.8

LT1370 · G08

100

0.2

0.4

0.6

0.1

0.9

0.3

0.5

0.7

1.0

Switch Saturation Voltage
vs Switch Current

DUTY CYCLE (%)

6.6

SWITCH CURRENT LIMIT (A)

7.4

7.2

7.8

8.2

7.0

6.8

7.6

8.0

LT1370 · G02

100

Switch Current Limit
vs Duty Cycle

SWITCH CURRENT (A)

SWITCH VOLTAGE (mV)

300

400

550

LT1370 · G01

200

100

250

350

450

500

150

125

TEMPERATURE (

1.8

INPUT VOLTAGE (V)

2.0

2.2

2.4

2.6

100

150

LT1370 · G03

2.8

3.0

125

Minimum Input Voltage
vs Temperature

TEMPERATURE (

SHUTDOWN DELAY (

SHUTDOWN THRESHOLD (V)

LT1370 · G04

0.2

0.6

0.8

1.0

2.0

1.4

0.4

1.6

1.8

1.2

100 125

150

SHUTDOWN THRESHOLD

SHUTDOWN DELAY

Shutdown Delay and Threshold
vs Temperature

Error Amplifier Output Current
vs Feedback Pin Voltage

FEEDBACK PIN VOLTAGE (V)

400

ERROR AMPLIFIER OUTPUT CURRENT (

300

200

100

300

100

0.1

200

0.3

0.2

REF

125

LT1370 · G06

TEMPERATURE (

MINIMUM SYNCHRONIZATION VOLTAGE (V

P-P

)

0.5

1.0

1.5

2.0

100

150

LT1370 · G05

2.5

3.0

125

SYNC

= 700kHz

Minimum Synchronization
Voltage vs Temperature

Error Amplifier Transconductance
vs Temperature

TEMPERATURE (

TRANSCONDUCTANCE (

mho)

200

600

800

1000

2000

1400

LT1370 · G09

400

1600

1800

1200

100 125

150

I (V

)

V (FB)

S/S Pin Input Current vs Voltage

LT1370

TYPICAL PERFOR

CE CHARACTERISTICS

TEMPERATURE (

FEEDBACK INPUT CURRENT (nA)

400

500

600

150

LT1370 · G11

300

200

100

800

700

125

REF

Feedback Input Current
vs Temperature

TEMPERATURE (

NEGATIVE FEEDBACK INPUT CURRENT (

LT1370 · G12

100 125

150

NFB

NFR

Negative Feedback Input Current
vs Temperature

Pin Threshold and High

Clamp Voltage vs Temperature

TEMPERATURE (

1.0

VOLTAGE (V)

1.4

2.2

LT1370 · G10

1.2

1.8

2.0

1.6

100 125

150

HIGH CLAMP

THRESHOLD

CTIO

: The Compensation pin is used for frequency compen-

sation, current limiting and soft start. It is the output of the
error amplifier and the input of the current comparator.
Loop frequency compensation can be performed with an
RC network connected from the V

pin to ground. See

Applications Information.

FB: The Feedback pin is used for positive output voltage
sensing and oscillator frequency shifting. It is the invert-
ing input to the error amplifier. The noninverting input of
this amplifier is internally tied to a 1.245V reference.

NFB: The Negative Feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 100k
source resistor.

S/S: Shutdown and Synchronization Pin. The S/S pin is
logic level compatible. Shutdown is active low and the
shutdown threshold is typically 1.3V. For normal opera-
tion, pull the S/S pin high, tie it to V

or leave it floating. To

synchronize switching, drive the S/S pin between 600kHz
and 800kHz. See Applications Information.

: Bypass Input Supply Pin with a Low ESR Capacitor,

F or More. The regulator goes into undervoltage lock-

out when V

drops below 2.5V. Undervoltage lockout

stops switching and pulls the V

pin low.

: The Switch pin is the collector of the power switch

and has large currents flowing through it. Keep the traces
to the switching components as short as possible to
minimize radiation and voltage spikes.

GND: Tie all ground pins to a good quality ground plane.
See Applications Information.

LT1370

BLOCK DIAGRA

OPERATIO

The LT1370 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage. Referring to the block
diagram, the switch is turned ON at the start of each
oscillator cycle. It is turned OFF when switch current
reaches a predetermined level. Control of output voltage is
obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90

phase shift at midfrequencies

in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely vary-
ing input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator pro-
vides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A

500kHz oscillator is the basic clock for all internal timing.
It turns on the output switch via the logic and driver
circuitry. Special adaptive antisat circuitry detects onset of
saturation in the power switch and adjusts driver current
instantaneously to limit switch saturation. This minimizes
driver dissipation and provides very rapid turn-off of the
switch.

A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce out-
put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases 10 times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps pro-
tect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator fre-
quency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.

NFBA

NFB

S/S

100k

50k

0.005

GND

LT1370 · BD

GND SENSE

1.245V

REF

5:1 FREQUENCY

SHIFT

OSC

SYNC

SHUTDOWN

DELAY AND RESET

LOW DROPOUT

2.3V REG

ANTI-SAT

LOGIC

DRIVER

SWITCH

COMP

LT1370

Unique error amplifier circuitry allows the LT1370 to
directly regulate negative output voltages. The negative
feedback amplifier's 100k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at 2.48V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. Consult LTC Marketing for units
that can regulate down to 1.25V.

The error signal developed at the amplifier output is
brought out externally. This pin (V

) has three different

functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regula-
tor operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (g

) type, so this voltage can

be externally clamped for lowering current limit. Like-
wise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the V

pin is pulled

below the control pin threshold, placing the LT1370 in an
idle mode.

Positive Output Voltage Setting

The LT1370 develops a 1.245V reference (V

REF

) from the

FB pin to ground. Output voltage is set by connecting the
FB pin to an output resistor divider (Figure 1). The FB pin
bias current represents a small error and can usually be
ignored for values of R2 up to 7k. The suggested value for
R2 is 6.19k. The NFB pin is normally left open for positive
output applications. Positive fixed voltage versions are
available (consult LTC Marketing).

Negative Output Voltage Setting

The LT1370 develops a 2.48V reference (V

NFR

) from the

NFB pin to ground. Output voltage is set by connecting the
NFB pin to an output resistor divider (Figure 2). The
30

A NFB pin bias current (I

NFB

) can cause output

voltage errors and should not be ignored. This has been
accounted for in the formula in Figure 2. The suggested
value for R2 is 2.49k. The FB pin is normally left open for
negative output applications.

Dual Polarity Output Voltage Sensing

Certain applications benefit from sensing both positive
and negative output voltages. One example is the "Dual
Output Flyback Converter with Overvoltage Protection"
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as
described above. When both the FB and NFB pins are used,

OUT

= V

REF

1 +

REF

OUT

( )

R1
R2

R1 = R2

(

)

OUT

1.245

LT1370 · F01

APPLICATIO S I FOR ATIO

Figure 1. Positive Output Resistor Divider

the LT1370 acts to prevent either output from going
beyond its set output voltage. For example, in this applica-
tion if the positive output were more heavily loaded than
the negative, the negative output would be greater and
would regulate at the desired set-point voltage. The posi-
tive output would sag slightly below its set-point voltage.
This technique prevents either output from going unregu-
lated high at no load.

Figure 2. Negative Output Resistor Divider

OUT

= V

NFB

+ I

NFB

(R1)

1 +

LT1370 · F02

NFB

NFR

NFB

OUT

( )

R1
R2

R1 =

+ 30 · 10

OUT

2.48

( ) ( )

2.48

OPERATIO

LT1370

Shutdown and Synchronization

The device has a dual function S/S pin which is used for
both shutdown and synchronization. This pin is logic level
compatible and can be pulled high, tied to V

or left

floating for normal operation. A logic low on the S/S pin
activates shutdown, reducing the part's supply current to
12

A. Typical synchronization range is from 1.05 to 1.8

times the part's natural switching frequency, but is only
guaranteed between 600kHz and 800kHz. A 12

s resetable

shutdown delay network guarantees the part will not go
into shutdown while receiving a synchronization signal.

Caution should be used when synchronizing above 700kHz
because at higher sync frequencies the amplitude of the
internal slope compensation used to prevent subhar-
monic switching is reduced. This type of subharmonic
switching only occurs when the duty cycle of the switch
is above 50%. Higher inductor values will tend to elimi-
nate this problem.

Thermal Considerations

Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces-
sive die temperatures. Typical thermal resistance is
30

C/W for the R package and 50

C/W for the T7 package

but these numbers will vary depending on the mounting
techniques (copper area, airflow, etc.). Heat is transferred
from the package via the tab.

Average supply current (including driver current) is:

= 4.5mA + DC(I

/45)

= Switch current

DC = Switch duty cycle

Switch power dissipation is given by:

= (I

)

)(DC)

= Output switch ON resistance

Total power dissipation of the die is the sum of supply
current times supply voltage, plus switch power:

D(TOTAL)

= (I

)(V

) + P

Surface mount heat sinks are available which can lower
package thermal resistance by two or three times. One
manufacturer, Wakefield Engineering, offers surface mount
heat sinks for the R package and can be reached at (617)
245-5900 or at www.wakefield.com.

Choosing the Inductor

For most applications the inductor will fall in the range of
2.2

H to 22

H. Lower values are chosen to reduce physi-

cal size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch, which has a 6A limit. Higher values also
reduce input ripple voltage and reduce core loss.

When choosing an inductor you need to consider maxi-
mum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault
current in the inductor, saturation and, of course, cost.
The following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.

1. Assume that the average inductor current for a boost

converter is equal to load current times V

OUT

/ V

and

decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 3A, for instance, a
3A inductor may not survive a continuous 6A overload
condition. Also be aware that boost converters are not
short-circuit protected and that, under output short
conditions, inductor current is limited only by the
available current of the input supply.

2. Calculate peak inductor current at full load current to

ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe-
cially with smaller inductors and lighter loads, so don't
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
saturate abruptly and other core materials fall in
between. The following formula assumes continuous
mode operation but it errs only slightly on the high side
for discontinuous mode, so it can be used for all
conditions.

APPLICATIO S I FOR ATIO

LT1370

APPLICATIO S I FOR ATIO

tested for low ESR, so they give the lowest ESR for a given
volume. To further reduce ESR, multiple output capaci-
tors can be used in parallel. The value in microfarads is
not particularly critical, and values from 22

F to greater

than 500

F work well, but you cannot cheat mother

nature on ESR. If you find a tiny 22

F solid tantalum

capacitor, it will have high ESR and output ripple voltage
will be terrible. Table 1 shows some typical solid tantalum
surface mount capacitors.

Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current

E CASE SIZE

ESR (MAX

)

RIPPLE CURRENT (A)

AVX TPS, Sprague 593D

0.1 to 0.3

0.7 to 1.1

AVX TAJ

0.7 to 0.9

0.4

D CASE SIZE

AVX TPS, Sprague 593D

0.1 to 0.3

0.7 to 1.1

AVX TAJ

0.9 to 2.0

0.36 to 0.24

C CASE SIZE

AVX TPS

0.2 (Typ)

0.5 (Typ)

AVX TAJ

1.8 to 3.0

0.22 to 0.17

B CASE SIZE

AVX TAJ

2.5 to 10

0.16 to 0.08

Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and AVX type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the

output capacitor. Solid

tantalum capacitors fail during very high

turn-on surges,

which do not occur at the output of regulators. High
discharge surges, such as when the regulator output is
dead-shorted, do not harm the capacitors.

Single inductor boost regulators have large RMS ripple
current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:

Output Capacitor Ripple Current (RMS)

RIPPLE

(RMS) = I

OUT

= I

OUT

1 DC

DC = Switch duty cycle

PEAK

= (I

OUT

)

= Minimum input voltage

f = 500kHz switching frequency

OUT

)

2(f)(L)(V

OUT

)

3. Decide if the design can tolerate an "open" core geom-

etry, like a rod or barrel, which has high magnetic field
radiation, or whether it needs a closed core, like a
toroid, to prevent EMI problems. One would not want an
open core next to a magnetic storage media, for
instance! This is a tough decision because the rods or
barrels are temptingly cheap and small and there are no
helpful guidelines to calculate when the magnetic field
radiation will be a problem.

4. Start shopping for an inductor that meets the

requirements of core shape, peak current (to avoid
saturation), average current (to limit heating) and fault
current. If the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts. Keep in mind that
all good things like high efficiency, low profile and high
temperature operation will increase cost, sometimes
dramatically.

5. After making an initial choice, consider the secondary

things like output voltage ripple, second sourcing, etc.
Use the experts in the LTC Applications Department if
you feel uncertain about the final choice. They have
experience with a wide range of inductor types and can
tell you about the latest developments in low profile,
surface mounting, etc.

Output Capacitor

The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. At 500kHz any polarized capacitor
is essentially resistive. To get low ESR takes

volume, so

physically smaller capacitors have high ESR. The ESR
range needed for typical LT1370 applications is 0.025

to 0.2

. A typical output capacitor is an AVX type TPS,

F at 25V (two each), with a guaranteed ESR less than

0.2

. This is a "D" size surface mount solid tantalum

capacitor. TPS capacitors are specially constructed and

LT1370

APPLICATIO S I FOR ATIO

Output Diode

The suggested output diode (D1) is a Motorola MBRD835L.
It is rated at 8A average forward current and 35V reverse
voltage. Typical forward voltage is 0.4V at 3A. The diode
conducts current only during switch OFF time. Peak re-
verse voltage for boost converters is equal to regulator
output voltage. Average forward current in normal opera-
tion is equal to output current.

Frequency Compensation

Loop frequency compensation is performed on the output
of the error amplifier (V

pin) with a series RC network.

The main pole is formed by the series capacitor and the
output impedance (

500k

) of the error amplifier. The

pole falls in the range of 2Hz to 20Hz. The series resistor
creates a "zero" at 1kHz to 5kHz, which improves loop
stability and transient response. A second capacitor, typi-
cally one-tenth the size of the main compensation capaci-
tor, is sometimes used to reduce the switching frequency
ripple on the V

pin. V

pin ripple is caused by output

voltage ripple attenuated by the output divider and multi-
plied by the error amplifier. Without the second capacitor,
V

pin ripple is:

Pin Ripple =

RIPPLE

= Output ripple (V

)

= Error amplifier transconductance

(

1500

mho)

= Series resistor on V

OUT

= DC output voltage

1.245(V

RIPPLE

)(g

)(R

)

OUT

)

To prevent irregular switching, V

pin ripple should be

kept below 50mV

Worst-case V

pin ripple occurs at

maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.0047

F capacitor on the V

pin reduces

switching frequency ripple to only a few millivolts. A low
value for R

will also reduce V

pin ripple, but loop phase

margin may be inadequate.

Input Capacitors

The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular and
does not contain large squarewave currents as is found in
the output capacitor. Capacitors in the range of 10

F to

100

F with an ESR of 0.1

or less work well up to full 6A

switch current. Higher ESR capacitors may be acceptable
at low switch currents. Input capacitor ripple current for a
boost converter is :

RIPPLE

f = 500kHz switching frequency

0.3(V

)(V

OUT

)

(f)(L)(V

OUT

)

The input capacitor can see a very high surge current when
a battery or high capacitance source is connected "live"
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed tantalum capaci-
tors specially tested for surge capability (AVX TPS series,
for instance) but even these units may fail if the input
voltage approaches the maximum voltage rating of the
capacitor during a high surge. AVX recommends derating
capacitor voltage by 2:1 for high surge applications.
Ceramic, OS-CON and aluminum electrolytic capacitors
may also be used and have a high tolerance to turn-on
surges.

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempt-
ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop "zero" at 5kHz to 50kHz that is instru-
mental in giving acceptable loop phase margin. Ceramic
capacitors remain capacitive to beyond 300kHz and usu-
ally resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.

LT1370

APPLICATIO S I FOR ATIO

Layout Considerations

For maximum efficiency, LT1370 switch rise and fall times
are made as short as possible. To prevent radiation and
high frequency resonance problems, proper layout of the
components connected to the switch node is essential. B
field (magnetic) radiation is minimized by keeping output
diode, switch pin and output bypass capacitor leads as
short as possible. Figure 3 shows recommended posi-
tions for these components. E field radiation is kept low by
minimizing the length and area of all traces connected to
the switch pin. A ground plane should always be used
under the switcher circuitry to prevent interplane
coupling.

The high speed switching current path is shown schemati-
cally in Figure 4. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.

More Help

For more detailed information on switching regulator
circuits, please see Application Note 19. Linear
Technology also offers a computer software program,
SwitcherCAD

, to assist in designing switching convert-

ers. In addition, our Applications Department is always
ready to lend a helping hand.

Figure 4

LOAD

OUT

SWITCH

NODE

LT1370 · F04

HIGH

FREQUENCY

CIRCULATING

PATH

NFB

GND

S/S

KEEP PATH FROM
V

OUTPUT DIODE,

OUTPUT CAPACITORS
AND GROUND RETURN
AS SHORT AS POSSIBLE

LT1370 · F03

Figure 3. Layout Considerations-- R Package

SwitcherCAD is a trademark of Linear Technology Corporation.

LT1370

TYPICAL APPLICATIO

Positive-to-Negative Converter with Direct Feedback

LT1370

2.7V TO 13V

*BH ELECTRONICS 501-0726

GND

NFB

LT1370 · TA03

S/S

D2
P6KE-15A
D3
1N4148

MBRD835L

C1
100

C2
0.047

0.0047

R1
2k

R3
2.49k
1%

R2
2.49k
1%

OUT

C4
100

OFF

3V
5V
9V

OUT

1.75A
2.25A

T1*

MAX I

OUT

Dual Output Flyback Converter with Overvoltage Protection

LT1370

2.7V TO 10V

*DALE LPE-5047-100MB

GND

NFB

LT1370 · TA04

S/S

P6KE-20A

1N4148

MBRS360T3

C1
22

6.19k

13k

C2
0.047

0.0047

R3
2k

R5
2.49k
1%

R4
12.1k
1%

OUT

15V

OUT

15V

C4
47

C5
47

OFF

2, 3

8, 9

T1*

4
10

LT1370

TYPICAL APPLICATIO

Two Li-Ion Cells to 5V SEPIC Converter**

LT1370

GND

4V TO 9V

LT1370 · TA05

S/S

C1
33

20V

C4
0.047

C5
0.0047

R1
2k

R3
6.19k
1%

R2
18.7k
1%

OUT

C3
100

10V

OFF

L1A*
6.8

L1B*
6.8

4.7

C1 = AVX TPSD 336M020R0200

C2 = TOKIN 1E475ZY5U-C304

C3 = AVX TPSD107M010R0100

BH ELECTRONICS 501-0726

INPUT VOLTAGE MAY BE GREATER OR
LESS THAN OUTPUT VOLTAGE

MBRD835L

4V
5V
7V
9V

OUT

2.2A
2.6A
2.8A

MAX I

OUT

Single Li-Ion Cell to 5V

LT1370

GND

LT1370 · TA06

S/S

L1*

C1**
100

10V

SINGLE
Li-Ion
CELL

C4**
100

10V

0.047

C3
0.0047

R2
6.19k
1%

R1
18.7k
1%

OUT

MBRD835L

OFF

COILCRAFT DO3316P-472
AVX TPSD107M010R0100

2.7V
3.3V
3.6V

OUT

2.5A

3.3A

MAX I

OUT

LT1370

PACKAGE DESCRIPTIO

Dimensions in inches (millimeters) unless otherwise noted.

R Package

7-Lead Plastic DD Pak

(LTC DWG # 05-08-1462)

R (DD7) 0396

0.026 0.036

(0.660 0.914)

0.143

+0.012
0.020

(

)

3.632

+0.305
0.508

0.040 0.060

(1.016 1.524)

0.013 0.023

(0.330 0.584)

0.095 0.115

(2.413 2.921)

0.004

+0.008
0.004

(

)

0.102

+0.203
0.102

0.050

0.012

(1.270

0.305)

0.059

(1.499)

TYP

0.045 0.055

(1.143 1.397)

0.165 0.180

(4.191 4.572)

0.330 0.370

(8.382 9.398)

0.060

(1.524)

TYP

0.390 0.415

(9.906 10.541)

TYP

0.300

(7.620)

0.075

(1.905)

0.183

(4.648)

0.060

(1.524)

0.060

(1.524)

0.256

(6.502)

BOTTOM VIEW OF DD PAK

HATCHED AREA IS SOLDER PLATED

COPPER HEAT SINK

T7 Package

7-Lead Plastic TO-220 (Standard)

(LTC DWG # 05-08-1422)

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

0.040 0.060

(1.016 1.524)

0.026 0.036

(0.660 0.914)

T7 (TO-220) (FORMED) 1197

0.135 0.165

(3.429 4.191)

0.700 0.728

(17.780 18.491)

0.045 0.055

(1.143 1.397)

0.165 0.180

(4.191 4.572)

0.095 0.115

(2.413 2.921)

0.013 0.023

(0.330 0.584)

0.620

(15.75)

TYP

0.155 0.195

(3.937 4.953)

0.152 0.202

(3.860 5.130)

0.260 0.320

(6.604 8.128)

0.147 0.155

(3.734 3.937)

DIA

0.390 0.415

(9.906 10.541)

0.330 0.370

(8.382 9.398)

0.460 0.500

(11.684 12.700)

0.570 0.620

(14.478 15.748)

0.230 0.270

(5.842 6.858)

LT1370

LINEAR TECHNOLOGY CORPORATION 1998

1370f LT/TP 0198 4K · PRINTED IN THE USA

PART NUMBER

DESCRIPTION

COMMENTS

LT1171

100kHz 2.5A Boost Switching Regulator

Good for Up to V

= 40V

LTC

1265

12V 1.2A Monolithic Buck Converter

Converts 5V to 3.3V at 1A with 90% Efficiency

LT1302

Micropower 2A Boost Converter

Converts 2V to 5V at 600mA in SO-8 Packages

LT1372

500kHz 1.5A Boost Switching Regulator

Also Regulates Negative Flyback Outputs

LT1373

Low Supply Current 250kHz 1.5A Boost Switching Regulator

90% Efficient Boost Converter with Constant Frequency

LT1374

500kHz 4.5A Buck Switching Regulator

Converts 12V to 3.3V at 2.5A in SO-8 Package

LT1376

500kHz 1.5A Buck Switching Regulator

Steps Down from Up to 25V Using 4.7

H Inductors

LT1512

500kHz 1.5A SEPIC Battery Charger

Input Voltage May Be Greater or Less Than Battery Voltage

LT1513

500kHz 3A SEPIC Battery Charger

Input Voltage May Be Greater or Less Than Battery Voltage

RELATED PARTS

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

(408) 432-1900

FAX: (408) 434-0507

TELEX: 499-3977

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