June 2000

MIC2182

Micrel

MIC2182

High-Efficiency Synchronous Buck Controller

Final Information

General Description

Micrel's MIC2182 is a synchronous buck (step-down) switch-
ing regulator controller. An all N-channel synchronous archi-
tecture and powerful output drivers allow up to a 20A output
current capabilty. The PWM and skip-mode control scheme
allows efficiency to exceed 95% over a wide range of load
current, making it ideal for battery powered applications, as
well as high current distributed power supplies.

The MIC2182 operates from a 4.5V to 32V input and can
operate with a maximum duty cycle of 86% for use in low-
dropout conditions. It also features a shutdown mode that
reduces quiescent current to 0.1

The MIC2182 achieves high efficiency over a wide output
current range by automatically switching between PWM and
skip mode. Skip-mode operation enables the converter to
maintain high efficiency at light loads by turning off circuitry
pertaining to PWM operation, reducing the no-load supply
current from 1.6mA to 600

A. The operating mode is inter-

nally selected according to the output load conditions. Skip
mode can be defeated by pulling the PWM pin low which
reduces noise and RF interference.

The MIC2182 is available in a 16-pin SOP (small-outline
package) and SSOP (shrink small-outline package) with an
operating range from 40

C to +85

Typical Application

VDD

BST

C6
0.1

R1
2k

R7
100k

4.5V to 30V*

OUT

3.3V/4A

GND

2.2nF

C4
1nF

C3
0.1

C5
0.1

C1
0.1

C9
4.7

16V

C11
22uf
35V
x2

SD103BWS

Q2*
Si4884

D1
B140

C7
220uf
10V

0.02

Q1*
Si4884

HSD

VSW

LSD

PGND

CSH

VOUT

VREF

SGND

GND

MIC2182-3.3BSM

VIN

EN/UVLO

PWM

COMP

SYNC

C13, 1nF

* 30V maximum input voltage limit is due

to standard 30V MOSFET selection.

See "Application Information" section for
5V to 3.3V/10A and other circuits.

4.5V30V* to 3.3V/4A Converter

Micrel, Inc. · 1849 Fortune Drive · San Jose, CA 95131 · USA · tel + 1 (408) 944-0800 · fax + 1 (408) 944-0970 · http://www.micrel.com

Features

· 4.5V to 32V Input voltage range
· 1.25V to 6V Output voltage range
· 95% efficiency
· 300kHz oscillator frequency
· Current sense blanking
· 5

impedance MOSFET Drivers

· Drives N-channel MOSFETs
· 600

A typical quiescent current (skip-mode)

· Logic controlled micropower shutdown (I

< 0.1

· Current-mode control
· Cycle-by-cycle current limiting
· Built-in undervoltage protection
· Adjustable undervoltage lockout
· Easily synchronizable
· Precision 1.245V reference output
· 0.6% total regulation
· 16-pin SOP and SSOP packages
· Frequency foldback overcurrent protection
· Sustained short-circuit protection at any input voltage
· 20A output current capability

Applications

· DC power distribution systems
· Notebook and subnotebook computers
· PDAs and mobile communicators
· Wireless modems
· Battery-operated equipment

MIC2182

Micrel

MIC2182

June 2000

Ordering Information

Part Number

Voltage

Temperature Range

Package

MIC2182BM

Adjustable

C to +85

16-pin narrow SOP

MIC2182-3.3BM

3.3V

C to +85

16-pin narrow SOP

MIC2182-5.0BM

5.0V

C to +85

16-pin narrow SOP

MIC2182BSM

Adjustable

C to +85

16-pin narrow SSOP

MIC2182-3.3BSM

3.3V

C to +85

16-pin narrow SSOP

MIC2182-5.0BSM

5.0V

C to +85

16-pin narrow SSOP

Pin Configuration

HSD

VSW

BST

LSD

PWM

COMP

SGND

MIC2182

PGND

VDD

VIN

VOUT

SYNC

EN/UVLO

CSH

Adjustable

16-pin SOP (M)

16-Pin SSOP (SM)

HSD

VSW

BST

LSD

PWM

COMP

SGND

MIC2182-x.x

PGND

VDD

VIN

VOUT

SYNC

EN/UVLO

VREF

CSH

Fixed

16-pin SOP (M)

16-Pin SSOP (SM)

June 2000

MIC2182

Micrel

Pin Description

Pin Number

Pin Name

Pin Function

Soft-Start (External Component): Connect external capacitor to ground to
reduce inrush current by delaying and slowing the output voltage rise time.
Rise time is controlled by an internal 5

A current source that charges an

external capacitor to V

PWM

PWM/Skip-Mode Select (Input): Low sets PWM-mode operation. 1nF
capacitor to ground sets automatic PWM/skip-mode selection.

COMP

Compensation (Output): Internal error amplifier output. Connect to capacitor
or series RC network to compensate the regulator control loop.

SGND

Small Signal Ground (Return): Route separately from other ground traces to
the () terminal of C

OUT

SYNC

Frequency Synchronization (Input): Optional. Connect to external clock
signal to synchronize the oscillator. Leading edge of signal above the
threshold terminates the switching cycle. Connect to SGND if unused.

EN/UVLO

Enable/Undervoltage Lockout (Input): Low-level signal powers down the
controller. Input below the 2.5V threshold disables switching and functions
as an accurate undervoltage lockout (UVLO). Input below the threshold
forces complete micropower (< 0.1

A) shutdown.

7 (fixed)

VREF

Reference Voltage (Output): 1.245V output. Requires 0.1

f capacitor to

ground.

7 (adj)

Feedback (Input): Regulates FB pin to 1.245V. See "Application Information"
for resistor divider calculations.

CSH

Current-Sense High (Input): Current-limit comparator noninverting input. A
built-in offset of 100mV between CSH and V

OUT

pins in conjunction with the

current-sense resistor set the current-limit threshold level. This is also the
positive input to the current sense amplifier.

VOUT

Current-Sense Low (Input): Output voltage feedback input and inverting
input for the current limit comparator and the current sense amplifier.

VIN

[Battery] Unregulated Input (Input): +4.5V to +32V supply input.

VDD

5V Internal Linear-Regulator (Output): V

is the external MOSFET gate

drive supply voltage and an internal supply bus for the IC. Bypass to SGND
with 4.7

F. V

can supply up to 5mA for external loads.

PGND

MOSFET Driver Power Ground (Return): Connects to source of synchro-
nous MOSFET and the () terminal of C

LSD

Low-Side Drive (Output): High-current driver output for external synchronous
MOSFET. Voltage swing is between ground and V

BST

Boost (Input): Provides drive voltage for the high-side MOSFET driver. The
drive voltage is higher than the input voltage by V

minus a diode drop.

VSW

Switch (Return): High side MOSFET driver return.

HSD

High-Side Drive (Output): High-current driver output for high-side MOSFET.
This node voltage swing is between ground and V

+ 5V V

diode drop

MIC2182

Micrel

MIC2182

June 2000

Electrical Characteristics

= 15V; SS = open; V

PWM

= 0V; V

SHDN

= 5V; I

LOAD

= 0.1A; T

= 25

C, bold values indicate 40

+85

C; Note 4; unless

noted

Parameter

Condition

Min

Typ

Max

Units

MIC2182 [Adjustable], (Note 5)

Feedback Voltage Reference

1.233

1.245

1.257

Feedback Voltage Reference

1.220

1.245

1.270

Feedback Voltage Reference

4.5V < V

< 32V, 0 < V

CSH

OUT

< 75mV

1.208

1.245

1.282

Feedback Bias Current

Output Voltage Range

1.25

Output Voltage Line Regulation

= 4.5V to 32V, V

CSH

OUT

= 50mV

0.03

%/V

Output Voltage Load Regulation

25mV < (V

CSH

OUT

) < 75mV (PWM mode only)

0.5

Output Voltage Total Regulation

0mV < (V

CSH

OUT

) < 75mV (full load range) 4.5V < V

< 32V

0.6

MIC2182-3.3

Output Voltage

3.267

3.3

3.333

Output Voltage

3.234

3.3

3.366

Output Voltage

4.5V < V

< 32V, 0 < V

CSH

OUT

< 75mV

3.201

3.3

3.399

Output Voltage Line Regulation

= 4.5V to 32V, V

CSH

OUT

= 50mV

0.03

%/V

Output Voltage Load Regulation

25mV < (V

CSH

OUT

) < 75mV (PWM mode only)

0.5

Output Voltage Total Regulation

0mV < (V

CSH

OUT

) < 75mV (full load range) 4.5V < V

< 32V

0.8

MIC2182-5.0

Output Voltage

4.95

5.0

5.05

Output Voltage

4.90

5.0

5.10

Output Voltage

6.5V < V

< 32V, 0 < V

CSH

OUT

< 75mV

4.85

5.0

5.150

Output Voltage Line Regulation

= 6.5V to 32V, V

CSH

OUT

= 50mV

0.03

%/V

Output Voltage Load Regulation

25mV < (V

CSH

OUT

) < 75mV (PWM mode only)

0.5

Output Voltage Total Regulation

0mV < (V

CSH

OUT

) < 75mV (full load range) 6.5V < V

< 32V

0.8

Input and VDD Supply

PWM Mode

PWM

= 0V, excluding external MOSFET gate drive current

1.6

2.5

Skip Mode

= 0mA, V

PWM

floating (1nF capacitor to ground)

600

1500

Shutdown Quiescent Current

EN/UVLO

= 0V

0.1

Digital Supply Voltage (V

)

= 0mA to 5mA

4.7

5.3

Undervoltage Lockout

upper threshold (turn on threshold)

4.2

lower threshold (turn off threshold)

4.1

Absolute Maximum Ratings

(Note 1)

Analog Supply Voltage (V

) ....................................... +34V

Digital Supply Voltage (V

) ......................................... +7V

Driver Supply Voltage (B

) .................................... V

+7V

Sense Voltage (V

OUT

, C

) ............................. 7V to 0.3V

Sync Pin Voltage (V

SYNC

) ................................ 7V to 0.3V

Enable Pin Voltage (V

EN/UVLO

) ...................................... V

Power Dissipation (P

)

SOP ................................................ 400mW @ T

= 85

SSOP ............................................. 270mW @ T

= 85

Ambient Storage Temperature (T

) ......... 65

C to +150

ESD, Note 3

Operating Ratings

(Note 2)

Analog Supply Voltage (V

) ........................ +4.5V to +32V

Ambient Temperature (T

) ......................... 40

C to +85

Junction Temperature (T

) ....................... 40

C to +125

Package Thermal Resistance

SOP

(

) .......................................................... 100

C/W

SSOP

(

) ........................................................ 150

C/W

June 2000

MIC2182

Micrel

Parameter

Condition

Min

Typ

Max

Units

Reference Output (Fixed Versions Only)

Reference Voltage

1.220

1.245

1.270

Reference Line Regulation

6V < V

< 32V

Reference Load Regulation

A < I

REF

< 100

Enable/UVLO

Enable Input Threshold

0.6

1.1

1.6

UVLO Threshold

2.2

2.5

2.8

Enable Input Current

EN/UVLO

= 5V

0.1

Soft Start

Soft-Start Current

= 0V

3.5

6.5

Current Limit

Current-Limit Threshold Voltage

CSH

= V

OUT

100

135

Error Amplifier

Error Sense Amplifier Gain

Current Amp

Current Sense Amplifier Gain

2.0

Oscillator Section

Oscillator Frequency

270

300

330

kHz

Maximum Duty Cycle

Minimum On-Time

OUT

= V

OUT(nominal)

+ 200mV

140

250

SYNC Threshold Level

0.7

1.3

1.9

SYNC Input Current

SYNC

= 5V

0.1

SYNC Minimum Pulse Width

200

SYNC Capture Range

Note 6

330

kHz

Frequency Foldback Threshold

measured at VOUT pin

0.75

0.95

1.15

Foldback Frequency

kHz

Gate Drivers

Rise/Fall Time

= 3000pF

Output Driver Impedance

source

8.5

sink

3.5

Driver Nonoverlap Time

PWM Input

PWM Input Current

PWM

= 0V

Note 1.

Exceeding the absolute maximum rating may damage the device.

Note 2.

The device is not guaranteed to function outside its operating rating.

Note 3.

Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.

Note 4.

C limits are 100% production tested. Limits over the operating temperature range are guaranteed by design and are not production tested.

Note 5.

> 1.3

OUT

(for the feedback voltage reference and output voltage line and total regulation).

Note 6.

See applications information for limitations on the maximum operating frequency.

MIC2182

Micrel

MIC2182

June 2000

0.2

0.4

0.6

0.8

1.0

1.2

1.4

1.6

1.8

2.0

-40 -20 0 20 40 60 80 100120140

CURRENT (mA)

TEMPERATURE (

Quiescent Current

vs. Temperature

PWM

Skip

0.05

0.10

0.15

0.20

-40 -20 0 20 40 60 80 100120140

TEMPERATURE (

-0.50

0.50

1.00

1.50

Quiescent Current

vs. Temperature

UVLO Mode

(mA)

SHUTDOWN

(

CURRENT

(
mA

)

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

12 16 20 24 28 32

CURRENT (mA)

INPUT VOLTAGE (V)

Quiescent Current

vs. Supply Voltage

PWM

Skip

-0.5

0.5

1.0

1.5

0.1

0.2

0.3

0.4

0.5

12 16 20 24 28 32

SUPPLY VOLTAGE (V)

Quiescent Current

vs. Supply Voltage

UVLO Mode

(mA)

SHUTDOWN

(

CURRENT (

1.236

1.238

1.240

1.242

1.244

1.246

1.248

1.250

1.252

1.254

1.256

12 16 20 24 28 32

REFERENCE VOLTAGE (V)

SUPPLY VOLTAGE (V)

REF

(Fixed Versions)

Line Regulation

1.200

1.210

1.220

1.230

1.240

1.250

1.260

200

400

600

800

1000

REFERENCE VOLTAGE (V)

LOAD CURRENT (

REF

(Fixed Versions)

Load Regulation

1.240

1.245

1.250

1.255

1.260

-40 -20 0 20 40 60 80 100120140

REFERENCE VOLTAGE (V)

TEMPERATURE (

REF

(Fixed Versions)

vs. Temperature

4.0

4.2

4.4

4.6

4.8

5.0

12 16 20 24 28 32

REGULATOR VOLTAGE (V)

SUPPLY VOLTAGE (V)

Line Regulation

4.80

4.85

4.90

4.95

5.00

REGULATOR VOLTAGE (V)

LOAD CURRENT (mA)

Load Regulation

4.82

4.84

4.86

4.88

4.90

4.92

4.94

4.96

4.98

-40 -20 0 20 40 60 80 100120140

REGULATOR VOLTAGE (V)

TEMPERATURE (

vs. Temperature

-10

-8

-6

-4

-2

-40 -20 0 20 40 60 80 100120140

FREQUENCY VARIATION (%)

TEMPERATURE (

Oscillator Frequency

vs. Temperature

Typical Characteristics

-1.0

-0.8

-0.6

-0.4

-0.2

0.2

0.4

0.6

0.8

1.0

12 16 20 24 28 32

FREQUENCY VARIATION (%)

SUPPLY VOLTAGE (V)

Oscillator Frequency

vs. Supply Voltage

MIC2182

Micrel

MIC2182

June 2000

Functional Description

See "Applications Information" following this section for com-
ponent selection information and Figure 14 and Tables 1
through 5 for predesigned circuits.

The MIC2182 is a BiCMOS, switched-mode, synchronous
step-down (buck) converter controller. Current-mode control
is used to achieve superior transient line and load regulation.
An internal corrective ramp provides slope compensation for
stable operation above a 50% duty cycle. The controller is
optimized for high-efficiency, high-performance dc-dc con-
verter applications.

The MIC2182 block diagrams are shown in Figure 2a and
Figure 2b.

The MIC2182 controller is divided into 6 functions.

· Control loop

- PWM operation
- Skip-mode operation

· Current limit

· Reference, enable and UVLO

· MOSFET gate drive

· Oscillator and sync

· Soft start

Control Loop

PWM and Skip Modes of Operation

The MIC2182 operates in PWM (pulse-width-modulation)
mode at heavier output load conditions. At lighter load condi-
tions, the controller can be configured to automatically switch
to a pulse-skipping mode to improve efficiency. The potential
disadvantage of skip mode is the variable switching fre-
quency that accompanies this mode of operation. The occur-
rence of switching pulses depends on component values as
well as line and load conditions. There is an external sync
function that is disabled in skip mode. In PWM mode, the
synchronous buck converter forces continuous current to
flow in the inductor. In skip mode, current through the inductor
can settle to zero, causing voltage ringing across the induc-
tor. Pulling the PWM pin (pin 2) low will force the controller to
operate in PWM mode for all load conditions, which will
improve cross regulation of transformer coupled, multiple
output configurations.

PWM Control Loop

The MIC2182 uses current-mode control to regulate the
output voltage. This method senses the output voltage (outer
loop) and the inductor current (inner loop). It uses inductor
current and output voltage to determine the duty cycle of the
buck converter. Sampling the inductor current removes the
inductor from the control loop, which simplifies compensa-
tion.

Current
Sense
Amp

1.245V

Error
Amp

CORRECTIVE

RAMP

PWM

COMPARATOR

RESET

Reference

Oscillator

COMP

100k

MIC2182 [adj.] PWM Mode

VOUT

CSH

VDD

VBST

VIN

HSD

BST

OUT

VSW

LSD

PGND

COMP

4.7

1.245V

OUT

CONTROL LOGIC AND

PULSE-WIDTH MODULATOR

= 0.2

-3

= 2

0.024V

PWM Mode
to Skip
Mode

LOW
FORCES
SKIP MODE

Figure 3. PWM Operation

June 2000

MIC2182

Micrel

A block diagram of the MIC2182 PWM current-mode control
loop is shown in Figure 3 and the PWM mode voltage and
current waveforms are shown in figure 5A. The inductor
current is sensed by measuring the voltage across the
resistor, R

. A ramp is added to the amplified current-sense

signal to provide slope compensation, which is required to
prevent unstable operation at duty cycles greater than 50%.

A transconductance amplifier is used for the error amplifier,
which compares an attenuated sample of the output voltage
with a reference voltage. The output of the error amplifier is
the COMP (compensation) pin, which is compared to the
current-sense waveform in the PWM block. When the current
signal becomes greater than the error signal, the comparator
turns off the high-side drive. The COMP pin (pin 3) provides
access to the output of the error amplifier and allows the use
of external components to stabilize the voltage loop.

Skip-Mode Control Loop

This control method is used to improve efficiency at light
output loads. At light output currents, the power drawn by the
MIC2182 is equal to the input voltage times the IC supply
current (I

). At light output currents, the power dissipated by

the IC can be a significant portion of the total output power,
which lowers the efficiency of the power supply. The MIC2182
draws less supply current in skip mode by disabling portions
of the control and drive circuitry when the IC in not switching.
The disadvantage of this method is greater output voltage
ripple and variable switching frequency.

A block diagram of the MIC2182 skip mode is shown in Figure
4. Skip mode voltage and current waveforms are shown in
figure 5B.

Low
Comp

Skip-Mode
Current
Limit

1.245V

0.07V

LOW
FORCES
PWM MODE

2%V

Reference

MIC2182 [adj.] Skip Mode

VOUT

CSH

VDD

VBST

VIN

HSD

BST

OUT

VSW

LSD

PGND

4.7

1.245V

OUT

Current
Sense
Amp

CONTROL LOGIC AND
SKIP-MODE LOGIC

= 2

ONE SHOT

LOW-SIDE DRIVER

ONE SHOT

Hysteresis
Comp

Figure 4. Skip-Mode Operation

MIC2182

Micrel

MIC2182

June 2000

A hysteretic comparator is used in place of the PWM error
amplifier and a current-limit comparator senses the inductor
current. A one-shot starts the switching cycle by momentarily
turning on the low side MOSFET to insure the high side drive
boost capacitor, Cbst, is fully charged. The high side MOS-
FET is turned on and current ramps up in the inductor, L1.
The high side drive is turned off when either the peak voltage
on the input of the current-sense comparator exceeds the
threshold, typically 35mV, or the output voltage rises above
the hysteretic threshold of the output voltage comparator.
Once the high side MOSFET is turned off, the load current
discharges the output capacitor, causing V

OUT

to fall. The

cycle repeats when V

OUT

falls below the lower threshold,

1%.

The maximum peak inductor current depends on the skip-
mode current-limit threshold and the value of the current-
sense resistor, R

35mV

inductor(peak)

sense

Figure 6 shows the improvement in efficiency that skip mode
makes when at lower output currents.

100

0.01

0.1

100

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

Figure 6. Efficiency

+ V

Reset

Pulse

HSD

LSD

LOAD

Figure 5a. PWM-Mode Timing

+1%

NOMINAL

OUT

LSD

HSD

one-shot

LIM(skip)

OUT

Figure 5b. Skip-Mode Timing

June 2000

MIC2182

Micrel

Switching from PWM to Skip Mode

The current sense amplifier in Figure 3 monitors the average
voltage across the current-sense resistor. The controller will
switch from PWM to skip mode when the average voltage
across the current-sense resistor drops below approximately
12mV. This is shown in Figure 7b. The average output current
at this transition level for is calculated below.

0.012

OUT(skipmode)

where:

0.012 = threshold voltage of the internal comparator

= current-sense resistor value

Switching from Skip to PWM Mode

The frequency of occurrence of the skip-mode current pulses
increase as the output current increases until the hysteretic
duty cycle reaches 100% (continuous pulses). Increasing the
current past this point will cause the output voltage will drop.
The low limit comparator senses the output voltage when it
drops below 2% of the set output and automatically switches
the converter to PWM mode.

The inductor current in skip mode is a triangular wave shape
a minimum value of 0 and a maximum value of 35mV/R

(see Figure 7b). The maximum average output current in skip
mode is the average value of the inductor waveform:

0.5

35mV

OUT(max skipmode)

The capacitor on the PWM pin (pin 2) is discharged when the
IC transitions from skip to PWM mode. This forces the IC to
remain in PWM mode for a fixed period of time. The added
delay prevents unwanted switching between PWM and skip
mode. The capacitor is charged with a 10uA current source
on pin 2. The threshold on pin 2 is 2.5V. The delay for a typical
1nF capacitor is:

1nF

2.5V

10 A

250 s

delay

PWM

threshold

source

where:

PWM

= capacitor connected to pin 2

Current Limit

The current-limit circuit operates during PWM mode. The
output current is detected by the voltage drop across the
external current-sense resistor (R

in Figure 2.). The cur-

rent-limit threshold is 100mV+35mV 25mV. The current-
sense resistor must be sized using the minimum current-limit
threshold. The external components must be designed to
withstand the maximum current limit. The current-sense
resistor value is calculated by the equation below:

75mV

OUT(max)

The maximum output current is:

135mV

OUT(max)

The current-sense pins CSH (pin 8) and V

OUT

(pin 9) are

noise sensitive due to the low signal level and high input
impedance. The PCB traces should be short and routed close
to each other. A small (1nF to 0.1

F) capacitor across the pins

will attenuate high frequency switching noise.

When the peak inductor current exceeds the current-limit
threshold, the current-limit comparator, in Figure 2, turns off
the high-side MOSFET for the remainder of the cycle. The
output voltage drops as additional load current is pulled from
the converter. When the output voltage reaches approxi-
mately 0.95V, the circuit enters frequency-foldback mode
and the oscillator frequency will drop to 60kHz while maintain-
ing the peak inductor current equal to the nominal 100mV
across the external current-sense resistor. This limits the
maximum output power delivered to the load under a short
circuit condition.

Reference, Enable and, UVLO Circuits

The output drivers are enabled when the following conditions
are satisfied:

· The V

voltage (pin 11) is greater than its

undervoltage threshold (typically 4.2V).

· The voltage on the enable pin is greater than the

enable UVLO threshold (typically 2.5V)

The internal bias circuit generates a 1.245V bandgap refer-
ence voltage for the voltage error amplifier and a 5V V

voltage for the gate drive circuit. The reference voltage in the
fixed-output-voltage versions of the MIC2182 is buffered and
brought to pin 7. The V

REF

pin should be bypassed to GND

(pin 4) with a 0.1

F capacitor. The adjustable version of the

MIC2182 uses pin 7 for output voltage sensing. A decoupling
capacitor on pin 7 is not used in the adjustable output voltage
version.

LIM(skip)

Inductor

Current

35mV THRESHOLD
ACROSS R

Figure 7a. Maximum Skip-Mode-Load Inductor Current

MIN(PWM)

Inductor

Current

12mV THRESHOLD
OF AVERAGE VOLTAGE
ACROSS R

Figure 7b. Minimum PWM-Mode-Load Inductor Current for PWM Operation

MIC2182

Micrel

MIC2182

June 2000

The enable pin (pin 6) has two threshold levels, allowing the
MIC2182 to shut down in a low current mode, or turn off output
switching in UVLO mode. An enable pin voltage lower than
the shutdown threshold turns off all the internal circuitry and
reduces the input current to typically 0.1

If the enable pin voltage is between the shutdown and UVLO
thresholds, the internal bias, V

, and reference voltages are

turned on. The soft-start pin is forced low by an internal
discharge MOSFET. The output drivers are inhibited from
switching and remain in a low state. Raising the enable
voltage above the UVLO threshold of 2.5V allows the soft-
start capacitor to charge and enables the output drivers.

Either of two UVLO conditions will pull the soft-start capacitor
low.

· When the V

drops below 4.1V

· When the enable pin drops below the 2.5V

threshold

MOSFET Gate Drive

The MIC2182 high-side drive circuit is designed to switch an
N-channel MOSFET. Referring to the block diagram in Figure
2, a bootstrap circuit, consisting of D2 and C

BST

, supplies

energy to the high-side drive circuit. Capacitor C

BST

charged while the low-side MOSFET is on and the voltage on
the V

pin (pin 15) is approximately 0V. When the high-side

MOSFET driver is turned on, energy from C

BST

is used to turn

the MOSFET on. As the MOSFET turns on, the voltage on the
V

pin increases to approximately V

. Diode D2 is re-

versed biased and C

BST

floats high while continuing to keep

the high-side MOSFET on. When the low-side switch is
turned back on, C

BST

is recharged through D2.

The drive voltage is derived from the internal 5V V

bias

supply. The nominal low-side gate drive voltage is 5V and the
nominal high-side gate drive voltage is approximately 4.5V
due the voltage drop across D2. A fixed 80ns delay between
the high- and low-side driver transitions is used to prevent
current from simultaneously flowing unimpeded through both
MOSFETs.

Oscillator and Sync

The internal oscillator is free running and requires no external
components. The nominal oscillator frequency is 300kHz. If
the output voltage is below approximately 0.95V, the oscilla-
tor operates in a frequency-foldback mode and the switching
frequency is reduced to 60kHz.

The SYNC input (pin 5) allows the MIC2182 to synchronize
with an external clock signal. The rising edge of the sync
signal generates a reset signal in the oscillator, which turns
off the low-side gate drive output. The high-side drive then
turns on, restarting the switching cycle. The sync signal is
inhibited when the controller operates in skip mode or during
frequency foldback. The sync signal frequency must be
greater than the maximum specified free running frequency
of the MIC2182. If the synchronizing frequency is lower,
double pulsing of the gate drive outputs will occur. When not
used, the sync pin must be connected to ground.

Figure 8 shows the timing between the external sync signal
(trace 2), the low-side drive (trace 1) and the high-side drive
(trace R1). There is a delay of approximately 250ns between
the rising edge of the external sync signal and turnoff of the
low-side MOSFET gate drive.

Some concerns of operating at higher frequencies are:

· Higher power dissipation in the internal V

regulator. This occurs because the MOSFET
gates require charge to turn on the device. The
average current required by the MOSFET gate
increases with switching frequency. This in-
creases the power dissipated by the internal
V

regulator. Figure 10 shows the total gate

charge which can be driven by the MIC2182
over the input voltage range, for different values
of switching frequency. The total gate charge
includes both the high- and low-side MOSFETs.
The larger SOP package is capable of dissipat-
ing more power than the SSOP package and
can drive larger MOSFETs with higher gate
drive requirements.

TIME

SYNC

SIGNAL

-
SIDE

DRIVE

HIGH-SIDE

DRIVE

Figure 8. Sync Waveforms

TIME

OUT

Figure 9. Startup Waveforms

June 2000

MIC2182

Micrel

· Reduced maximum duty cycle due to switching

transition times and constant delay times in the
controller. As the switching frequency increased,
the switching period decreases. The switching
transition times and constant delays in the
MIC2182 start to become noticeable. The effect
is to reduce the maximum duty cycle of the
controller. This will cause the minimum input to
output differential voltage (dropout voltage) to
increase.

100

12 16 20 24 28 32

GATE CHARGE (nC)

SUPPLY VOLTAGE (V)

400kHz

300kHz

500kHz

SOP

Figure 10a. SOP Gate Charge vs. Input Voltage

100

12 16 20 24 28 32

GATE CHARGE (nC)

SUPPLY VOLTAGE (V)

SSOP

400kHz

300kHz

500kHz

Figure 10b. SSOP Gate Charge vs. Input Voltage

It is recommended that the user limits the maximum synchro-
nized frequency to 600kHz. If a higher synchronized fre-
quency is required, it may be possible and will be design
dependent. Please consult Micrel applications for assis-
tance.

Soft Start

Soft start reduces the power supply input surge current at
startup by controlling the output voltage rise time. The input
surge appears while the output capacitance is charged up. A
slower output rise time will draw a lower input surge current.
Soft start may also be used for power supply sequencing.

The soft-start voltage is applied directly to the PWM compara-
tor. A 5uA internal current source is used to charge up the
soft-start capacitor. The capacitor is discharged when either
the enable voltage drops below the UVLO threshold (2.5V) or
the V

voltage drops below the UVLO level (4.1V).

The part switches at a minimum duty cycle when the soft-start
pin voltage is less than 0.4V. This maintains a charge on the
bootstrap capacitor and insures high-side gate drive voltage.
As the soft-start voltage rises above 0.4V, the duty cycle
increases from the minimum duty cycle to the operating duty
cycle. The oscillator runs at the foldback frequency of 60kHz
until the output voltage rises above 0.95V. Above 0.95V, the
switching frequency increases to 300kHz (or the sync'd
frequency), causing the output voltage to rise a greater rate.
The rise time of the output is dependent on the soft-start
capacitor, output capacitance, output voltage, and load cur-
rent. The oscilloscope photo in Figure 9 show the output
voltage and the soft-start pin voltage at startup.

Minimum Pulse Width

The MIC2182 has a specified minimum pulse width. This
minimum pulse width places a lower limit on the minimum
duty cycle of the buck converter. When the MIC2182 is
operating in forced PWM mode (pin 2 low) and when the
output current is very low or zero, there is a limit on the ratio
of V

OUT

. If this limit is exceeded, the output voltage will

rise above the regulated voltage level. A minimum load is
required to prevent the output from rising up. This will not
occur for output voltages greater than 3V.

Figure 11 should be used as a guide when the MIC2182 is
forced into PWM-only mode. The actual maximum input
voltage will depend on the exact external components used
(MOSFETs, inductors, etc.).

INPUT VOLTAGE (V)

OUTPUT VOLTAGE (V)

Figure 11. Max. Input Voltage in Forced-PWM Mode

This restriction does not occur when the MIC2182 is set to
automatic mode (pin 2 connected to a capacitor) since the
converter operates in skip mode at low output current.

MIC2182

Micrel

MIC2182

June 2000

Applications Information

The following applications information includes component
selection and design guidelines. See Figure 14 and Tables 1
through 5 for predesigned circuits.

Inductor Selection

Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak to peak inductor
ripple current. Generally, higher inductance values are used
with higher input voltages. Larger peak to peak ripple currents
will increase the power dissipation in the inductor and
MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple
current. Smaller peak to peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current.

The inductance value is calculated by the equation below.

)

0.2 I

OUT

IN(max)

OUT

IN(max)

OUT(max)

× ×

where:

= switching frequency

0.2 = ratio of ac ripple current to dc output current

IN(max)

= maximum input voltage

The peak-to-peak inductor current (ac ripple current) is:

)

OUT

IN(max)

OUT

IN(max)

× ×

The peak inductor current is equal to the average output
current plus one half of the peak to peak inductor ripple
current.

0.5 I

OUT(max)

The RMS inductor current is used to calculate the I

·R losses

in the inductor.

(rms)

inductor

OUT(max)

Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2182 requires the use of ferrite
materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply. This
is especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.

The power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At lower

output currents, the core losses can be a significant contribu-
tor. Core loss information is usually available from the mag-
netics vendor.

Copper loss in the inductor is calculated by the equation
below:

(rms)

inductor Cu

inductor

winding

The resistance of the copper wire, R

winding

, increases with

temperature. The value of the winding resistance used should
be at the operating temperature.

1 0.0042 (T

)

winding(hot)

winding(20 C)

hot

20 C

× +

(

)

where:

HOT

= temperature of the wire

under operating load

= ambient temperature

winding(20

is room temperature winding resistance

(usually specified by the manufacturer)

Current-Sense Resistor Selection

Low inductance power resistors, such as metal film resistors
should be used. Most resistor manufacturers make low
inductance resistors with low temperature coefficients, de-
signed specifically for current-sense applications. Both resis-
tance and power dissipation must be calculated before the
resistor is selected. The value of R

SENSE

is chosen based on

the maximum output current and the maximum threshold
level. The power dissipated is based on the maximum peak
output current at the minimum overcurrent threshold limit.

75mV

SENSE

OUT(max)

The maximum overcurrent threshold is:

135mV

overcurrent(max)

The maximum power dissipated in the sense resistor is:

D(R

)

overcurrent(max)

SENSE

MOSFET Selection

External N-channel logic-level power MOSFETs must be
used for the high- and low-side switches. The MOSFET gate-
to-source drive voltage of the MIC2182 is regulated by an
internal 5V V

regulator. Logic-level MOSFETs, whose

operation is specified at V

= 4.5V must be used.

It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75

C rise in junc-

tion temperature will increase the channel resistance of the
MOSFET by 50% to 75% of the resistance specified at 25

This change in resistance must be accounted for when
calculating MOSFET power dissipation.

Total gate charge is the charge required to turn the MOSFET
on and off under specified operating conditions (V

and

). The gate charge is supplied by the MIC2182 gate drive

circuit. At 300kHz switching frequency and above, the gate

June 2000

MIC2182

Micrel

charge can be a significant source of power dissipation in the
MIC2182. At low output load this power dissipation is notice-
able as a reduction in efficiency. The average current re-
quired to drive the high-side MOSFET is:

G[high-side](avg)

where:

G[high-side](avg)

average high-side MOSFET gate current

= total gate charge for the high-side MOSFET

taken from manufacturer's data sheet
with V

= 5V.

The low-side MOSFET is turned on and off at V

= 0

because the freewheeling diode is conducting during this
time. The switching losses for the low-side MOSFET is
usually negligable. Also, the gate drive current for the low-
side MOSFET is more accurately calculated using C

ISS

= 0 instead of gate charge.

For the low-side MOSFET:

G[low-side](avg)

ISS

Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2182 due to gate
drive is:

gate drive

IN G[high-side](avg)

G[low-side](avg)

(

)

A convenient figure of merit for switching MOSFETs is the on-
resistance times the total gate charge (R

DS(on)

). Lower

numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2182. Power dissipation in the MIC2182 package limits
the maximum gate drive current. Refer to Figure 10 for the
MIC2182 gate drive limits.

Parameters that are important to MOSFET switch selection
are:

· Voltage rating

· On-resistance

· Total gate charge

The voltage rating of the MOSFETs are essentially equal to
the input voltage. A safety factor of 20% should be added to
the V

DS(max)

of the MOSFETs to account for voltage spikes

due to circuit parasitics.

The power dissipated in the switching transistor is the sum of
the conduction losses during the on-time (P

conduction

) and the

switching losses that occur during the period of time when the
MOSFETs turn on and off (P

conduction

where:

(rms)

conduction

AC(off)

AC(on)

= on-resistance of the MOSFET switch.

Making the assumption the turn-on and turnoff transition
times are equal, the transition time can be approximated by:

ISS

OSS

where:

ISS

and C

OSS

are measured at V

= 0.

= gate drive current (1A for the MIC2182)

The total high-side MOSFET switching loss is:

V ) I

where:

= switching transition time

(typically 20ns to 50ns)

= freewheeling diode drop, typically 0.5V.

it the switching frequency, nominally 300kHz

The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.

RMS Current and MOSFET Power Dissipation Calculation

Under normal operation, the high-side MOSFET's RMS
current is greatest when V

is low (maximum duty cycle). The

low-side MOSFET's RMS current is greatest when V

is high

(minimum duty cycle). However, the maximum stress the
MOSFETs see occurs during short circuit conditions, where
the output current is equal to I

overcurrent(max)

. (See the Sense

Resistor section). The calculations below are for normal
operation. To calculate the stress under short circuit condi-
tions, substitute I

overcurrent(max)

for I

OUT(max)

. Use the formula

below to calculate D under short circuit conditions.

0.063 1.8 10

short circuit

The RMS value of the high-side switch current is:

(rms)

SW(highside)

OUT(max)

(rms)

1 D I

SW(low side)

OUT(max)

(

)

where:

D = duty cycle of the converter

OUT

= efficiency of the converter.

Converter efficiency depends on component parameters,
which have not yet been selected. For design purposes, an
efficiency of 90% can be used for V

less than 10V and 85%

can be used for V

greater than 10V. The efficiency can be

more accurately calculated once the design is complete. If the
assumed efficiency is grossly inaccurate, a second iteration
through the design procedure can be made.

For the high-side switch, the maximum dc power dissipation
is:

(rms)

switch1(dc)

DS(on)1

SW1

MIC2182

Micrel

MIC2182

June 2000

For the low-side switch (N-channel MOSFET), the dc power
dissipation is:

(rms)

switch2(dc)

DS(on)2

SW 2

Since the ac switching losses for the low side MOSFET is
near zero, the total power dissipation is:

low-side MOSFET(max)

switch2(dc)

The total power dissipation for the high side MOSFET is:

highsideMOSFET(max)

SWITCH 1(dc)

External Schottky Diode

An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 80ns The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must be
able to handle the peak current.

80ns

D(avg)

OUT

× ×

The reverse voltage requirement of the diode is:

(rrm)

diode

The power dissipated by the Schottky diode is:

diode

D(avg)

where:

= forward voltage at the peak diode current

The external Schottky diode, D2, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode is
used, it must be rated to handle the peak and average current.
The body diode has a relatively slow reverse recovery time
and a relatively high forward voltage drop. The power lost in
the diode is proportional to the forward voltage drop of the
diode. As the high-side MOSFET starts to turn on, the body
diode becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the

circuit inductance will cause ringing during the high-side
MOSFET turn-on.

An external Schottky diode conducts at a lower forward
voltage preventing the body diode in the MOSFET from
turning on. The lower forward voltage drop dissipates less
power than the body diode. The lack of a reverse recovery
mechanism in a Schottky diode causes less ringing and less
power loss. Depending on the circuit components and oper-
ating conditions, an external Schottky diode will give a 1/2%
to 1% improvement in efficiency. Figure 12 illustrates the
difference in noise on the VSW pin with and without a
Schottky diode.

Output Capacitor Selection

The output capacitor values are usually determined by the
capacitors ESR (equivalent series resistance). Voltage rating
and RMS current capability are two other important factors in
selecting the output capacitor. Recommended capacitors are
tantalum, low-ESR aluminum electrolytics, and OS-CON.

The output capacitor's ESR is usually the main cause of
output ripple. The maximum value of ESR is calculated by:

ESR

OUT

where:

OUT

= peak to peak output voltage ripple

= peak to peak inductor ripple current

The total output ripple is a combination of the ESR and the
output capacitance. The total ripple is calculated below:

(1 D)

OUT

ESR

× -

(

)

where:

D = duty cycle

OUT

= output capacitance value

= switching frequency

The voltage rating of capacitor should be twice the output
voltage for a tantalum and 20% greater for an aluminum
electrolytic or OS-CON.

The output capacitor RMS current is calculated below:

(rms)

OUT

The power dissipated in the output capacitor is:

(rms)

DISS(C

ESR(C

)

OUT

)

Input Capacitor Selection

The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may fail
when subjected to high inrush currents, caused by turning the
input supply on. Tantalum input capacitor voltage rating
should be at least 2 times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage derating.

TIME

WITH

FREEWHEELING DIODE

WITHOUT

FREEWHEELING DIODE

Figure 12. Switch Output Noise

With and Without Shottky Diode

June 2000

MIC2182

Micrel

The input voltage ripple will primarily depend on the input
capacitors ESR. The peak input current is equal to the peak
inductor current, so:

inductor(peak)

ESR(C )

The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at the
maximum output current. Assuming the peak to peak induc-
tor ripple current is low:

(rms)

D (1 D)

OUT(max)

× -

The power dissipated in the input capacitor is:

(rms)

DISS(C )

ESR(C )

Voltage Setting Components

The MIC2182-3.3 and MIC2182-5.0 ICs contain internal
voltage dividers that set the output voltage. The MIC2182
adjustable version requires two resistors to set the output
voltage as shown in Figure 13.

Error
Amp

MIC2182 [adj.]

REF

1.245V

Figure 13. Voltage-Divider Configuration

The output voltage is determined by the equation:

REF

× +

Where: V

REF

for the MIC2182 is typically 1.245V.

A typical value of R1 can be between 3k and 10k. If R1 is too
large it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small in value it will decrease the
efficiency of the power supply, especially at low output loads.

Once R1 is selected, R2 can be calculated using:

REF

Voltage Divider Power Dissipation

The reference voltage and R2 set the current through the
voltage divider.

divider

REF

The power dissipated by the divider resistors is:

(R1 R2) I

divider

Efficiency Calculation and Considerations

Efficiency is the ratio of output power to input power. The
difference is dissipated as heat in the buck converter. Under
light output load, the significant contributors are:

· Supply current to the MIC2182

· MOSFET gate-charge power (included in the IC

supply current)

· Core losses in the output inductor

To maximize efficiency at light loads:

· Use a low gate-charge MOSFET or use the

smallest MOSFET, which is still adequate for
maximum output current.

· Allow the MIC2182 to run in skip mode at lower

currents.

· Use a ferrite material for the inductor core, which

has less core loss than an MPP or iron power
core.

Under heavy output loads the significant contributors to
power loss are (in approximate order of magnitude):

· Resistive on-time losses in the MOSFETs

· Switching transition losses in the MOSFETs

· Inductor resistive losses

· Current-sense resistor losses

· Input capacitor resistive losses (due to the

capacitors ESR)

To minimize power loss under heavy loads:

· Use logic-level, low on-resistance MOSFETs.

Multiplying the gate charge by the on-resistance
gives a Figure of merit, providing a good bal-
ance between low and high load efficiency.

· Slow transition times and oscillations on the

voltage and current waveforms dissipate more
power during turn-on and turnoff of the
MOSFETs. A clean layout will minimize parasitic
inductance and capacitance in the gate drive
and high current paths. This will allow the fastest
transition times and waveforms without oscilla-
tions. Low gate-charge MOSFETs will transition
faster than those with higher gate-charge
requirements.

· For the same size inductor, a lower value will

have fewer turns and therefore, lower winding
resistance. However, using too small of a value
will require more output capacitors to filter the
output ripple, which will force a smaller band-
width, slower transient response and possible
instability under certain conditions.

· Lowering the current-sense resistor value will

decrease the power dissipated in the resistor.
However, it will also increase the overcurrent
limit and will require larger MOSFETs and
inductor components.

· Use low-ESR input capacitors to minimize the

power dissipated in the capacitors ESR.

Decoupling Capacitor Selection

The 4.7

F decoupling capacitor is used to minimize noise on

the VDD pin. The placement of this capacitor is critical to the
proper operation of the IC. It must be placed right next to the

MIC2182

Micrel

MIC2182

June 2000

pins and routed with a wide trace. The capacitor should be a
good quality tantalum. An additional 1

F ceramic capacitor

may be necessary when driving large MOSFETs with high
gate capacitance. Incorrect placement of the V

decoupling

capacitor will cause jitter or oscillations in the switching
waveform and large variations in the overcurrent limit.

A 0.1

F ceramic capacitor is required to decouple the VIN.

The capacitor should be placed near the IC and connected
directly to between pin 10 (Vcc) and pin 12 (PGND).

PCB Layout and Checklist

PCB layout is critical to achieve reliable, stable and efficient
performance. A ground plane is required to control EMI and
minimize the inductance in power, signal and return paths.

The following guidelines should be followed to insure proper
operation of the circuit.

· Signal and power grounds should be kept

separate and connected at only one location.
Large currents or high di/dt signals that occur
when the MOSFETs turn on and off must be
kept away from the small signal connections.

· The connection between the current-sense

resistor and the MIC2182 current-sense inputs
(pin 8 and 9) should have separate traces,
routed from the terminals directly to the IC pins.
The traces should be routed as closely as
possible to each other and their length should be
minimized. Avoid running the traces under the
inductor and other switching components. A 1nF
to 0.1

F capacitor placed between pins 8 and 9

will help attenuate switching noise on the current
sense traces. This capacitor should be placed
close to pins 8 and 9.

· When the high-side MOSFET is switched on, the

critical flow of current is from the input capacitor
through the MOSFET, inductor, sense resistor,
output capacitor, and back to the input capacitor.
These paths must be made with short, wide
pieces of trace. It is good practice to locate the
ground terminals of the input and output capaci-
tors close to each.

· When the low-side MOSFET is switched on,

current flows through the inductor, sense
resistor, output capacitor, and MOSFET. The
source of the low-side MOSFET should be
located close to the output capacitor.

· The freewheeling diode, D1 in Figure 2, con-

ducts current during the dead time, when both
MOSFETs are off. The anode of the diode
should be located close to the output capacitor
ground terminal and the cathode should be
located close to the input side of the inductor.

· The 4.7

F capacitor, which connects to the VDD

terminal (pin 11) must be located right at the IC.
The VDD terminal is very noise sensitive and
placement of this capacitor is very critical.
Connections must be made with wide trace. The
capacitor may be located on the bottom layer of
the board and connected to the IC with multiple
vias.

· The V

bypass capacitor should be located

close to the IC and connected between pins 10
and 12. Connections should be made with a
ground and power plane or with short, wide
trace.

June 2000

MIC2182

Micrel

VDD

BST

C6
0.1

R1
2k

R7
100k

OUT

GND

2.2nF

C4
1nF

C3
0.1

C5
0.1

C1
0.1

50V

C9
4.7

16V

C11
(table)

SD103BWS

Q2
(table)

D1
(table)

C7
(table)

C12
0.1

50V

(table)

Q1
(table)

HSD

VSW

LSD

PGND

CSH

VOUT

VREF

SGND

GND

MIC2182

VIN

EN/UVLO

PWM

COMP

SYNC

C13, 1nF

Figure 14. Basic Circuit Diagram for Use with Tables 3 through 6

Table 1. Specifications for Figure 14 and Tables 3 through 6

Manufacturer

Telephone Number (USA)

Web Address

AVX

(803) 946-0690

www.avxcorp.com

Central Semiconductor

(516) 435-1110

www.centralsemi.com

Coiltronics

(561) 241-7876

www.coiltronics.com

IRC

(704) 264-8861

(310) 322-3331

www.irf.com

Micrel

(408) 944-0800

www.micrel.com

Vishay/Lite On

(805) 446-4800

www.vishay-liteon.com

(diodes)

Vishay/Siliconix

(800) 554-5665

www.siliconix.com

(MOSFETs)

Vishay/Dale

(800) 487-9437

www.vishaytechno.com

(inductors and resistors)

Sumida

(847) 956-0666

www.japanlink.com/sumida

Table 2. Component Suppliers

Predesigned Circuits

A single schematic diagram, shown in Figure 14, can be used
to build power supplies ranging from 3A to 10A at the common
output voltages of 1.8V, 2.5V, 3.3V, and 5V. Components that
vary, depending upon output current and voltage, are listed
in the accompanying Tables 3 through 6.

Power supplies larger than 10A can also be constructed
using the MIC2182 using larger power-handling compo-
nents.

The "Power Supply Operating Characteristics" graphs follow-
ing the component and vendor tables provide useful informa-
tion about the actual performance of some of these circuits.

MIC2182

Micrel

MIC2182

June 2000

3A (6.5V30V)

4A (6.5V30V)

5A (6.5V30V)

10A (6.5V10V)

Reference

Part No. / Description

qty: 2

TPSE227M010R0100

TPSV227M010R0060

TPSV337M010R0060

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 330

F 10V,

0.1

ESR,

0.1

ESR,

0.06

ESR,

0.06

ESR,

output filter capacitor

C11

qty: 2

qty: 3

qty: 4

TPSE226M035R0300

TPSV107M020R0085

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 100

F 20V,

0.3

ESR,

0.3

ESR,

0.3

ESR,

0.06

ESR,

input filter capacitor

qty: 1 B140, Vishay,

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

qty: 1 CDRH127-100,

qty: 1 CDRH127-100

qty: 1 UP4B-3R3,

Sumida Inductor,

Sumida,

Coiltronics,

H 4A,

H 5A,

3.3

H 11A,

output inductor

qty: 1 Si4800, Siliconix,

qty: 1 Si4884, Siliconix,

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

qty: 1 Si4884, Siliconix,

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

WSL-2010 .020 1%,

WSL-2512 .015 1%,

WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W,

Vishay, 0.02, 1%, 0.5W,

Vishay, 0.015, 1%, 1W,

current sense resistor

MIC2182-5.0BSM or

MIC2182-5.0BM

Table 3. Components for 5V Output

3A (4.5V30V)

4A (4.5V30V)

5A (4.5V30V)

10A (4.5V5.5V)

Reference

Part No. / Description

qty: 2

TPSE227M010R0100

TPSV227M010R0060

TPSV477M006R0055

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 470

F 6.3V,

0.1

ESR,

0.1

ESR,

0.06

ESR,

0.055

ESR,

output filter capacitor

C11

qty: 2

qty: 3

TPSE226M035R0300

TPSV227M016R0075

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 220

F 16V,

0.3

ESR,

0.3

ESR,

0.3

ESR,

0.075

ESR,

input filter capacitor

filter capacitor

qty: 1 B140, Vishay,

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

qty: 1 CDRH127-100,

qty: 1 CDRH127-100

qty: 1 UP4B-3R3,

Sumida Inductor,

Sumida,

Coiltronics,

H 4A,

H 5A,

3.3

H 11A,

output inductor

qty: 1 Si4800, Siliconix,

qty: 2 Si4884, Siliconix,

low-side MOSFET

qty: 1 Si4800, Siliconix,

qty: 1 Si4884, Siliconix,

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

WSL-2010 .020 1%,

WSL-2512 .015 1%,

WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W,

Vishay, 0.02, 1%, 0.5W,

Vishay, 0.015, 1%, 1W,

current sense resistor

MIC2182-3.3BSM or

MIC2182-3.3BM or

MIC2182-3.3BM

MIC2182-3.3BSM

Table 4. Components for 3.3V Output

June 2000

MIC2182

Micrel

3A (4.5V30V)

4A (4.5V30V)

5A (4.5V30V)

10A (4.5V5.5V)

Reference

Part No. / Description

qty: 2

TPSE227M010R0100

TPSV227M010R0060

TPSV447M006R0055

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 470

F 6.3V,

0.1

ESR,

0.1

ESR,

0.06

ESR,

0.06

ESR,

output filter capacitor

C11

qty: 2

qty: 3

TPSE226M035R0300

TPSV227M016R0075

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 220

F 16V,

0.3

ESR,

0.3

ESR,

0.3

ESR,

0.06

ESR,

input filter capacitor

qty: 1 B140, Vishay,

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

qty: 1 CDRH127-100,

qty: 1 CDRH127-100

qty: 1 UP4B-3R3,

Sumida Inductor,

Sumida,

Coiltronics,

H 4A,

H 5A,

3.3

H 11A,

output inductor

qty: 1 Si4800, Siliconix,

qty: 1 Si4884, Siliconix,

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

WSL-2010 .025 1%,

WSL-2010 .020 1%,

WSL-2512 .015 1%,

WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W,

Vishay, 0.02, 1%, 0.5W,

Vishay, 0.015, 1%, 1W,

current sense resistor

MIC2182BSM or

MIC2182BM

Table 5. Components for 2.5V Output

3A (4.5V30V)

4A (4.5V30V)

5A (4.5V8V)

10A (4.5V5.5V)

Reference

Part No. / Description

qty: 2

TPSE227M010R0100

TPSV227M010R0060

TPSV447M006R0055

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 220

F 10V,

AVX, 470

F 6.3V,

0.1

ESR,

0.1

ESR,

0.06

ESR,

0.06

ESR,

output filter capacitor

C11

qty: 2

TPSE226M035R0300

TPSV227M016R0075

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 22

F 35V,

AVX, 220

F 16V,

0.3

ESR,

0.3

ESR,

0.3

ESR,

0.06

ESR,

input filter capacitor

qty: 1 B140, Vishay,

qty: 1 B330, Vishay,

freewheeling diode

qty: 1 CDRH125-100,

qty: 1 CDRH127-100,

qty: 1 CDRH127-100

qty: 1 UP4B-3R3,

Sumida Inductor,

Sumida,

Coiltronics,

H 4A,

H 5A,

3.3

H 11A,

output inductor

qty: 1 Si4800, Siliconix,

qty: 1 Si4884, Siliconix,

qty: 2 Si4884, Siliconix

low-side MOSFET

qty: 1 Si4800, Siliconix,

qty: 2 Si4884, Siliconix,

high-side MOSFET

qty: 1

qty: 2

WSL-2010 .025 1%,

WSL-2010 .020 1%,

WSL-2512 .015 1%,

WSL-2512 .015 1% ,

Vishay, 0.025, 1%, 0.5W,

Vishay, 0.02, 1%, 0.5W,

Vishay, 0.015, 1%, 1W,

current sense resistor

MIC2182BSM or

MIC2182BM

Table 6. Components for 1.8V Output

MIC2182

Micrel

MIC2182

June 2000

Effect of Soft-Start Capacitor (C

) Value

On Output Voltage Waveforms

During Turn-On

(10A Power Supply Configuration)

Effect of Soft-Start Capacitor (C

) Value

On Output Voltage Waveforms

During Turn-On

(4A Power Supply Configuration)

Normal (300kHz Switching Frequency) and

Output Short-Circuit (60kHz) Conditions

Switch Node (Pin 15) Waveforms

Converter Waveforms

QTY: 2
Si4884
HIGH-SIDE
MOSFETS

SWITCH-NODE

VOLTAGE

INDUCTOR CURRENT

LOW-SIDE MOSFET

GATE-TO-SOURCE VOLTAGE

HIGH-SIDE MOSFET

GATE-TO-SOURCE VOLTAGE

QTY: 2
Si4884
LOW-SIDE
MOSFETS

= 7V

L1 = 3.3

OUT

= 3.3V

OUT

= 10A

PIN 15

SW+HSD

PIN 16

-
SIDE

MOSFET

HIGH-SIDE

MOSFET

(2A/div)

HIGH-SIDE

DRIVE VOLTAGE

REFERENCED TO GROUND

10Amps

Typical Skip-Mode Waveforms

OUT

Pin 15

(0.5A/div)

Typical PWM-Mode Waveforms

OUT

Pin 15

(0.5A/div)

Power Supply Operating Characteristics

June 2000

MIC2182

Micrel

-40

-20

100

120

150

180

210

10x10

100x10

1x10

10x10

100x10

300x10

GAIN (dB)

PHASE (

)

FREQUENCY (Hz)

Bode Plot

(4A Power Supply Configuration)

GAIN

PHASE

Load Transient Response

and Bode Plot

(4A Power Supply Configuration)

= 12V

OUT

= 3.3V

L1 = 10

R2 = 20m

OUT

2A/div

OUT

Load Transient Response

and Bode Plot

(10A Power Supply Configuration)

= 6V

OUT

= 3.3V

L1 = 3.3

R2 = 7.5m

OUT

5A/div

OUT

-40

-20

100

120

150

180

210

10x10

100x10

1x10

10x10

100x10

300x10

GAIN (dB)

PHASE (

)

FREQUENCY (Hz)

Bode Plot

(10A Power Supply Configuration)

GAIN

PHASE

100

0.01

0.1

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

R2 = 7.5m

L1 = 3.3

2 high-side MOSFETs: Si4884
2 low-side MOSFETs: Si4884

Efficiency

(10A Power Supply Configuration)

100

0.01

0.1

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

= 5V

R2 = 15m

L1 = 10

1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800

5V Efficiency

(4A Power Supply Configuration)

100

0.01

0.1

EFFICIENCY (%)

OUTPUT CURRENT (A)

12V Efficiency

(4A Power Supply Configuration)

Skip

PWM

= 12V

R2 = 15m

L1 = 10

1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800

100

0.01

0.1

EFFICIENCY (%)

OUTPUT CURRENT (A)

Skip

PWM

= 24V

R2 = 15m

L1 = 10

1 high-side MOSFET: Si4800
1 low-side MOSFET: Si4800

24V Efficiency

(4A Power Supply Configuration)

Ð­Ð»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: MIC2182-5.0

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: MIC2182-5.0