SP8852E

2·7GHz Parallel Load Professional Synthesiser

Preliminary Information

Supersedes January 1996 version, DS4237 - 1.2

DS4237 - 2.0 June 1998

The SP8852E is one of a family of parallel load synthesisers

containing all the elements apart from the loop amplifier to
fabricate a PLL synthesis loop. Other parts in the series are
the SP8854E which has hard wired reference counter pro-
gramming and requires only a single 16-bit programming
word, and the SP8855E which is fully programmable using
hard wired links or switches.

The SP8852E is programmed using a 16-bit parallel data

bus. Data can be stored in one of two internal buffers, selected
by a single address bit on the input interface. In order to fully
program the device, two 16-bit words are required, one to
select the RF division ratio (A and M counters) and phase
detector gain, and one to set the 10-bit reference divider
count, phase detector state and sense. Once the reference
divide ratio has been set, frequency changes can be made by
a single 16-bit data load entry to the RF divider chain.

and F

REF

outputs are reversed by the phase

detector sense bit in the F1/F2 programming word, bit
12. The above diagram is correct when bit 12 is high.

HC44

FEATURES

2·7 GHz Operating Frequency

Single 5V Supply

Low Power Consumption <1·3W

High Comparison Frequency : 20MHz

High Gain Phase Detector : 1mA/rad

Zero `Dead Band' Phase Detector

Wide Range of RF and Reference Division Ratios

Programming by Dual Word Data Transfer

ABSOLUTE MAXIMUM RATINGS

Supply voltage
Operating temperature
Storage temperature
Prescaler and reference input voltage
Data inputs

Junction temperature

0·3V to 16V

C to1100

C to 1150

2·5Vp-p

10·3V

20·3V

175

ORDERING INFORMATION

SP8852E KG HCAR Non-standard temperature range,

C to

100

C, standard product screening

SP8852E IG HCAR Industrial temperature range,

C to

C, standard product screening

THERMAL DATA

= 5

C/W

= 53

C/W

ESD PROTECTION

1000V, human body model

Fig. 1 Pin connections - top view

1 44

B4
B3
B2
B1
B0

0V (PRESCALER)

RF INPUT
RF INPUT

(PRESCALER)

VEE

LOCK DETECT

STROBE
ADDRESS
NC
NC
NC
NC
NC
NC
NC
NC
NC

C-LOCK DETECT

SET

CHARGE PUMP

OUTPUT

CHARGE PUMP

REF

REF OSC CAP

ACIT

REF IN/CR

YST

B10

B12

B13

B14

B15

SP8852E

SP8852E

Pin

1-11, 40-44

13 (RF INPUT)

14 (RF INPUT)

17 (LOCK DETECT INPUT)

18 (C-LOCK DETECT)

19 (R

SET

)

20 (CHARGE PUMP OUTPUT)

21 (CHARGE PUMP REF)

if pin 23 is high

REF

if pin 23 is low

REF

if pin 23 is high

27 (Ref. oscillator capacitor)

28 (REF IN/XTAL)

29-37

38 (ADDRESS)

39 (STROBE)

Description

These are the inputs to the 16-bit data bus. When pin 38 is high the data goes to the buffers for
the A counter, M counter and phase detector gain. When pin 38 is low the data goes to the buffers
for the reference counter and the phase detector state (see Table 4). Open circuit = 1 (high) on
these pins. Data is transparent from pins to the selected buffers when pin 39 (STROBE) is high
and frozen in buffers when pin 39 is low.

Balanced inputs to the RF preamplifier. For single-ended operation the signal is AC-coupled into
pin 13 with pin 14 AC-decoupled to ground (or vice-versa). Pins 13 and 14 are internally DC
biased.

A current sink into this pin is enabled when the lock detect circuit indicates lock. Used to give
an external indication of phase lock.

A capacitor connected to this point determines the lock detect integrator time constant and can
be used to vary the sensitivity of the phase lock indicator.

An external resistor from pin 19 to V

sets the charge pump output current.

The phase detector output is a single ended charge pump sourcing or sinking current to the
inverting input of an external loop filter. The direction is controlled by bit 12 of the reference word.
For bit 12 = 1 and F

or RF phase leads Ref phase pin 20 will sink current (see Table 3).

Connected to the non-inverting input of the loop filter to set the optimum DC bias.

Not Connected.

Not connected.

RF divider output pulses. F

= RF input frequency/(M.N

A). Pulse width = 8 RF input cycles

(1 cycle of the divide by 8 prescaler output).

Reference divider output pulses. F

REF

= reference input frequency/R. Pulse width = high period

of Ref input.

Leave open circuit if an external reference is used. See Fig. 5 for typical connection for use as
an onboard crystal oscillator.

This pin is the input buffer amplifier for an external reference signal. This amplifier provides the
active element if an onboard crystal oscillator is used.

Not connected.

Controls which buffer the data on the input bus goes to. Pin 38 high sends data to the RF divider
group of functions. Pin 38 low sends data to the Ref divider group of functions (see Fig. 6). Open
circuit = high.

When pin 39 is high the A, M, and R counters are held in the reset state and the charge pump
output is disabled. The data on the input bus is loaded into the buffers selected by the ADDRESS
input state (pin 38) when pin 39 goes low. When pin 39 is low the data is fixed in the buffers, the
buffers are loaded into the counter and control register, all the counters are active, and the
charge pump is enabled. Open circuit = high.

Table 1 Pin descriptions

SP8852E

ELECTRICAL CHARACTERISTICS

The Electrical Characteristics are guaranteed over the following range of operating conditions unless otherwise stated

AMB

C to

100

C (KG parts),

C to

C (IG parts); V

= 4·75V to 5·25V

Characteristic

Conditions

Supply current

RF input sensitivity

RF division ratio

Reference division ratio

Comparison frequency

Reference input frequency

Reference input voltage

REF

output voltage high

REF

output voltage low

LOCK DETECT output voltage

CHARGE PUMP current

Input bus logic level high

Input bus logic level low

Input bus current source

Input bus current sink

Up/down current matching

CHARGE PUMP REFERENCE voltage

SET

current

SET

voltage

C-LOCK DETECT current

STROBE pulse width

Data setup time

1·4

2·0

3·4

5·4

3·5

200

1·6

0·5

100

240

16383

1023

100

500

1·7

2·5

4·1

6·5

0·5

180

0·8

1·4

300

1·5

2·3

3·8

6·1

1·6

Pin

18, 26

13,14

13,14, 24

28, 25

28, 24, 25

24, 25

19, 20, 21

1-11, 38-44

Typ.

Max.

Min.

dBm

MHz

dBm

Units

Value

100MHz to 2·7GHz. See note 3.

Ref division ratio >2. See note 1

WRT V

, 2·2k

to 0V

WRT V

, 2·2k

to 0V

OUT

= 3mA

PIN20

= V

PIN21

, I

PIN19

= 1·6mA,

multiplication factor = 1

PIN20

= V

PIN21

, I

PIN19

= 1·6mA,

multiplication factor = 1·5

PIN20

= V

PIN21

, I

PIN19

= 1·6mA,

multiplication factor = 2·5

PIN20

= V

PIN21

, I

PIN19

= 1·6mA,

multiplication factor = 4·0

= 0V

= V

PIN20

= V

PIN21

, I

PIN19

= 1·6mA

PIN19

= 1·6mA, current

multiplication factor = 1·0

PIN19

= 1·6mA, current

multiplication factor = 4·0

Note 2

PIN19

= 1·6mA

PIN18

= 4·7V

Note 3

NOTES
1. Lower frequencies may be used provided that slew rates are maintained.
2. Pin 19 current

multiplication factor must be less than 5mA if charge pump accuracy is to be maintained.

3. Guranteed but not tested.

SP8852E

43 42 41 40

23 24

25 26 27 28

REF IN

REF

10n

2·2k

100p

VCO

LOOP FILTER

30V

SP8852E

33p

100p

10MHZ

Application using
crystal reference

CONTROL

MICRO

ADDRESS

STROBE

VALUES DEPEND
ON APPLICATION

OP27

ETC

Fig. 5 Typical application diagram

DESCRIPTION

Prescaler and

AM counter The programmable divider

chain is of

A and M counter construction and therefore

contains a dual modulus front end prescaler, an

A counter

which controls the dual modulus ratio and an

M counter which

performs the bulk multi-modulus division. A programmable
divider of this construction has a division ratio of

A and

a minimum integer steppable division ratio of

N(N

1), where

N is the prescaler value.

Data Entry and Storage

Data is loaded from the 16-bit bus into one of the internal

buffers by applying a positive pulse to the STROBE input. The
input bus can be driven from TTL or CMOS logic levels. When

STROBE is low, the inputs are isolated and the data can be
changed without affecting the programmed state.

The data is loaded into the RF buffer when the address

input is high and into the reference buffer when low. When the
STROBE input is taken high, the

A and M and reference

counters are reset and the input data is applied to the internal
storage register. When STROBE is again taken low, the data
on the input bus is stored in the selected register and the
counters released. The STROBE input is level triggered so
that if the data is changed whilst the input is high, the final
value before STROBE goes low will be stored.

In order to prevent disturbances on the VCO control voltage

when frequency changes are made, the STROBE input disables

SP8852E

the charge pump outputs when high. During this period the
VCO control voltage will be maintained by the loop filter
components around the loop amplifier, but due to the com-
bined effects of the amplifier input current and charge pump
leakage a gradual change will occur. In order to reduce the
change, the duration of the strobe pulse should be minimised.
Selection of a loop amplifier with low input current will reduce
the VCO voltage droop during the strobe pulse and result in
minimum reference sidebands from the synthesiser.

Reference Input

The reference source can be either driven from an external

sine or square wave source of up to 100MHz or a crystal can
be connected as shown in Fig. 5.

Phase Comparator and Charge Pump

The SP8852E has a digital phase/frequency comparator

driving a charge pump with programmable current output.
The charge pump current level at the minimum gain setting is
approximately equal to the current fed into the R

SET

input, pin

19, and can be increased by programming the bus according
to Table 2 by up to 4 times.

1·6V

SET

Pin 19 current =

Phase detector gain =

PIN19

(mA)

multiplication factor

mA/rad

Sense bit (bit 12)

Pin 20

Current source

Current sink

Output for RF phase lag

Table 3

Bit 11

Bit 10

Phase detector state

Enabled, F

and F

REF

off

Enabled, F

and F

REF

Disabled by lock detect, F

and F

REF

Disabled, F

and F

REF

The charge pump connections to the loop amplifier consist

of the charge pump output and the charge pump reference.
The matching of the charge pump up and down currents will
only be maintained if the charge pump output is held at a
voltage equal to the charge pump reference using an
operational amplifier to produce a virtual earth condition at pin
20. The lock detect circuit can drive an LED to give visual
indication of phase lock or provide an indication to the control
system if a pullup resistor is used in place of the LED. A small
capacitor connected form the C-LOCK DETECTOR pin to
ground may be used to delay lock detect indication and
remove glitches produced by momentary phase coincidence
during lock up.

To allow for control direction changes introduced by the

design of the PLL, bit 12 on the input bus address 0 can be
programmed to reverse the sense of the phase detector by
transposing the F

and F

REF

connections. In order that any

external phase detector will also be reversed by this program-
ming bit, the F

and F

REF

outputs are also interchanged by

bit 12 as shown in Table 3.

ADDRESS

PIN 40
BIT 15

PIN 11

BIT 0

NOT USED

PHASE

DETECTOR

STATE

CONTROL

(SEE TABLE 4)

PHASE

DETECTOR

SENSE

CONTROL

(SEE TABLE 3)

10-BIT REFERENCE COUNTER

Fig. 6a Reference word bit allocation

ADDRESS

PHASE

DETECTOR

GAIN

CONTROL

(SEE TABLE 2)

M COUNTER

3-BIT

A COUNTER

Fig. 6b RF division ratio bit allocation

PIN 40
BIT 15

PIN 11

BIT 0

Fig. 6 Programming data format

Bit 15

Bit 14

Current multiplication factor

1·0

1·5

2·5

4·0

Table 2

Table 4

The F

and F

REF

signals to the phase detector are

available on pins 24 and 25 and may be used to monitor the
frequency input to the phase detector or used in conjunction
with an external phase detector. These outputs may be
programmed by bits 10 and 11 of word 0 according to Table 4.
State 3, where the outputs are disabled by the lock detect
circuit, is useful where the user wishes to use an external
phase detector. The internal phase/frequency detector may
be used to pull the loop into lock and an automatic switch-over
to the external phase detector made. When the F

and F

REF

outputs are to be used at high frequencies, an external pull
down resistor of minimum value 330

may be connected to

ground to reduce the fall time of the output pulse.

SP8852E

Fig. 7g F

and F

REF

outputs

Fig. 7h Reference oscillator

PD,

REF

OUTPUTS

24, 25

296

3·3mA

100

60k

40k

CAPACITOR

CRYSTAL

Fig. 7 Interface circuit diagrams (continued)

APPLICATIONS

RF Layout

The SP8852E can operate with input frequencies up to

2·7GHz but to obtain optimum performance, good RF layout
practices should be used. A suitable layout technique is to use
double sided printed circuit board with through plated holes.
Wherever possible the top surface on which the SP8852E is
mounted should be left as a continuous sheet of copper to
form a low impedance ground plane. The ground pins 12 and
16 should be connected directly to the ground plane.

Pins such as V

and the unused RF input should be

decoupled with chip capacitors mounted as close to the
device pin as possible, with a direct connection to the ground
plane; suitable values are 10nF for the power supplies and
<1nF for the RF input pin (a lower value should be used
sufficient to give good decoupling at the RF frequency of
operation). A larger decoupling capacitor mounted as close
as possible to pin 26 should be used to prevent modulation of
V

by the charge pump pulses. The R

SET

resistor should also

be mounted close to the R

SET

pin to prevent noise pickup. The

capacitor connected from the charge pump output should be
a chip component with short connections to the SP8852E. All
signals such as the programming inputs, RF IN, REFERENCE
IN and the connections to the op-amp are best taken through
the pc board adjacent to the SP8852D with through plated
holes allowing connections to remote points without
fragmenting the ground plane.

Lock Detect Circuit

The lock detect circuit uses the up and down correction

pulses from the phase detector to determine whether the loop
is in or out of lock. When the loop is locked, both up and down
pulses are very narrow compared to the reference frequency,
but the pulse width in the out of lock condition continuously
varies, depending on the phase difference between the outputs
of the reference and RF counters. The logical AND of the up
and down pulses is used to switch a 20mA current sink to pin
18 and a 50k

resistor provides a load to V

. The circuit is

shown in Fig. 7c.

When lock is established, the narrow pulses from the

phase detector ensure that the current source is off for the
majority of the time and so pin 18 will be pulled high by the
50k

resistor. A voltage comparator with a switching threshold

at about 4·7V monitors the voltage at pin 18 and switches pin
17 low when pin 18 is more positive than the 4·7V threshold.

When the loop is unlocked, the frequency difference at the

counter outputs will produce a cyclic change in pulse width
from the phase detector outputs with a frequency equal to the
difference at the reference and RF counter outputs. A small
capacitor connected to pin 18 prevents the indication of false
phase lock conditions at pin 17 for momentary phase
coincidence. Because of the variable width pulse nature of the
signal at pin 18 the calculation of a suitable capacitor value is
complex, but if an indication with a delay amounting to several
times the expected lock up time is acceptable, the delay will
be approximately equal to the time constant of the capacitor
on pin 18 and the internal 50k

resistor. If a faster indication

is required, comparable with the loop lock up time, the
capacitor will need to be 2 to 3 times smaller than the time
constant calculation suggests. The time to respond to an out
of lock condition is 2 to 3 times less than that required to
indicate lock.

Charge Pump Circuit

The charge pump circuit converts the variable width up and

down pulses from the phase detector into adjustable current
pulses which can be directly connected to the loop amplifier.
The magnitude of the current and therefore the phase detec-
tor gain can be modified when new frequency data is entered
to compensate for change in the VCO gain characteristic over

Programming Bus

The input pins are designed to be compatible with TTL or

CMOS logic with a switching threshold set at about 2·4V by
three forward biased base-emitter diodes. The inputs will be
taken high by an internal pull up resistor if left open circuit but
for best noise immunity it is better to connect unused inputs
directly to V

or ground.

RF Inputs

The prescaler has a differential input amplifier to improve

input sensitivity. Generally the input drive will be single ended
and the RF signal should be AC coupled to either of the inputs
using a chip capacitor.The remaining input should be decoupled
to ground, again using a chip capacitor. The inputs can be driven
differentially but the input circuit should not provide a DC path
between inputs or to ground.

SP8852E

its frequency band. The charge pump pulse current is determined
by the current fed into pin 19 and is approximately equal to pin
19 current when the programmed multiplication ratio is 1. The
circuit diagram Fig. 7e shows the internal components on pin 19
which mirror the input current into the charge pump. The voltage
at pin 19 will be approximately 1·6V above ground due to two V

drops in the current mirror. This voltage will exhibit a negative
temperature coefficient, causing the charge pump current to
change with chip temperature by up to 10% over the full military
temperature range if the current programming resistor is
connected to V

as shown in the application diagram, Fig. 5. In

critical applications where this change in charge pump current
would be too large the resistor to pin 19 could be increased in
value and connected to a higher supply to reduce the effect of V

variation on the current level. A suitable resistor connected to a
30V supply would reduce the variation in pin 19 current due to
temperature to less than 1·5%. Alternatively a stable current
source could be used to set pin 19 current.

The charge pump output on pin 20 will only produce

symmetrical up and down currents if the voltage is equal to that
on the voltage reference pin 21. In order to ensure that this
voltage relationship is maintained, an operational amplifier must
be used as shown in the typical application Fig. 5. Using this
configuration pin 20 voltage will be forced to be equal to that on
pin 21 since the operational amplifier differential input voltage will
be no more than a few millivolts (the input offset voltage of the
amplifier).

When the synthesiser is first switched on or when a frequency

outside the VCO range is programmed, the amplifier output will
limit, allowing pin 20 voltage to differ from that on pin 21. As soon
as an achievable frequency value is programmed and the
amplifier output starts to slew the correct voltage relationship
between pin 20 and 21 will be restored. Because of the importance
of voltage equality between the charge pump reference and
output pins, a resistor should never be connected in series with
the operational amplifier inverting input and pin 20, as is the case
with a phase detector giving voltage outputs. Any current drawn
from the charge pump reference pin should be limited to the few
microamps input current of a typical operational amplifier. A
resistor between the charge pump reference and the non-
inverting input could be added to provide isolation but the value
should not be so high that more than a few millivolts drop are
produced by the amplifier input current.

When selecting a suitable amplifier for the loop filter, a

number of parameters are important; input offset voltage in
most designs is only a few millivolts and an offset of 5mV will
produce a mismatch in the up and down currents of about 4%
with the charge pump multiplication factor set at 1. The
mismatch in up and down currents caused by input offset
voltage will be reduced in proportion to the charge pump
multiplication factor in use.

If the linearity of the phase detector about the normal phase

locked operating point is critical, the input offset voltage of most
amplifiers can be adjusted to near zero by means of a
potentiometer. The charge pump reference voltage on pin 21 is
about 1·3V below the positive supply and will change with
temperature and with the programmed charge pump multiplication
factor. In many cases it is convenient to operate the amplifier with
the negative power supply pin connected to 0V as this removes
the need for an additional power supply. The amplifier selected
must have a common mode range to within 3·4V (minimum
charge pump reference voltage) of the negative supply pin to
operate correctly without a negative supply. Most popular
amplifiers can be operated from a 30V positive supply to give a
wide VCO voltage drive range and have adequate common

mode range to operate with inputs at

3·4V with respect to the

negative supply.

Input bias and offset current levels to most operational

amplifiers are unlikely to be high enough to significantly affect the
accuracy of the charge pump circuit currents but the bias current
can be important in reducing reference side bands and local
oscillator drift during frequency changes.

When the loop is locked, the charge pump produces only very

narrow pulses of sufficient width to make up for any charge lost
from the loop filter components during the reference cycle. The
charge lost will be due to leakage from the charge pump output
pin and to the amplifier input bias current, the latter usually being
more significant. The result of the lost charge is a sawtooth ripple
on the VCO control line which frequency modulates the phase
locked oscillator at the reference frequency and its harmonics. A
similar effect will occur whenever the strobe input is taken high
during a programming sequence. In this case the charge pump
is disabled when the strobe input is high and any leakage current
will cause the oscillator to drift off frequency. To reduce this
effect, the duration of the strobe pulse should be minimised.

and F

REF

Outputs

These outputs provide access to the outputs from the RF and

reference dividers and are provided for monitoring purposes
during product development or test, and for connection of an
external phase detector if required. The output circuit is of ECL
type, the circuit diagram being shown in Fig. 7g. The outputs can
be enabled or disabled under software control by the address 0
control word but are best left in the disabled state when not
required as the fast edge speeds on the output can increase the
level of reference sidebands on the synthesised oscillator.

The emitter follower outputs have no internal pulldown resis-

tor to save current and if the outputs are required an external
pulldown resistor should be fitted. The value should be kept as
high as possible to reduce supply current, about 2·2k

being

suitable for monitoring with a high impedance oscilloscope probe
or for driving an AC-coupled 50

load. A minimum value for the

pulldown resistor is 330

When the F

and F

REF

outputs are disabled the output level

will be at the logic low level of about 3·5V so that the additional
supply current due to the load resistors will be present even when
the outputs are disabled.

Reference Input

The reference input circuit functions as an input amplifier or

crystal oscillator. When an external reference signal is used this
is simply AC-coupled to pin 28, the base of the input emitter
follower. When a low phase noise synthesiser is required the
reference signal is critical since any noise present here will be
multiplied by the loop. To obtain the lowest possible phase noise
from the SP8852E it is best to use the highest possible reference
input frequency and to divide this down internally to obtain the
required frequency at the phase detector. The amplitude of the
reference input is also important, and a level close to the
maximum will give the lowest noise.

When the use of a low reference input frequency say 4 to

10MHz is essential some advantage may be gained by using a
limiting amplifier such as a CMOS gate to square up the
reference input. In cases where a suitable reference signal is not
available, it may be more convenient to use the input buffer as a
crystal oscillator in this case the emitter follower input transistor
is connected as a Colpitts oscillator with the crystal connected
from the base to ground and with the feedback necessary for
oscillation provided by a capacitor tap at the emitter. The
arrangement is shown inset in Fig. 5.

SP8852E

FROM

CHARGE PUMP

TO VCO

FROM

CHARGE PUMP

REFERENCE

Fig. 8 Third order loop filter circuit diagram

Loop Filter Design

Generally, the third order filter configuration shown in Fig. 8

gives better results than the more commonly used second order
because the reference sidebands are reduced. Three equations
are required to determine values for the three constants, where

The equations are:

(

)

...(2)

...(3)

...(1)

tan F

1
2

cos

where
K

is the phase detector gain factor in mA/radian

is the VCO gain factor in radians/seconds/V

is the division ratio from VCO to reference frequency

is the natural loop frequency

is the phase margin, normally set to 45

Since the phase detector used is linear over a range of 2

radians, the phase detector gain is given by:

These values can now be substituted in equation (1) to obtain
a value for

and in equations (2) and (3) to determine values

for

and

Example

Calculate values for a loop with the following parameters:

Frequency to be synthesised

1000MHz

Reference frequency

10MHz

Division ratio

1000MHz/100MHz = 100

VCO gain factor

10MHz/V

phase margin

Phase comparator current

6·3mA

The phase detector gain factor

= 6·3/2p = 1mA/radian

mA/radian

Phase comparator current setting

From equation (3):

tan

100kHz

cos 45

= 659

628319

0·4142

(100kHz

)

659

= 3·844

Using these values in equation (1):

100

(100kHz

)

10MHz/V

[A]

where A =

1
2

(100kHz

)

(3·844

)

(100kHz

)

(659

)

= 3·84

From equation (2):

= 1·59

2·415

39·48

62832

6·833

1·1714

1
2

Substituting for C2:

3·844

659

0·0153

= 829·4

659

829·4

= 0·794nF

Now,

= 3·84nF

(

)

or,

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piece of equipment. It is the user's responsibility to fully determine the performance and suitability of any equipment using such information and to ensure that any publication or
data used is up to date and has not been superseded. Manufacturing does not necessarily include testing of all functions or parameters. These products are not suitable for use in
any medical products whose failure to perform may result in significant injury or death to the user. All products and materials are sold and services provided subject to Mitel's
conditions of sale which are available on request.

Ð­Ð»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: SP8852

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: SP8852