Semiconductor Components Industries, LLC, 2005

January, 2005 - Rev. 2

Publication Order Number

NCP1239/D

NCP1239

Low-Standby High
Performance PWM Controller

Housed in SO-16 the NCP1239 represents a major leap toward

ultra-compact Switch Mode Power Supplies specifically tailored for
medium to high power off-line applications, e.g. notebook adapters.
The NCP1239 offers everything needed to build a rugged and efficient
power supply, including a dedicated event management to drive a
Power Factor Correction (PFC) front-end circuitry. The circuit
disables the front-end PFC stage while still in fault or standby
conditions by interrupting the PFC controller powering for improved
no-load consumption figures. As soon as normal operating mode
recovers, the NCP1239 feeds back the PFC that wakes-up.

When power demand is low, the IC automatically enters the

so-called skip-cycle mode and provides excellent efficiency at light
loads. Because this occurs at a user adjustable low peak current, no
acoustic noise takes place.

Features

Current-Mode Operation with Internal Ramp Compensation

Internal High-Voltage Current Source for loss-less Startup

Adjustable Skip-Cycle Capability

Selectable Soft-Start Period

Internal Frequency Dithering for Improved EMI Signature

Go-to-Standby Signal for PFC Front-Stage

Large V

Operation from 12.2 V to 36 V

500 mV Overcurrent Limit

500 mA/-800 mA Peak Current Capability

5 V/10 mA Pinned-out Reference Voltage

Adjustable Switching Frequency up to 250 kHz.

Overload Protection Independent of the Auxiliary V

Adjustable Over Power Compensation (NCP1239F)

Programmable Maximum Duty Cycle (NCP1239V)

Typical Applications

High Power AC/DC Adapters for Notebooks etc.

Offline Battery Chargers

Telecom and PC Power Supplies

Flyback Applications (NCP1239F) and Forward Applications

(NCP1239V)

Device

Package

Shipping

ORDERING INFORMATION

NCP1239FDR2

SO-16

2500/Tape & Reel

NCP1239 = Device Code
x

= F or V

= Assembly Location

= Wafer Lot

= Year

= Work Week

MARKING
DIAGRAM

SO-16

FD or VD SUFFIX

CASE 751B

NCP1239xD

AWLYWW

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PIN CONNECTIONS

Over Power
Limit

Skip Adjust

Gnd

SS/Timer

Drv

Brown-out

Fault Detect

REF5V

GTS

For information on tape and reel specifications,

including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.

NCP1239VDR2

SO-16

2500/Tape & Reel

Max Duty-
Cycle

Skip Adjust

Gnd

SS/Timer

Drv

Brown-out

Fault Detect

REF5V

GTS

NCP1239F

NCP1239V

NCP1239

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ELECTRICAL CHARACTERISTICS

(For typical values T

= 25

C, for min/max values T

= 0

C to +125

C, Max T

= 150

= 20 V unless otherwise noted.)

Symbol

Rating

Pin

Min

Typ

Max

Unit

Supply Section

CCON

Turn-on Threshold Level, V

Going up

15.5

16.4

17.5

CCOFF

Minimum Operating Voltage after Turn-on

10.5

11.2

12.2

HYST1

Difference (V

CCON

- V

CCOFF

)

4.5

5.1

CCLATCH

Decreasing Level at which the Latch-off Phase ends

6.5

6.9

7.2

CCRESET

Level at which the Internal Logic gets reset

4.0

CC1

Internal IC Consumption, no output load on Pin 12 (@I

= 20

NCP1239F

NCP1239V

-
-

2.1
2.6

3.0
4.0

CC2a

Internal IC Consumption, 1 nF output load on Pin 12

NCP1239F (65 kHz)

NCP1239V (118 kHz)

-
-

3.1
4.2

3.8
6.5

CC2b

Internal IC Consumption, 1 nF output load on Pin 12

NCP1239F (100 kHz)

NCP1239V (182 kHz)

-
-

3.9
5.5

5.0
8.5

CC2c

Internal IC Consumption, 1 nF output load on Pin 12

NCP1239F (130 kHz)

NCP1239V (236 kHz)

-
-

4.6
6.7

5.9
9.6

CC3

Internal IC Consumption, latchoff phase

(NCP1239F and NCP1239V)

0.40

0.75

Internal Startup Current Source

C1_hv

High-Voltage Current Source (sunk by Pin 16), V

= 10 V

2.0

4.0

5.3

C1_VCC

Startup Charge Current flowing out of the V

Pin, V

=10 V

1.8

3.6

4.5

High-Voltage Current Source, V

= 0

4.2

5 V Reference Voltage (REF5V)

REF5V

Reference Voltage

@ No load on Pin 2
@ I

pin2

= 5 mA

4.7
4.6

5.0
4.9

5.2
5.1

Iref

Current Capability

5.0

Drive Output

Vcl

Output Voltage Positive Clamp

11.5

13.6

rise

Output Voltage Rise-Time @ CL = 1 nF, 10-90% of output signal

fall

Output Voltage Fall-Time @ CL = 1 nF, 10-90% of output signal

source

High State Voltage Drop @ I

pin12

= 3 mA and V

= 12 V

2.5

3.3

source

Source Current Capability (@ V

pin12

= 0 V)

500

Sink Resistance @ V

pin12

=1 V

3.8

7.5

sink

Sink Current Capability (@ V

pin12

= 10 V)

800

Oscillator

fsw

Recommended Switching Frequency Range

250

kHz

Vosc

Pin 4 Voltage @ Rt = 100 k

1.6

Kosc

Product (Switching Frequency times the Rt Pin 4 resistance) (Note 1)

@ 65 kHz and 130 kHz (NCP1239F)
@ 118 kHz and 236 kHz (NCP1239V)

6050

11000

6500

11800

6950

12600

kHz*k

fsw

Internal Modulation Swing, in percentage of fsw

3.5

Dmax

Maximum Duty-Cycle

75.5

80.0

83.0

1. The nominal switching frequency fsw equals: fsw = K

OSC

/Rt. The implemented jittering makes the switching frequency continuously vary

around this nominal value (

3.5% variation).

NCP1239

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ELECTRICAL CHARACTERISTICS

(For typical values T

= 25

C, for min/max values T

= 0

C to +125

C, Max T

= 150

= 20 V unless otherwise noted.)

Symbol

Unit

Max

Typ

Min

Pin

Rating

Current Limitation

Limit

Maximum Internal Set-Point

0.84

0.90

0.95

DEL_CS

Propagation Delay from V

pin10

> I

Limit

to gate turned off

(Pin 12 loaded by 1 nF)

130

220

LEB-65kHz

Leading Edge Blanking Duration (Pins 9 and 10) @ 65 kHz

(NCP1239F)

9, 10

420

LEB-130kHz

Leading Edge Blanking Duration (Pins 9 and 10) @ 130 kHz

(NCP1239F)

9, 10

230

LEB-118kHz

Leading Edge Blanking Duration (Pin 10) @ 118 kHz (NCP1239V)

320

LEB-236kHz

Leading Edge Blanking Duration (Pin 10) @ 236 kHz (NCP1239V)

170

Over Power Limit (NCP1239F)

ocp

Internal Current Source of the Over Power Limit Pin

@ 1 V on Pin 5 and V

pin9

= 0.5 V

@ 2 V on Pin 5 and V

pin9

= 0.5 V

120

160

100
185

opl

Over Power Limitation Threshold

@ T

= 25

@ T

= 0

C to 125

0.48
0.47

0.50
0.50

0.52
0.52

DEL_OCP

Propagation Delay from V

pin9

> V

opl

to gate turned off

(Pin 12 loaded by 1 nF)

130

220

Maximum Duty-Cycle (Dmax) Control (NCP1239V)

Dmax

Pin 9 Current Source @ V

pin9

= 1.0 V and V

pin9

= 2.0 V

max

Maximum Duty Cycle @ 118 kHz and V

pin9

= 1.0 V

Dmax

Dmax Coefficient @ 118 kHz and V

pin9

= 1.0 V (Note 2)

1.10

1.30

1.53

%/k

Soft-Start and Timer

Soft-Start or Jittering charge current @ V

pin6

= 2.4 V

110

disch

Jittering Discharge Current @ V

pin6

= 2.4 V

107

137

jitter

Jittering Saw-Tooth Lower Threshold

1.67

1.80

1.89

jitter

Jittering Saw-Tooth Upper Threshold

2.85

3.00

3.20

timer

Timer Peak Threshold

4.0

4.3

4.6

timerC

Timer Charge Current @ V

pin6

= 3.5 V and Pin 8 open

3.9

5.2

6.4

timerD

Timer Discharge Current @ V

pin6

= 3.5 V and Pin 8 open

400

Feedback Section

Rup

Internal Pullup Resistor

Ifb

Source Current @ V

pin8

= 0.5 V

200

Iratio

Pin 8 to current Setpoint division ratio

3.0

Internal Ramp Compensation

ramp

Internal Resistor

ramp

Internal Saw-Tooth Amplitude

3.2

2. K

Dmax

is the proportionality coefficient that links the maximum duty-cycle to the Pin 9 resistor: Dmax = K

Dmax

pin9

. K

Dmax

is defined in

the "Maximum Duty-Cycle Limitation" section of the operating description.

NCP1239

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PIN FUNCTION DESCRIPTION

Pin No.

Pin Name

Function

Pin Description

GTS

Shuts the PFC down in

standby

The standby detection block changes Pin 1 state in accordance to the mode
(standby or normal mode). Pin1 is designed to drive an external pnp transistor that
connects or disconnects the NCP1239's V

to the PFC's.

REF5V

A 5V reference voltage

This pin helps to internally bias the controller but can also be used to power
surrounding logic gates for any purposes. The typical output current is 10 mA. This
voltage source is disabled during the circuit startup and latched-off phases. A
100 nF filtering capacitor must be placed between Pin 2 and ground.

Fault Detect

Enables to permanently

shutdown the part

If the Pin 3 voltage exceeds 2.4 V, the circuit is permanently shut down. This pin
can be used to monitor the voltage across a thermistor in order to protect the
application from excessive heating and/or to detect an overvoltage condition.

Timing resistor

Pin4 resistor allows a precise frequency programming from 20 kHz up to 250 kHz.

Brown-Out

This pin receives a portion of the bulk capacitor to authorize operation above a
certain level of mains only. It also serves to elaborate an offset voltage on Pin 9
used for Over Power Compensation.

SS/Timer

Performs soft-start and

fault timeout

During Power on and fault conditions, the capacitor connected to this pin ensures a
soft-start period. When a fault is detected, this pin is internally brought high by a
current source. If 4.3 V are reached, the fault is confirmed and the circuit enters an
auto-recovery burst mode, otherwise the pin goes back to a lower value and
oscillates to perform frequency jittering.

Skip Adjust

Adjust skip level

By adjusting the skip-cycle level, it is possible to fight against noisy transformers
and modify the standby detection thresholds. Keep Pin 7 open to operate with the
default levels (skip threshold setpoint: 140 mV, normal mode recovery setpoint:
250 mV).

Feedback signal

An opto-coupler collector pulls this pin low to regulate

Over Power

Limit

(NCP1239F)

Enables a precise peak

current clamp and then an

accurate Over Power

Detection

This pin delivers a current proportional to V

pin5

, an image of the high voltage rail.

Inserting a resistor between Pin 9 and the current sense resistor, an offset
proportional to the input voltage is built. Such offset compensates the circuit and
power switch propagation delays for an accurate power limitation in the whole input
voltage range.

Max Duty-

Cycle

(NCP1239V)

Enables to precisely
clamp the maximum

duty-cycle.

This terminal sources a constant current. Connect a resistor between Pin 9 and
Ground to select the maximum duty-cycle.

The current sense input

This pin receives the primary current information via a sense element. By inserting
a resistor in series with this pin, it becomes possible to introduce ramp
compensation.

Ground

The IC ground

Drv

Drives the MOSFET

By offering up to +500 mA/-800 mA peak, this pin lets you drive large Qg
MOSFET's. It is clamped to 16 V maximum not to exceed the maximum
gate-source voltage of most power MOSFET's.

Supplies the controller

This pin accepts up to 36 V from an auxiliary winding.

Creepage distance.

The high-voltage startup

This pin connects to the bulk capacitor to generate the startup current.

NCP1239

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Figure 3. NCP1239F Internal Circuit Architecture

Vcc < 4V

450mV

0.5V / 0.25V

REF5V

SS / timer

PFC_Vcc

Fault
detect

Skip
adjust

Vdd

/ 3

to Skip

20k

0.9V

Oscillator

Gnd

32k

Drv

Vcc

Vdd

Regul

UVLOs
Latch
Reset

Error flag

Over Power
Limit

Vdd

2.5V

Vdd

2.5V

100k

Fault

Vdd

Soft-Start
and timer
management

Stby_detect

Error_Flag

Stby

OVL

Vcc<7V

stdwn

Vstop

PWM Latch

Output
Buffer

BO_out

Jittering
Modulation

"Jittered"
Reference

Jittering
Modulation

CLK

0.5V

BO_in

Soft-Start
Ipk limit

Internal
Thermal
Shutdown

TSD

Divider by 2

Skip

Startup Phase

(Vcc<VccOFF)

1mA

Vcc

10k

Vstop

regOUT

Stby_detect

25r

15r

FB<Vpin1 => Skip high

FB>1.6*Vpin1 =>Stby_detect RESET

pfcOFF

pfcON

UVLO

14V

clamp

pfcON

OUTon

Ramp
Compensation

3.2V

BO_in

A/V x V

pin5

LEB

NCP1239

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Figure 4. NCP1239V Internal Circuit Architecture

Vcc < 4V

450mV

0.5V / 0.25V

REF5V

SS / timer

PFC_Vcc

Fault
detect

Skip
adjust

Vdd

/ 3

to Skip

20k

0.9V

Oscillator

Gnd

32k

Drv

Vcc

Vdd

Regul

UVLOs
Latch
Reset

Error flag

Dmax

Vdd

2.5V

Vdd

2.5V

100k

Fault

Vdd

Soft-Start
and timer
management

Stby_detect

Error_Flag

Stby

OVL

Vcc<7V

stdwn

Vstop

PWM Latch

Output
Buffer

BO_out

Jittering
Modulation

"Jittered"
Reference

Jittering
Modulation

CLK

BO_in

Soft-Start
Ipk limit

Internal
Thermal
Shutdown

TSD

Divider by 2

Skip

Startup Phase

(Vcc<VccOFF)

1mA

Vcc

10k

Vstop

regOUT

Stby_detect

25r

15r

FB<Vpin1 => Skip high

FB>1.6*Vpin1 =>Stby_detect RESET

pfcOFF

pfcON

UVLO

14V

clamp

pfcON

OUTon

Ramp
Compensation

3.2V

Idmax

LEB

OSC

NCP1239

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Figure 5. High Voltage Current Source

vs. Temperature @ V

= 10 V

TYPICAL PERFORMANCE CHARACTERISTICS

Figure 6. High Voltage Current Source

vs. Temperature @ V

= 0 V

Figure 7. High Voltage Pin Leakage Current

vs. Temperature

Figure 8. V

Startup Threshold

vs. Temperature

Figure 9. V

Turn-Off Threshold

vs. Temperature

Figure 10. V

Latched-Off vs. Temperature

TEMPERATURE (

125

100

-25

(mA)

TEMPERATURE (

125

100

5.0

125

100

TEMPERATURE (

125

100

TEMPERATURE (

11.00

11.10

11.20

11.30

11.40

125

100

-25

6.82

6.84

6.86

6.88

C1_HV

, (mA)

Pin16 Leakage Current (

CCON

(V)

TEMPERATURE (

125

100

-25

CCOFF

(V)

CCLA

TCH

(V)

6.0

10.80

10.90

6.90

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

5.0

4.0

3.0

2.0

1.0

16.25

16.30

16.35

16.40

16.45

16.50

16.55

NCP1239

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TEMPERATURE (

125

100

-25

1.60

2.20

2.80

125

100

-25

2.0

2.5

3.0

3.5

4.0

TEMPERATURE (

0.20

0.30

0.35

0.40

0.45

0.50

125

100

-25

2.35

2.40

2.45

2.55

2.60

2.0

3.0

7.0

CC1

(mA)

CC2

(mA)

TEMPERATURE (

125

100

-25

CC3

(mA)

Vdrop (V)

TEMPERATURE (

125

100

-25

SINK (

)

2.50

1.0

5.0

6.0

4.0

TYPICAL PERFORMANCE CHARACTERISTICS

Figure 11. No Load Circuit Consumption

vs. Temperature

Figure 12. Circuit Consumption

(1 nF on driver Pin 12) vs. Temperature

Figure 13. Latched-Off Mode Consumption

vs. Temperature

Figure 14. REF5V Voltage Source

vs. Temperature

Figure 15. Driver High State Voltage Drop

vs. Temperature

Figure 16. Driver Sink Resistance

vs. Temperature

0.25

4.70

4.85

4.90

4.95

TEMPERATURE (

125

100

-25

130 kHz

REF5V (V)

4.80

0 mA

5 mA

10 mA

1.80

2.00

2.40

2.60

4.5

5.0

100 kHz

65 kHz

0.55

0.60

5.00

5.05

2.70

2.75

2.65

4.75

NCP1239

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11250

11350

11450

11550

11650

11750

11850

11950

12050

12150

12250

TEMPERATURE (

125

100

-25

125

100

-25

6420

6460

6500

6540

6580

Dmax (%)

osc

(kHz*k

)

125

130 kHz

TYPICAL PERFORMANCE CHARACTERISTICS

Figure 17. Driver Voltage Clamp vs. Temperature

Figure 18. Maximum Duty Cycle vs. Temperature

(NCP1239F)

Figure 19. Oscillator K

osc

Parameter vs. Temperature

osc

= fsw * R

pin4

) (NCP1239F)

TEMPERATURE (

125

100

-25

12.5

13.5

14.0

14.5

15.0

Vcl CLAMP VOL

AGE (V)

13.0

6440

6480

6520

6560

200 kHz

65 kHz

TEMPERATURE (

125

100

osc

(kHz*k

)

osc1

@ 130 kHz

Figure 20. Oscillator K

osc

Parameter vs. Temperature

osc

= fsw * R

pin4

) (NCP1239V)

osc2

@ 260 kHz

TEMPERATURE (

120

140

180

125

100

-25

0.47

0.48

0.49

0.51

0.53

TEMPERATURE (

100

-25

ocp

(

opl

(V) 0.50

100

Figure 21. Pin 9 Current vs. Temperature

(@ V

pin9

= 0.5 V) (NCP1239F)

Figure 22. Over Power Limitation Threshold vs.

Temperature (NCP1239F)

160

BO=2 V

BO=1 V

0.52

NCP1239

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TYPICAL PERFORMANCE CHARACTERISTICS

0.492

0.494

TEMPERATURE (

125

100

-25

BO_th_H (V)

0.493

0.495

0.496

0.497

TEMPERATURE (

125

100

-25

410

415

420

430

435

skip

(mV)

425

TEMPERATURE (

125

100

-25

710

720

730

750

760

stby-

out

(mV)

740

445

450

440

770

0.490

0.491

TEMPERATURE (

125

100

-25

0.87

0.89

0.91

0.93

Limit

(V)

0.88

0.90

0.92

Figure 23. Maximum Current Setpoint vs.

Temperature

Figure 24. Default Feedback Threshold for

Standby Detection vs. Temperature

Figure 25. Default Feedback Level for Normal

Operation Recovery

Figure 26. Brown-Out Upper Threshold vs.

Temperature

TEMPERATURE (

125

100

-25

0.240

0.244

0.250

(V)

0.242

0.246

0.248

TEMPERATURE (

125

100

-25

2.46

2.48

fault

(V)

2.40

2.44

2.42

2.36

2.38

Figure 27. Brown-Out Low Threshold vs.

Temperature

Figure 28. Fault Detect Threshold vs. Temperature

NCP1239

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APPLICATION INFORMATION

The NCP1239 includes all necessary features to help

building a rugged and safe switch-mode power supply. The
following details the major benefits brought by
implementing the NCP1239 controller:

Current-mode operation with internal ramp

compensation: implementing peak current mode control,
the NCP1239 offers an internal ramp compensation signal
that can easily be summed up to the sensed current.
Subharmonic oscillations can thus be fought via the
inclusion of a simple resistor,

500 mV Current Sense threshold for Over Power Limit

(NCP1239F): the NCP1239 operating in current mode, the
circuit Pin 10 monitors the current to modulate its level
according to the power demand. Due to the ramp
compensation, one must generally note that the Pin 10
voltage is not the exact image of the inductor current. A
precise current limitation being essential, the NCP1239
features a separate current sense pin (Pin 9) for an accurate
overcurrent detection. The low threshold of this protection
(500 mV) avoids excessive losses in the current sense
resistor and improves the efficiency. In addition, Pin 9
sources a current that proportional to the high-voltage rail,
compensates the current-sense and turn off delays at high
line. A resistor inserted between Pin 9 and the sensing
resistor offsets the Pin 9 current-sense information to build
a precise overload protection, independent of the mains
input.

Large V

operation: the NCP1239 offers an extended

range up to 36 V, bringing greater flexibility in Flyback

or Forward applications.

Internal high-voltage startup switch: reaching low

levels of standby power represents a difficult exercise when
the controller requires an external, lossy, resistor connected
to the bulk capacitor. Due to an internal logic, the controller
disables the high-voltage current source after startup which
no longer hampers the consumption in no-load situations.

Skip-cycle capability: a continuous flow of pulses is not

compatible with no-load standby power requirements.
Slicing the switching pattern in bunch of pulses drastically
reduces overall losses but can, in certain cases, bring
acoustic noise in the transformer. Due to a skip operation
taking place at low peak currents only, no mechanical noise
appears in the transformer. Furthermore, the skip threshold
is made programmable to allow the best trade-off between
noise and efficiency.

Standby Detect/Shutdown of the PFC front-stage: The

NCP1239 incorporates an internal logic that is able to detect

a standby situation. Pin1 state changes in accordance to the
detected mode (standby or normal mode). Simply connect a
pnp transistor between the NCP1239 V

and the PFC

controller one and drive it using Pin 1, to enable the PFC
stage in normal mode and disable it in standby.

Soft-start: the capacitor connected to Pin 6 provides a

soft-start sequence that precludes the main power switch
from being stressed upon startup. The same voltage is also
used to perform frequency jittering and timing for the fault
condition detection.

Major Fault Detection: the circuit detects when Pin 3

voltage exceeds 2.4 V. When this occurs, the NCP1239
considers that a major fault is present and as a consequence,
the circuit gets permanently latched-off. In this mode, the
circuit needs the V

to go down below 4 V to reset, for

instance when the user un-plugs the SMPS. This capability
is mainly intended to detect an overvoltage condition or/and
an over-heating of the application that would be sensed by
a thermistor.

Brown-out detection: by monitoring the level on Pin 5

during normal operation, the controller protects the SMPS
against low mains conditions. When the Pin 5 voltage falls
below 250 mV, the controllers stops pulsing until this level
goes back to 500 mV to prevent any instability.

Short-circuit protection: short-circuit and especially

overload protections are difficult to implement when a
strong leakage inductance affects the transformer (the
auxiliary winding level does not properly collapse

...

). Here,

every time the feedback pin is at its maximum (higher than
5 V practically), an error flag is asserted and the circuit
activates a timer that is programmed by the Pin 6 capacitor.
If Pin 6 reaches 4.3 V while the error flag is still present, the
controller stops the pulses and goes into a latch-off phase,
operating in a low-frequency burst-mode. As soon as the
fault disappears, the SMPS resumes its operation. The
latch-off phase can also be initiated, more classically, when
V

drops below UVLO (11.2 V typical).

Adjustable frequency and Internal dithering for

improved EMI signature: Pin 4 offers a means to precisely
adjust the switching frequency through a simple resistor to
ground. Frequency operation is allowed up to 250 kHz. By
modulating the internal switching frequency with the Pin 6
saw-tooth (100 Hz with 390 nF), natural energy spread
appears and softens the controller's EMI signature.

5 V reference voltage: a 5 V regulator is provided to help

biasing any external circuitry in the vicinity of the controller.
This reference voltage can typically supply up to 10 mA.

NCP1239

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Startup Sequence

When the power supply is first connected to the mains

outlet, the internal current source (typically 3.6 mA) is
biased and charges up the V

capacitor. When the voltage

on this V

capacitor reaches the V

CCON

level (typically

16.4 V), the current source turns off and no longer wastes
any power. At this time, the energy stored by the V

capacitor serves to supply the controller and the auxiliary
supply is supposed to take over before V

collapses below

CCOFF

. Figure 33 shows the internal arrangement of this

structure:

Figure 33.

The current source brings V

above 16.4 V and then turns off

3.6 mA/0

Aux

16.4 V /
11.2 V

As soon as V

reaches 16.4 V, driving pulses are

delivered on Pin 12 and the auxiliary winding grows up the
V

pin. Because the output voltage is below the target (the

SMPS is starting up), the feedback pin is at its maximum
voltage. A resistor divider outputs the third of the feedback
voltage that forms the current setpoint. This setpoint is
clamped and the limitation level slowly increases until it
reaches 0.9V during the soft start time. In nominal operation,
the setpoint clamp keeps equal to 0.9 V (refer to Figure 34).

As soon as the feedback voltage is high enough to activate

the 0.9 V setpoint clamp (during the startup period but also
anytime an overload occurs), an internal error flag is
asserted, testifying that the system is pushed to the
maximum power. At that moment, a 100 ms time period
(typically, with C

pin6

=390 nF that also corresponds to 7.5 ms

soft -start) starts while a logic block observes this error flag.
If the error flag keeps asserted all along the 100ms period,
then the controller assumes that the power supply really
undergoes a fault condition and immediately stops all pulses
to enter a safe burst operation. The 100 ms timer enables to
distinguish a startup phase (shorter than 100 ms) from an
overload condition. If the error flag is released before the
100 ms period has elapsed, the controller concludes that no
error is present and resets the timer to use it for other
purposes (e.g. frequency dithering).

Figure 34. Current Control

Pin 10 monitors the power switch current and compares it to the current setpoint (one third of the feedback voltage). The
current setpoint is limited by the soft-start during the power-on sequence and permanently clamped to 0.9 V In the
NCP1239F, a second pin (Pin 9) monitors the current to clamp the power.

CLK

Current Sense
Comparator

Current Sense

Over Power
Comparator

Feedback

Rramp

Rsense

Vdd

0.9 V

Vin

Overcurrents Compensation

to
Standby Management
(Skipping, GTS)

LEB

PWM Latch

Rcomp

Over Power
Limit

500 mV

LEB

Pin 5 (Brown-Out)

/ 3

Ramp Compensation

oscillator

Soft-start

20k

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PFC Startup Sequence

To ensure an adequate startup sequence of both PWM

section and the PFC stage, some logic and timing need to be
included as shown on the internal diagram. The key point
here is the fact that the PFC always starts after the PWM
section. As a result, the SMPS must be designed to cope with
transient universal mains operation. Why this? Because of
the light-to-heavy load transition where a case exists when
the PFC is off, the PWM in standby and the load is suddenly
applied. In this scenario, the PWM section must sustain the
entire transient period that lasts until the PFC re-starts since
it has been deactivated for standby.

The standby detection block generates an internal signal

"pfcON" that controls Pin 1 in accordance to the operation
mode:

- "pfcON" is high in normal mode and a current source
draws 1 mA from Pin 1,

- "pfcON" is low in standby to disable the 1 mA current
source. A 10 k

resistor pulls up Pin 1 to V

This configuration makes it ideal to drive a pnp transistor

that connects or disconnects the NCP1239 V

to the PFC

controller one (refer to Figure 37). The "pfcON" signal is
activated following Figure 36 diagram. Let's split this
drawing in different time periods to clearly depict signal
assertions:

Power on: during this time, V

rises up, the V

capacitor being charged by the 3.6 mA current source. When
V

exceeds V

CCON

(16.4 V typ.), driving pulses are

delivered to the MOSFET in an attempt to crank the power
supply. V

collapses (because the V

capacitor alone

delivers the energy) until sufficient auxiliary voltage is built
up in order to take over the startup sequence and thus
self-supply the controller. As long as the output voltage has
not reached its wished value, the controller pushes for the
maximum peak current. During the soft-start (7.5 ms with
390 nF on Pin 6), the maximum permissible current linearly
increases till the maximum peak setpoint is reached, the

internal 0.9 V Zener diode actively clamping the current
amplitude to (0.9 V/Rsense). During this time, the NCP1239
asserts an error flag. A maximum current condition being
observed, the circuit determines if this state results from
either a normal response (startup or a transient period) or a
fault condition. To make the difference, each time the error
flag is asserted, a 100 ms timer starts to count down. If the
error flag keeps asserted for the 100 ms period, there is a
fault and the PWM controller enters a safe, auto-recovery,
burst mode to limit the dissipated heat (see below for more
details). During the Power-on sequence, "pfcON" keeps
low to pull-up Pin 1 to V

until the error flag is down.

When the error flag is down, the power supply has entered
regulation, its auxiliary voltage is stable, then Pin 1 can turn
low (1 mA sink current) to safely allow PFC operation.

Entering Standby: when skip-cycle starts to activate, a

100 ms countdown takes place and the logic observes the
skip activity. If the skip activity is still there at the end of the
100 ms, then standby is confirmed and the NCP1239 pulls
up Pin 1 to V

to shut down the PFC.

Leaving standby: in this case, as soon as the skip-cycle

activity disappears, the circuit immediately re-activates the
1 mA sinking current source of Pin 1, to enable the PFC:
there is no reaction delay in this situation.

Short-circuit condition: a short circuit is detected on the

primary side by measuring the time the error flag is asserted.
As explained, if this flag is asserted longer than 100 ms, then
the PWM stops oscillating and enters a safe burst mode. In
this case, Pin 1 is pulled up to V

and the PFC is shut down.

During the burst, it is not activated (PFC is off) until the fault
goes away and the power supply resumes operation. The
PFC being shut off in short-circuit conditions, it naturally
reduces the main MOSFET stress.

Latch-off mode: if the controller is permanently

latched-off due to a major fault (Pin 3 detection of an OVP
or an excessive external temperature), the PFC is kept off
(Pin 1 being tied to V

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The PFC controller connection is really straightforward as

testified by Figure 37: simply connect to Pin 1, the base of
a pnp transistor that connects the PFC's V

to the NCP1239

one (perhaps add a small decoupling capacitor like a 0.1

on the PFC) and this is all! The PFC startup network goes

away as it is fully supplied by the PWM auxiliary winding
and even high quiescent current devices do not hamper the
standby power since they are completely disconnected in
standby.

Figure 37.

The NCP1239 turns off the pnp Q1 during the standby so that the PFC controller is no longer supplied in this mode.

NCP1239

PFC Controller

PFC_V

PFC stage

Rectified
AC line

Short-Circuit or Overload Condition

The NCP1239 differs from other controllers in the sense

that a fault condition is detected independently of the
auxiliary voltage level. In auxiliary supply-based power
supplies, it is necessary that the (isolated) secondary output
conditions properly reflects on the (non-isolated) auxiliary
winding in order to instruct the controller on what is
happening on the other side of the transformer. For the
following reasons, it sometimes becomes extremely
difficult to build an efficient short-circuit protection
circuitry and even more difficult to implement over power

detection (e.g. the output load is 25% above the nominal
value but Vout is still present).

The primary leakage inductance is high: this is probably

the main reason why building efficient short-circuit
detection is difficult. When the power switch opens, the
leakage inductance superimposes a large overvoltage spike
on the drain voltage. This spike is seen on the secondary side
but also on the auxiliary winding. Unfortunately, since the
V

capacitor and the auxiliary diode form a peak rectifier,

the auxiliary V

often depends on this peak value rather

than the true plateau which corresponds to the output level.

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Figure 38.

236U

240U

244U

248U

252U

15.0

5.00

15.0

25.0

Leakage effect:
Vpeak = 24.2V

"clean" plateau
V = 13.4V

The leakage effect seen on the auxiliary side pulls-up the final level peak-rectified by the diode

On Figure 38's example, one can clearly observe the

difference between the peak and the real plateau DC level.
The delta is around 10 V, which obviously degrades the
auxiliary image of the secondary side. When a short-circuit
occurs, the leakage can be so strong that the whole plateau
has dropped to a few volts, but the leakage contribution
becomes so energetic (Ip = Ip max.) that even a few

duration is enough to prevent V

auxiliary from collapsing

and thus stopping the pulses. Needless to say that over power
detection is simply impossible.

Low standby power requirement decreases V

no-load: this is particularly true if you try to reach less than
100 mW at high line. Due to skip-cycle, the continuous flow
of pulses turns into bunches of pulses (sometimes 1-2 pulses
only) that can be spaced by 50ms or more in certain cases.
The energy content in each bunch of pulses does not suffer
any attenuation. For instance, to lower Figure 38's peak, you
could think of inserting a resistor with the auxiliary diode to
form a low pass filter with the V

capacitor. Unfortunately,

it would drastically reduce the V

capacitor refueling

current and V

could not be maintained. To compensate

that effect, a solution could be to increase the turn ratio, but
then the peak rectification problem comes back again.

As one can see, a short-circuit protection free of the V

level would be the best solution. This is exactly what the
NCP1239 delivers with the internal 100 ms timer (390 nF
being connected to Pin 6). As soon as the internal 0.9 V error
flag is asserted high, a 100 ms timer gets started. If the error
flag keeps asserted during the 100 ms period, then the
controller detects a true fault condition and stops pulsing the
output. If this is a simple transient overload, e.g. the error
flag goes back to a normal level before the 100 ms period has

elapsed, nothing happens and the controller continues
working normally.

When a fault is detected, we have seen that the controller

stops delivering pulses. At this time, V

starts to drop

because the power supply is locked off. When the V

drops

below V

CCOFF

(11.2 V typical), it enters a so-called

latch-off phase where the internal consumption is reduced
down to about 400

mA. The V

capacitor continues to

deplete, but at a lower rate. When V

finally reaches the

latch-off level (around 6.9 V), the startup current source
turns on and pulls V

above V

CCON

, exactly as a startup

sequence would do. When V

exceeds V

CCON

(16.4 V),

pulses are delivered and can last 100 ms maximum if there
is enough voltage or can be prematurely interrupted if V

falls below V

CCOFF

. Figure 39 shows the difference between

these two cases. As already explained, in short-circuit
bursts, the PFC section is not validated.

The short-circuit protection features a so-called

auto-recovery circuitry. That is to say, during the 100 ms
period, the power supply attempts to startup. If the fault has
gone, then the controller resumes from the fault and the
power supply operates again. If the fault is still present, the
pulses are stopped at the end of the 100 ms section (Tpulse)
for a given time period Tfault. At the end of Tfault, a new
100 ms attempt is made and so on. To avoid any thermal
runaway, a burst duty-cycle defined by Tpulse/(Tfault+
Tpulse) below 10% is desirable ((Tfault+Tpulse) is the burst
period). If the 100 ms is made by an internal timer in
conjunction with the Pin 6 capacitor, the Tfault duration
builds on the V

capacitor which is charged/discharged

two times. Figure 40 on the following page portrays this
behavior.

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If by design we have selected a 47

mF V

capacitor, it

becomes easy to evaluate the burst period and its duty-cycle.
This can be done by properly identifying all time events on
Figure 40 and applying the classical formula: t = C *

DV / i.

To simplify, let's consider t1 starts while V

= V

CCOFF

Then:

t1: I = I

3 = 400

mA,

V= 11.2 6.9 = 3 V

t1 = 505 ms

t2: I = 3.6 mA,

V= 16.4 6.9 = 9.5 V

! t2 = 124 ms

t3: I = 400

mA,

V= 16.4 11.2 = 5.2 V

! t3 = 611 ms

t'1 = t1= 505 ms

t'2 = t2 = 124 ms

The total period duration is thus the sum of all these events

which leads to Tfault = 1793 ms. If Tpulse = 100 ms, then
our burst duty-cycle equals 100/(1869 + 100)

5%, which

is excellent.

In fact, the calculation assumption, t1 starts while

= V

CCOFF

, gives the worse case since the duty cycle is

calculated in the case where Tpulse exactly equals the active
phase duration (switching period when V

decreases from

CCON

to V

CCOFF

In fact, Tpulse is generally:

- shorter than the switching phase period. In this
case, t1 is longer since the latched off phase starts
earlier (at a V

higher than V

CCOFF

). As a

consequence, the final duty cycle is lower than
previously estimated,
- longer than the switching phase period. In this case,
the circuit detects an overload condition simply
because V

drops below V

CCOFF

(11.2 V) before

the fault timer has elapsed. Tpulse is lower than 100
ms and as a result the duty cycle is also lower.

(Major) Fault Detection and Latched Off Mode

The NCP1239 features a fast comparator that

permanently

monitors the "Fault Detect" pin level. If for any

reason this level exceeds 2.4 V (typical), the part
immediately stops oscillating and stays latched off until the
user cycles down the power supply. This enables the SMPS
designer to externally shut down the part in particular when
a major default occurs, e.g. an Overvoltage Protection
(OVP). Figure 41 shows what happens when the part is
latched:

CCOFF

Figure 41.

Drv

Latch-off phase level

Logic reset level

Pin 3

Stop!

2.4 V

When Vpin3 exceeds 2.4 V, NCP1239 permanently latches-off the output pulses

0until its V

goes below 4 V. The figure can

illustrate a case where a thermistor supplied by REF5V is connected to Pin 3 to detect excessive temperatures of the application
(refer to application schematic).

The user has unplugged, reset!

CCON

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Pin 3 can serve to build an Overvoltage Protection by

placing a Zener between the voltage to measure (e.g., V

)

and Pin 3 (refer to application schematic). If a 15 V Zener is
applied, the Pin 3 comparator will switch when (V

- 15 V)

exceeds the 2.4 V internal reference, that is, when V

higher than 17.5 V.

This pin can also monitor the temperature using an

external thermistor (refer to application schematic).
Thermistors can be of Negative Temperature Coefficient
(NTC) type (the resistance decreases versus the
temperature) or of Positive Temperature Coefficient (PTC)
type (the resistance increases versus the temperature). Let's
assume that a NTC thermistor is used (as in the application
schematic). Placing it between the 5 V reference voltage
(REF5V) and Pin 3, and a classical resistance between Pin 3
and ground, the Pin 3 voltage equals:

Vpin3

)

Rthermistor

5 V

where R and R

thermistor

are respectively the resistor and the

thermistor resistance.

thermistor

decreasing versus the temperature, the Pin 3

voltage (V

pin3

) increases when the temperature grows up.

For instance, the thermistor resistance can be in the range

of 500 k

at 25

C and as low as 5 k

at 130

C that as an

example, one can take as the temperature limit the
application must not exceed. Choosing R equal to 5k, the
Pin 3 voltage at 130

C that equates:

Vpin3(130

5 k

)

5 k

5 V

2.5 V

triggers the fault comparator.

This example illustrates that one must just select the

bottom resistor so that it exhibits the same resistance as the
thermistor at the temperature to be detected.

If the thermistor is a PTC, it must be placed between Pin 3

and ground. One must place a resistor between the 5 V
reference voltage and Pin 3. Similarly, the resistor must be
selected so that its resistance equals the thermistor one at the
temperature to be detected.

Brown-Out and Over Power Limitation

SMPS are designed for a given input range. When the

input voltage is too low (brown-out), the SMPS tends to
compensate by sinking an increased current from the line.
As a result the power components may suffer from an
excessive heating and ultimately the SMPS may be
destroyed. To avoid such a risk, the NCP1239 incorporates
a brown-out detection that monitors the portion of the input
voltage that is applied to Pin 5.

Figure 42.

pin5

CMP

240 mV 500 mV

An hysteresis comparator monitors the SMPS input voltage

Rupper

CMP

Driver is off
as long as
CMP is low

Driver

Rlower

500 mV if CMP is low
240 mV if CMP is high

Also called "Bulk OK" signal (BOK), the Brown-Out

(BO) protection prevents the power supply from being
adversely destroyed in case the mains drops to a very low
value. When it detects such a situation, the NCP1239 no
longer pulses but waits until the bulk voltage goes back to its
normal level. A certain amount of hysteresis needs to be
provided since the bulk capacitor is affected by some ripple,
especially at low input levels. For that reason, when the BO

comparator toggles, the internal reference voltage changes
from 500 mV to 240 mV. This effect is not latched: that is to
say, when the bulk capacitor is below the target, the
controller does not deliver pulses. As soon as the input
voltage grows-up and reaches the level imposed by the
resistive divider, pulses are passed to the internal driver and
activate the MOSFET.

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Figure 43 offers a way to connect the elements around Pin 5 to create a Brown-Out detection:

Figure 43.

ac line

Preconverter

PFC

Cbulk

Cfil

Rupper

Rlower

to converter

Input
Filtering
Capacitor

Example where the voltage of the bulk capacitor is used for the brown-out Protection

The calculation procedure for Rupper and Rlower is easy.

The first level transition is always clean: the SMPS is not
working during the startup sequence and there exists no
ripple superimposed on Cbulk. Supposed we want to start
the operation at Vbulk = Vtrip = 120 VDC (i.e., VinAC =
85 V).

1. Fix a bridge current Ib compatible with your

standby requirements, for instance an Ib of 50

mA.

2. Then evaluate Rlower by: Rlower = 0.5/Ib =

10 k

3. Calculate Rupper by: (Vtrip 0.5 V)/Ib =

(120 0.5)/50

mA = 2.39 M

The second threshold, the level at which the power supply

stops (VBO), depends on the capacitor Cfil but also on the
selected bulk capacitor. Furthermore, when the load varies,
the ripple also does and increases as Vin drops. If Cfil allows
a too high ripple, chances exist to prematurely stop the
converter. By increasing Cfil, you have the ability to select
the amount of hysteresis you want to apply. The less ripple
appears on a Pin 5, the larger the gap between Vtrip and VBO
(the maximum being VBO = Vtrip/2). The best way to assess
the right value of Cfil, is to use a simple simulation sketch
as the one depicted by Figure 44. A behavioral source loads
the rectified DC line and adjusts itself to draw a given

amount of power, actually the power of your converter
(35 W in our example). The equation associated to Bload
instructs the simulator not to draw current until the
Brown-Out converter gives the order, just like what the real
converter will do. As a result, Vbulk is free of ripple until the
node CMP goes high, giving the green light to switch pulses.
The input line is modulated by the "timing" node which
ramps up and down to simulate a slow startup/turn-off
sequence. Then, by adjusting the Cfil value, it becomes
possible to select the right turn-off AC voltage. Figure 45
portrays the typical signal you can expect from the
simulator. We measured a turn-on voltage of 85 VAC
whereas the turn-off voltage is 72 VAC. Further increasing
Cfil lowers this level (for instance, a 1

mF capacitor gives

VBO = 65 VAC in the example).

As we have seen, the load variations will modify this

turn-off level. To remove the dependency between VBO
and the load, it is possible to directly sense the rectified input
line present at the PFC stage input, as shown in Figure 46.
In that case, there still exists the input line ripple, but this
ripple is independent of the load. By adjusting Cfil
capacitance and the divider section, you can build a
brown-out detection independent of the load.

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Figure 46.

ac line

Preconverter

PFC

Cbulk

Cfil

Rupper

Rlower

to converter

Rectified AC Line Sensing

Input
filtering
Capacitor

A second option to directly sense the mains

This second option that directly senses the input voltage

(see Figure 46), enables a more direct under-mains
detection. Even in a brown-out conditions, the PFC
pre-converter may be able to maintain a sufficient bulk
voltage, possibly at the price of some excessive stress.
Measuring the rectified AC line instead of the bulk voltage,
the NCP1239 more surely protects the PFC stage in
brown-out conditions.

Using:

- Rlower = 10 k

- Rupper = 2. 39 M

- Cfil = 1

mF,

One obtains the following voltage thresholds:

- Vtrip = 85 Vrms,
- VBO = 65 Vrms.

Over Power Limit (NCP1239F)

Overload conditions may push the converter to draw an

excessive power (which generally increases versus the input
voltage). One must avoid such a behavior:

a) not to have to dimension the converter for a power

higher than the nominal one,

b) to meet SMPS specifications that often request the

power not to exceed a given level.

In addition, it is not recommended to provide the output

with more power than normally necessary. To the light of
these statements, it becomes interesting to accurately limit
the amount of power drawn from the AC line in fault
conditions. The easiest way to do so consists of clamping the
peak current since in a discontinuous mode flyback
converter, the input power (Pin) can be calculated as
follows: Pin = 1/2 * Lp * Ip

* fsw, where Lp is the primary

inductor, Ip

is the inductor peak current and fsw is the

switching frequency.

Practically, a sense resistor converts the primary current

into a voltage that is compared to a voltage reference. When
the voltage representative of the current exceeds the voltage
reference, the controller turns off the power switch. The
theoretical maximum peak current is then: Imax =
Vocp/Rsense, where Vocp is the reference voltage (or
overcurrent protection threshold) and Rsense is the sense
resistor.

Unfortunately, the controller cannot turn off the power

switch immediately when it detects that the current exceeds
its maximum permissible level. Internal propagation delays
differ the drive turn low. In addition, the power switch needs
some time to turn off. Finally, the real current stop can be
250ns or more delayed. During this time, the current
continues ramping up so that an overcurrent is obtained.

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Actual Peak Current

High Input
Voltage

Low Input
Voltage

Vopl/Rsense

Wished Maximum

Peak Current

Figure 47.

The propagation delay (

dt) produces overcurrents (dI

at low line,

at high line in the figure) that are proportional to the input volt-

age. As a consequence, the actual maximum current and then the power limit gets higher when the AC line increases.

I max

Vocp

Rsense

)

Vin

@ d

, where Vin is the converter

input voltage and

t is the total delay in turning off the power

switch.

The NCP1239 enables the compensation of the second

term in the Imax equation for a precise limitation of the peak
current. A current source (I

pin9

) proportional to the Pin 5

voltage flows out of Pin 9. Since Pin 5 receives a voltage
proportional to the input voltage for brown-out detection,
I

pin9

is proportional to the input voltage too. An external

resistor Rcomp can be connected between Pin 9 and the
positive terminal of Rsense, so that Pin 9 monitors the
following voltage:

Vpin9

[Rsense

(Ip

)

Ipin9)]

)

(Rcomp

Ipin9)

pin9

being small compared to the inductor current, the Pin 9

voltage simplifies as follows:

Vpin9

(Rsense

Ip)

)

(Rcomp

Ipin9)

pin9

is proportional to the Pin 5 voltage (80

mA/V*V

pin5

see parameters specification table) and V

pin5

is a portion of

the input voltage (V

pin5

= k

*Vin). Finally,

Ipin9

A V

kBO

Vin

The voltage V

pin9

is compared to the internal reference

Vocp. When V

pin9

reaches Vocp, the corresponding

threshold current (Ipth) is deducted from:

Vopl

(Rsense

Ipth)

)

(Rcomp

A V

kBO

Vin)

Then,

Ipth

Vopl

(Rcomp

A V

kBO

Vin)

Rsense

Taking into account the overcurrent resulting from the
propagation delays, the maximum current is finally:

I max

Vocp

Rsense

Rcomp

A V

kBO

Vin

Rsense

)

Vin

@ d

Choosing Rcomp so that

Rcomp

A V

kBO

Rsense

) d

the current limit is made constant in the whole input voltage
range (Imax = Vocp/Rsense).

As an example, let's assume that:

- the minimum input voltage for operation is 100 V =>
k

=0.5/100=0.005,

- Rsense is 0.25

- Lp=500

mH,

- The total propagation delays are

t = 350 ns,

Then, the Rcomp resistor should be:

Rcomp

+ d

Rsense

m @

kBO

350 n

0.25

m @

0.005

500 m

[

438

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Figure 48. NCP1239F

An (averaged) portion of the input voltage is applied to the brown-out pin. A current source proportional to this voltage, flows through an
external resistor Rcomp to form an offset proportional to the (average) input voltage. Rcomp should be selected so that the offset compen-
sates the overcurrent sensed by the current sensing resistor Rsense.

CLK

to
Current Sense
Comparator

Rcomp

Current Sense

to Brown-Out
Comparator

Brown-Out

Over Power
Limit

Rramp

Rsense

comp

0.5V

Vdd

pin5

Vin

Rbo1

Rbo2

Cbo

comp

= k*R

comp

pin5

A/V*V

pin5

LEB

PWM Latch

Maximum Duty Cycle Limitation (NCP1239V)

Pin 9 sources a 55

mA current. By placing a resistor

between this pin and ground, one builds a voltage that forces
the maximum on-time. Practically the Pin 9 voltage is
compared to the positive ramp of the internal oscillator and
the power switch is allowed to be on, only when the ramp is
below V

pin9

Then the maximum on-time is given by:

(ton)max

Cosc

Vpin9

Iosc

where Cosc and Iosc are respectively the capacitor and the
charging current of the oscillator.

pin9

being the product of the Pin 9 current by the Pin 9

resistance (R

pin9

external resistor connected to Pin 9),

results in:

(ton)max

Cosc

IDmax

Iosc

Rpin9

, where I

Dmax

is the

Pin 9 current source.

One can deduct the maximum duty cycle (Dmax) by

dividing by the period T:

Dmax

Cosc

IDmax

Iosc

Rpin9

KDmax

Rpin9

where

KDmax

Cosc

IDmax

Iosc

Dmax

is specified within the parameters' table. Please

note that Cosc and Iosc are the internal capacitor and current
(respectively), that set the switching period (T). Hence,

Cosc

Iosc

is a constant and K

Dmax

is independent of the

switching frequency.

In the NCP1239F, the maximum duty-cycle is fixed (80%
typically).

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Soft-Start

The NCP1239 features an internal soft-start activated

during the Power On sequence (PON). As soon as V

reaches 16.4 V, the current setpoint is gradually increased
from nearly zero up to the maximum clamping level (e.g.
0.9 V/Rsense). This situation lasts a programmable time that
is adjusted by the Pin 6 capacitor (7.5 ms typically with
C

pin6

= 390 nF). Further to that time period, the current

setpoint is blocked to 0.9 V/Rsense until the supply enters

regulation. The soft-start is also activated at each start of the
active phase of fault burst operation. Every restart attempt
is followed by a soft-start activation.

Generally speaking, the soft-start will be activated when

ramps up either from zero (fresh power-on sequence)

or 6.9 V, the latch-off threshold after an overload detection
(OVL) for instance. Figure 49 shows the soft-start behavior.
The time scales are purposely shifted to offer a better zoom
portion.

Figure 49.

Soft-start is activated during a startup sequence or an OVL condition

16.4V

6.9V

Internal Ramp Compensation

Ramp compensation is a known mean to cure

sub-harmonic oscillations. These oscillations take place at
half the switching frequency and occur only during
Continuous Conduction Mode (CCM) with a duty-cycle
greater than 50%. To lower the current loop gain, one usually
injects between 50 and 100% of the inductor down-slope.
Figure 50 depicts how internally the ramp is generated:

Figure 50.

Inserting a resistor in series with the current sense

information brings ramp compensation

LEB

32k

Rramp

Rsense

from
setpoint

3.2V

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In the NCP1239, the ramp features a swing of 3.2 V.

Suppose we select a 65 kHz version. Over a 65 kHz
frequency, it corresponds to a 130 mV/ms ramp. In our
FLYBACK design, let's assume that our primary inductance
Lp is 350 mH, and the SMPS delivers 12 V with a Np:Ns
ratio of 1:0.1. The OFF time slope of the primary current is:

(Vout

)

Vf)

Ns
Np

that is, 371 mA/ms or 37 mV/ms, once

projected over a 0.1

W Rsense for instance. If we select 75%

of the down-slope as the required amount of ramp
compensation, then we shall inject 27 mV/ms. Our internal
compensation being of 208 mV/ms, the divider ratio
(divratio) between R

ramp

and the 32 k

W is 0.178. A few lines

of algebra to determine R

ramp

Rramp

19 k

divratio

divratio)

6.92 k

The ramp is disabled during standby (i.e., when pfcON is

low). This inhibition avoids that the ramp compensation
modifies the setpoint above which the NCP1239 enables
PFC.

Frequency Jittering

Frequency jittering is a method used to soften the EMI

signature by spreading the energy in the vicinity of the main
switching component. NCP1239 offers a +3.5% deviation of
the nominal switching frequency. The sweep saw-tooth is
internally generated and modulates the clock up and down
with a period depending on the Pin 6 capacitor (10 ms
typically with 390 nF, 10 mS * Cpin6 / 390 nF in general).
Again, if one selects a 65 kHz version, the frequency will
equal 65 kHz in the middle of the ripple and will increase as
V

pin6

rises or decrease as V

pin6

ramps down. Figure 51

portrays the behavior we have adopted:

65kHz

67.6kHz

62.4kHz

Internal
ramp

Internal
sawtooth

10ms

Figure 51.

The V

pin6

ramp is used to introduce frequency jittering on the oscillator saw-tooth

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Skipping Cycle Mode

The NCP1239 automatically skips switching cycles when

the output power demand drops below a given level. This is
accomplished by monitoring the FB pin. In normal
operation, Pin 8 imposes a current setpoint accordingly to
the load value. If the load demand decreases, the internal
loop asks for less peak current. When this setpoint reaches
a fixed determined level, the IC prevents the current from
decreasing further down and starts to blank the output
pulses: the IC enters the so-called skip-cycle mode, also
named controlled burst operation. The default skip-cycle
current is internally frozen to 30% of the maximum peak
current which is 0.5 V/Rsense The power transfer now
depends upon the width of the pulse bunches (Figure 52).

Suppose we have the following component values:

Lp, primary inductance = 350

fsw , switching frequency = 65 kHz
Ip skip = 600 mA (or 140 mV/Rsense)

The theoretical power transfer is therefore:

1/2 * Lp * Ip

* fsw = 4 W

If this IC enters skip-cycle mode with a bunch length of

10 ms over a recurrent period of 100 ms, then the total power
transfer is:

4 W * 10 ms/100 ms = 400 mW
To better understand how this skip-cycle mode takes

place, a look at the operation mode versus the FB level
immediately gives the necessary insight:

4.3 V, FB Pin open
2.7 V upper dynamic range

Normal current mode operation

0.43 V

Skip-cycle operation
Ip

MIN

= 150 mV/Rsense

FB Pin Voltage

Figure 52.

When FB is below the skip-cycle threshold (0.43 V by

default), the circuit skips the switching cycle. When the IC
enters the skip-cycle mode, the peak current cannot go

below (0.43 V/3)/Rsense or 140 mV/Rsense. Figure 53
shows different values of pulse widths when the SMPS starts
to skip-cycles at different power levels:

Power P1

Power P2

Power P3

Figure 53.

Output pulses at various power levels (X = 5

ms/div) P1 < P2 < P3

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315.4U

882.7U

1.450M

2.017M

2.585M

300.0M

200.0M

100.0M

25% of max Ip

Max peak
current

Figure 54.

The skip-cycle takes place at low peak currents which guaranties noise free operation

PFC Inhibition in Standby

The circuit detects a light load condition by permanently

monitoring the skip-cycle comparator activity: in normal
load condition this comparator keeps quiet. As soon as the
load strongly decreases, this comparator starts to toggle at a
low frequency rate: we are entering skip-cycle and the
opto-coupler operates in a digital manner, ON/OFF.
Figure 54 shows the way skip-cycle is detected. In skip
mode, the feedback voltage oscillates around V

pin7

(If no

voltage is applied to the Pin 7, a 430 mV voltage source
supplies a default value through a high impedance resistor).
In these conditions, the skip comparator ("COMP1") that
turns on and off (to adjust the skip mode bunches of pulses),
sets the standby detection latch. A second comparator
("COMP2") compares the feedback voltage (FB or V

pin8

) to

1.7*V

pin7

As long as the load keeps light, FB does not exceed

1.7*V

pin7

(i.e., 0.74 V typical if no voltage is forced to

Pin 7). A timer counts down and if COMP2 keeps high for
100 ms (typically with 390 nF on Pin 6), the NCP1239
considers that the system runs in the standby mode. Pin 1
turns high, a 10 k

resistor tying the pin to V

. If as shown

in Figure 37, Pin 1 directly drives a pnp transistor that is
connected between V

and the PFC V

, this switch turns

off in standby. As a result, this transistor stops feeding the
PFC V

and ultimately shuts the PFC down.

As soon as FB exceeds 1.7*V

pin7

, the circuit leaves the

standby mode without any delay by forcing a 1 mA sinking
current source on Pin 1, that re-activates the pnp transistor
and then the PFC stage.

One can note that there is a 1/3 ratio between the actual

current setpoint and the feedback value FB. Therefore the
default thresholds for standby detection and normal mode
recovery (0.43 V, 0.74 V) actually corresponds to the
140 mV and 250 mV setpoints.

Figure 55.

A delay is inserted to avoid false tripping of the GTS signal

70%

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Figure 56.

One clearly sees that the GTS signal does not react to the fugitive low FB Pin condition during startup

Figure 57.

Internal Go-To-Standby signal elaboration

Skip

0.43V

FB < V

pin7

=> Skip high

COMP1

COMP2

Stby_detect

GTS

pin1

100 ms timer (*)
(SS and timer block)

15r

25r

100k

Skip
Adjust

REF5V

FB > 1.7 * V

pin7

=> Stby_detect RESET

(*) the 100 ms delay is programmed by the Pin 6 capacitor

Suppose our Flyback controller is built with a transformer

primary inductance of 250

mH. To pass 120 W, we assume

that a peak current of 4.2 A was needed. Due to these
numbers, we can easily now when the GTS signal will be
asserted:

Lp, primary inductance = 250

h = 85%
fsw, switching frequency = 65 kHz

Pout

h @

fsw

4.2 A

Ip skip = 30% of Ip max = 1.26 A

The theoretical region at which the SMPS will enter

standby is: 1/2 * Lp * Ip

* fsw *

h 11 W. This number

can vary depending on the line level since the propagation
delay becomes a sensitive parameter, and on the efficiency
that is difficult to precisely predict in light load conditions.
The peak current at which the SMPS will leave standby is
48% of the peak current which means that a power of 28 W
is necessary to re-trigger the PFC.

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PACKAGE DIMENSIONS

SO-16

FD SUFFIX

CASE 751B-05

ISSUE J

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: MILLIMETER.

3. DIMENSIONS A AND B DO NOT INCLUDE

MOLD PROTRUSION.

4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)

PER SIDE.

5. DIMENSION D DOES NOT INCLUDE DAMBAR

PROTRUSION. ALLOWABLE DAMBAR

PROTRUSION SHALL BE 0.127 (0.005) TOTAL

IN EXCESS OF THE D DIMENSION AT

MAXIMUM MATERIAL CONDITION.

SEATING

PLANE

X 45

8 PL

-B-

-A-

0.25 (0.010)

-T-

16 PL

0.25 (0.010)

DIM

MIN

MAX

MIN

MAX

INCHES

MILLIMETERS

9.80

10.00

0.386

0.393

3.80

4.00

0.150

0.157

1.35

1.75

0.054

0.068

0.35

0.49

0.014

0.019

0.40

1.25

0.016

0.049

1.27 BSC

0.050 BSC

0.19

0.25

0.008

0.009

0.10

0.25

0.004

0.009

5.80

6.20

0.229

0.244

0.25

0.50

0.010

0.019

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to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability
arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages.
"Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All
operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights
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NCP1239/D

The product described herein (NCP1239), may be covered by one or more of the following U.S. patents: 6,362,067, 6,385,060, 6,429,709. There may be
other patents pending.

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