Semiconductor Components Industries, LLC, 2004
October, 2004 - Rev. 6
1
Publication Order Number:
NCP1421/D
NCP1421
600 mA Sync-Rect PFM
Step-Up DC-DC Converter
with True-Cutoff and
Ring-Killer
NCP1421 is a monolithic micropower high-frequency step-up
switching converter IC specially designed for battery-operated
hand-held electronic products up to 600 mA loading. It integrates
Sync-Rect to improve efficiency and to eliminate the external
Schottky Diode. High switching frequency (up to 1.2 MHz) allows
for a low profile, small-sized inductor and output capacitor to be
used. When the device is disabled, the internal conduction path from
LX or BAT to OUT is fully blocked and the OUT pin is isolated from
the battery. This True-Cutoff function reduces the shutdown current
to typically only 50 nA. Ring-Killer is also integrated to eliminate
the high-frequency ringing in discontinuous conduction mode. In
addition to the above, Low-Battery Detector, Logic-Controlled
Shutdown, Cycle-by-Cycle Current Limit and Thermal Shutdown
provide value-added features for various battery-operated
applications. With all these functions on, the quiescent supply
current is typically only 8.5
mA. This device is available in the
compact and low profile Micro8
t package.
Features
Pb-Free Package is Available
High Efficiency: 94% for 3.3 V Output at 200 mA from 2.5 V Input
88% for 3.3 V Output at 500 mA from 2.5 V Input
High Switching Frequency, up to 1.2 MHz (not hitting current limit)
Output Current up to 600 mA at V
IN
= 2.5 V and V
OUT
= 3.3 V
True-Cutoff Function Reduces Device Shutdown Current to
typically 50 nA
Anti-Ringing Ring-Killer for Discontinuous Conduction Mode
High Accuracy Reference Output, 1.20 V
$1.5%, can Supply
2.5 mA Loading Current when V
OUT
> 3.3 V
Low Quiescent Current of 8.5
mA
Integrated Low-Battery Detector
Open Drain Low-Battery Detector Output
1.0 V Startup at No Load Guaranteed
Output Voltage from 1.5 V to 5.0 V Adjustable
1.5 A Cycle-by-Cycle Current Limit
Multi-function Logic-Controlled Shutdown Pin
On Chip Thermal Shutdown with Hysteresis
Typical Applications
Personal Digital Assistants (PDA)
Handheld Digital Audio Products
Camcorders and Digital Still Cameras
Hand-held Instruments
Conversion from one to two Alkaline, NiMH, NiCd Battery Cells to
3.0-5.0 V or one Lithium-ion cells to 5.0 V
White LED Flash for Digital Cameras
Device
Package
Shipping
ORDERING INFORMATION
NCP1421DMR2
Micro8
4000 Tape & Reel
http://onsemi.com
For information on tape and reel specifications,
including part orientation and tape sizes, please
refer to our Tape and Reel Packaging Specifications
Brochure, BRD8011/D.
Micro8
DM SUFFIX
CASE 846A
1
8
PIN CONNECTIONS
FB
OUT
LBI/EN
LBO
REF
LX
GND
BAT
1
8
2
3
4
7
6
5
MARKING
DIAGRAM
1421
= Device Code
A
= Assembly Location
Y
= Year
W
= Work Week
1421
AYW
(Top View)
NCP1421DMR2G
Micro8
(Pb-Free)
4000 Tape & Reel
NCP1421
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2
Chip
Enable
Figure 1. Detailed Block Diagram
_ZCUR
_MSON
_CEN
_PFM
_TSDON
_MAINSW2ON
_MAINSWOFD
_SYNSW2ON
_SYNSWOFD
_V
REFOK
CONTROL LOGIC
20 mV
+
-
PFM
Voltage
Reference
REF
4
LBI/EN
2
+
-
FB
1
+
-
ZLC
+
TRUE CUTOFF
CONTROL
V
DD
GND
V
DD
GND
+
-
+
GND
R
SENSE
GND
SENSEFET
t
M1
V
DD
M3
BAT
5
LX
7
OUT
8
V
BAT
6
GND
V
OUT
LBO
3
_ILIM
0.5 V
1.20 V
M2
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3
PIN FUNCTION DESCRIPTIONS
Pin
Symbol
Description
1
FB
Output Voltage Feedback Input.
2
LBI/EN
Low-Battery Detector Input and IC Enable. With this pin pulled down below 0.5 V, the device is disabled and
enters the shutdown mode.
3
LBO
Open-Drain Low-Battery Detector Output. Output is LOW when V
LBI
is < 1.20 V. LBO is high impedance in
shutdown mode.
4
REF
1.20 V Reference Voltage Output, bypass with 1.0
m
F capacitor. If this pin is not loaded, bypass with 300 nF
capacitor; this pin can be loaded up to 2.5 mA @ V
OUT
= 3.3 V.
5
BAT
Battery input connection for internal ring-killer.
6
GND
Ground.
7
LX
N-Channel and P-Channel Power MOSFET drain connection.
8
OUT
Power Output. OUT also provides bootstrap power to the device.
MAXIMUM RATINGS
(T
C
= 25
C unless otherwise noted.)
Rating
Symbol
Value
Unit
Power Supply (Pin 8)
V
OUT
-0.3, 5.5
V
Input/Output Pins (Pin 1-5, Pin 7)
V
IO
-0.3, 5.5
V
Thermal Characteristics
Micro8 Plastic Package
Thermal Resistance Junction-to-Air
P
D
R
q
JA
520
240
mW
_
C/W
Operating Junction Temperature Range
T
J
-40 to +150
_
C
Operating Ambient Temperature Range
T
A
-40 to +85
_
C
Storage Temperature Range
T
stg
-55 to +150
_
C
Maximum ratings are those values beyond which device damage can occur. Maximum ratings applied to the device are individual stress limit values
(not normal operating conditions) and are not valid simultaneously. If these limits are exceeded, device functional operation is not implied, damage
may occur and reliability may be affected.
1. This device contains ESD protection and exceeds the following tests:
Human Body Model (HBM)
2.0 kV per JEDEC standard: JESD22-A114. *Except OUT pin, which is 1k V.
Machine Model (MM)
200 V per JEDEC standard: JESD22-A115. *Except OUT pin, which is 100 V.
2. The maximum package power dissipation limit must not be exceeded.
PD
+
TJ(max)
*
TA
R
q
JA
3. Latchup Current Maximum Rating:
150 mA per JEDEC standard: JESD78.
4. Moisture Sensitivity Level: MSL 1 per IPC/JEDEC standard: J-STD-020A.
NCP1421
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4
ELECTRICAL CHARACTERISTICS
(V
OUT
= 3.3 V, T
A
= 25
C for typical value, -40
C
v
T
A
v
85
C for min/max values unless
otherwise noted.)
Characteristic
Symbol
Min
Typ
Max
Unit
Operating Voltage
V
IN
1.0
-
5.0
V
Output Voltage Range
V
OUT
1.5
-
5.0
V
Reference Voltage
(V
OUT
= 3.3 V, I
LOAD
= 0
m
A, C
REF
= 200 nF, T
A
= 25
C)
V
REF_NL
1.183
1.200
1.217
V
Reference Voltage
(V
OUT
= 3.3 V, I
LOAD
= 0
m
A, C
REF
= 200 nF, T
A
= -40
C to 85
C)
V
REF_NL
1.174
-
1.220
V
Reference Voltage Temperature Coefficient
TC
VREF
-
0.03
-
mV/
C
Reference Voltage Load Current
(V
OUT
= 3.3 V, V
REF
= V
REF_NL
"
1.5% C
REF
= 1.0
m
F) (Note 5
)
I
REF
-
2.5
-
mA
Reference Voltage Load Regulation
(V
OUT
= 3.3 V, I
LOAD
= 0 to 100
m
A, C
REF
= 1.0
m
F)
V
REF_LOAD
-
0.05
1.0
mV
Reference Voltage Line Regulation
(V
OUT
from 1.5 V to 5.0 V, C
REF
= 1.0
m
F)
V
REF_LINE
-
0.05
1.0
mV/V
FB Input Threshold (I
LOAD
= 0 mA, T
A
= 25
C)
V
FB
1.192
1.200
1.208
V
FB Input Threshold (I
LOAD
= 0 mA, T
A
= -40
C to 85
C)
V
FB
1.184
-
1.210
V
LBI Input Threshold (I
LOAD
= 0 mA, T
A
= -40
_
C to 85
_
C)
V
LBI
1.162
1.230
V
LBI Input Threshold (T
A
= 25
_
C)
V
LBI
1.182
1.200
1.218
V
Internal NFET ON-Resistance
R
DS(ON)_N
-
0.3
-
W
Internal PFET ON-Resistance
R
DS(ON)_P
-
0.3
-
W
LX Switch Current Limit (N-FET) (Note 7)
I
LIM
-
1.5
-
A
Operating Current into BAT
(V
BAT
= 1.8 V, V
FB
= 1.8 V, V
LX
= 1.8 V, V
OUT
= 3.3 V)
I
QBAT
-
1.3
3
m
A
Operating Current into OUT (V
FB
= 1.4 V, V
OUT
= 3.3 V)
I
Q
-
8.5
14
m
A
LX Switch MAX. ON-Time (V
FB
= 1.0 V, V
OUT
= 3.3 V, T
A
= 25
_
C)
t
ON
0.46
0.72
1.15
m
s
LX Switch MIN. OFF-Time (V
FB
= 1.0 V, V
OUT
= 3.3 V, T
A
= 25
_
C)
t
OFF
-
0.12
0.22
m
s
FB Input Current
I
FB
-
1.0
50
nA
True-Cutoff Current into BAT
(LBI/EN = GND, V
OUT
= 0, V
IN
= 3.3 V, LX = 3.3 V)
I
BAT
-
50
-
nA
BAT-to-LX Resistance (V
FB
= 1.4 V, V
OUT
= 3.3 V) (Note 7)
R
BAT_LX
-
100
-
W
LBI/EN Input Current
I
LBI
-
1.5
50
nA
LBO Low Output Voltage (V
LBI
= 0, I
SINK
= 1.0 mA)
V
LBO_L
-
-
0.2
V
Soft-Start Time (V
IN
= 2.5 V, V
OUT
= 5.0 V, C
REF
= 200 nF) (Note 6)
T
SS
-
1.5
20
ms
EN Pin Shutdown Threshold (T
A
= 25
C)
V
SHDN
0.35
0.5
0.67
V
Thermal Shutdown Temperature (Note 7)
T
SHDN
-
-
145
C
Thermal Shutdown Hysteresis (Note 7)
T
SDHYS
-
30
-
C
5. Loading capability increases with V
OUT.
6. Design guarantee, value depends on voltage at V
OUT.
7. Values are design guaranteed.
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5
TYPICAL OPERATING CHARACTERISTICS
1.180
1.185
1.190
1.195
1.200
1.205
0.0
0.1
0.2
0.3
0.4
0.5
0.6
-40
-20
0
20
40
60
80
100
AMBIENT TEMPERATURE, T
A
/
C
SWITCH ON RESIST
ANCE, R
DS
(
O
N)
/
W
P-FET (M2)
N-FET (M1)
V
OUT
= 3.3 V
-40
-20
0
20
40
60
80
100
AMBIENT TEMPERATURE, T
A
/
C
REFERENCE VOL
T
AGE, V
RE
F
/V
0.5
0.6
0.7
0.8
0.9
1.0
-40
-20
0
20
40
60
80
100
0.6
0.9
1.1
1.4
1.6
0
50
100
150
200
250
T
A
= 25
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
MINIMUM ST
A
R
TUP BA
TTER
Y
V
O
L
T
AGE, V
BA
T
T
/V
Figure 2. Reference Voltage vs. Output Current
Figure 3. Reference Voltage vs. Voltage at OUT Pin
Figure 4. Reference Voltage vs. Temperature
Figure 5. Switch ON Resistance vs. Temperature
Figure 6. L
X
Switch Max. ON Time vs. Temperature
Figure 7. Minimum Startup Battery Voltage vs.
Loading Current
1.180
1.190
1.200
1.210
1.220
1
10
100
1000
V
OUT
= 3.3 V
L = 10
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
C
REF
= 1.0
m
F
T
A
= 25
_
C
AMBIENT TEMPERATURE, T
A
/
C
L
X
SWITCH MAXIMUM, ON TIME, t
ON
/
m
S
OUTPUT CURRENT, I
LOAD
/mA
REFERENCE VOL
T
AGE, V
RE
F
/V
V
IN
= 1.5 V
V
IN
= 2.0 V
V
IN
= 2.5 V
1.180
1.190
1.210
1.220
C
REF
= 200 nF
I
REF
= 0 mA
T
A
= 25
C
VOLTAGE AT OUT PIN, V
OUT
/V
REFERENCE VOL
T
AGE, V
RE
F
/V
V
OUT
= 3.3 V
C
REF
= 200 nF
I
REF
= 0 mA
1.5
2
2.5
3
3.5
4
4.5
5
1.200
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6
TYPICAL OPERATING CHARACTERISTICS
50
60
70
80
90
100
1
10
100
1000
V
IN
= 1.5 V
V
OUT
= 1.8 V
L = 2.2
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
50
60
70
80
90
100
1
10
100
1000
V
IN
= 1.5 V
V
OUT
= 5.0 V
L = 2.2
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
50
60
70
80
90
100
1
10
100
1000
V
IN
= 2.0 V
V
OUT
= 3.3 V
L = 10
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
50
60
70
80
90
100
1
10
100
1000
V
IN
= 2.5 V
V
OUT
= 5.0 V
L = 6.8
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
50
60
70
80
90
100
1
10
100
1000
V
IN
= 2.5 V
V
OUT
= 3.3 V
L = 10
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
50
60
70
80
90
100
1
10
100
1000
Figure 8. Efficiency vs. Load Current
Figure 9. Efficiency vs. Load Current
Figure 10. Efficiency vs. Load Current
Figure 11. Efficiency vs. Load Current
Figure 12. Efficiency vs. Load Current
Figure 13. Efficiency vs. Load Current
V
IN
= 3.3 V
V
OUT
= 5.0 V
L = 12
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
OUTPUT LOADING CURRENT, I
LOAD
/mA
EFFICIENCY/%
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7
TYPICAL OPERATING CHARACTERISTICS
-10
-5
5
10
10
100
1000
V
OUT
= 3.3 V
L = 5.6
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
0
10
20
30
40
50
1.5
1.7
1.9
2.1
2.3
2.5
Figure 14. Output Voltage Change vs. Load
Current
Figure 15. Output Voltage Change vs. Load
Current
Figure 16. Battery Input Voltage vs. Output Ripple
Voltage
Figure 17. Low Battery Detect
Figure 18. No Load Operating Current vs. Input
Voltage at OUT Pin
V
IN
= 2.5 V
V
IN
= 2.0 V
0
OUTPUT LOADING CURRENT, I
LOAD
/mA
OUTPUT VOL
T
AGE CHANGE/%
-10
-5
5
10
10
100
1000
V
OUT
= 5.0 V
L = 5.6
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
V
IN
= 3.3 V
V
IN
= 1.5 V
0
OUTPUT LOADING CURRENT, I
LOAD
/mA
OUTPUT VOL
T
AGE CHANGE/%
V
IN
= 2.5 V
300 mA
BATTERY INPUT VOLTAGE, V
BATT
/V
RIPPLE VOL
T
AGE, V
R
I
PPLE
/mV
p-
p
100 mA
500 mA
V
IN
= 2.5 V
V
OUT
= 3.3 V
L = 6.8
m
H
C
IN
= 22
m
F
C
OUT
= 22
m
F
T
A
= 25
_
C
Upper Trace:
Input Voltage Waveform, 1.0 V/Division
Lower Trace: Output Voltage Waveform, 2.0 V/Division
Figure 19. Startup Transient Response
2.5
5.0
7.5
10
12.5
15
1.5
2.0
2.5
3.0
3.5
5.0
INPUT VOLTAGE AT OUT PIN, V
OUT
/V
NO LOAD OPERA
TING CURRENT
, I
BA
T
T
/
m
A
4.0
4.5
Upper Trace:
Voltage at LBI Pin, 1.0 V/Division
Lower Trace: Voltage at LBO Pin, 1.0 V/Division
V
IN
= 2.5 V
V
OUT
= 5.0 V
I
LOAD
= 10 mA
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8
TYPICAL OPERATING CHARACTERISTICS
(V
IN
= 2.5 V, V
OUT
= 3.3 V, I
LOAD
= 50 mA; L = 5.6
m
H, C
OUT
= 22
m
F)
Upper Trace: Output Voltage Ripple, 20 mV/Division
Lower Trace: Voltage at Lx pin, 1.0 V/Division
Figure 20. Discontinuous Conduction Mode
Switching Waveform
(V
IN
= 2.5 V, V
OUT
= 3.3 V, I
LOAD
= 500 mA; L = 5.6
m
H, C
OUT
= 22
m
F)
Upper Trace: Output Voltage Ripple, 20 mV/Division
Lower Trace: Voltage at LX pin, 1.0 V/Division
Figure 21. Continuous Conduction Mode
Switching Waveform
Figure 22. Line Transient Response for V
OUT
= 3.3 V
Figure 23. Line Transient Response For V
OUT
= 5.0 V
(V
IN
= 1.5 V to 2.5 V; L = 5.6
m
H, C
OUT
= 22
m
F, I
LOAD
= 100 mA)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Battery Voltage, V
IN,
1.0 V/Division
(V
IN
= 1.5 V to 2.5 V; L = 5.6
m
H, C
OUT
= 22
m
F, I
LOAD
= 100 mA)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Battery Voltage, V
IN,
1.0 V/Division
Figure 24. Load Transient Response For V
IN
= 2.5 V
Figure 25. Load Transient Response For V
IN
= 3.0 V
(V
OUT
= 5.0 V, I
LOAD
= 50 mA to 500 mA; L = 5.6
m
H, C
OUT
= 22
m
F)
Upper Trace: Output Voltage Ripple, 100 mV/Division
Lower Trace: Load Current, I
LOAD
, 500 mA/Division
(V
OUT
= 3.3 V, I
LOAD
= 50 mA to 500 mA; L = 5.6
m
H, C
OUT
= 22
m
F)
Upper Trace: Output Voltage Ripple, 50 mV/Division
Lower Trace: Load Current, I
LOAD
, 500 mA/Division
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DETAILED OPERATION DESCRIPTION
NCP1421 is a monolithic micropower high-frequency
step-up voltage switching converter IC specially designed
for battery operated hand-held electronic products up to
600 mA loading. It integrates a Synchronous Rectifier to
improve efficiency as well as to eliminate the external
Schottky diode. High switching frequency (up to 1.2 MHz)
allows for a low profile inductor and output capacitor to be
used. Low-Battery Detector, Logic-Controlled Shutdown,
and Cycle-by-Cycle Current Limit provide value-added
features for various battery-operated applications. With all
these functions ON, the quiescent supply current is
typically only 8.5
mA. This device is available in a compact
Micro8 package.
PFM Regulation Scheme
From the simplified functional diagram (Figure 1), the
output voltage is divided down and fed back to pin 1 (FB).
This voltage goes to the non-inverting input of the PFM
comparator whereas the comparator's inverting input is
connected to the internal voltage reference, REF. A
switching cycle is initiated by the falling edge of the
comparator, at the moment the main switch (M1) is turned
ON. After the maximum ON-time (typically 0.72
mS)
elapses or the current limit is reached, M1 is turned OFF
and the synchronous switch (M2) is turned ON. The M1
OFF time is not less than the minimum OFF-time
(typically 0.12
mS), which ensures complete energy
transfer from the inductor to the output capacitor. If the
regulator is operating in Continuous Conduction Mode
(CCM), M2 is turned OFF just before M1 is supposed to be
ON again. If the regulator is operating in Discontinuous
Conduction Mode (DCM), which means the coil current
will decrease to zero before the new cycle starts, M1 is
turned OFF as the coil current is almost reaching zero. The
comparator (ZLC) with fixed offset is dedicated to sense
the voltage drop across M2 as it is conducting; when the
voltage drop is below the offset, the ZLC comparator
output goes HIGH and M2 is turned OFF. Negative
feedback of closed-loop operation regulates voltage at
pin 1 (FB) equal to the internal reference voltage (1.20 V).
Synchronous Rectification
The Synchronous Rectifier is used to replace the
Schottky Diode to reduce the conduction loss contributed
by the forward voltage of the Schottky Diode. The
Synchronous Rectifier is normally realized by powerFET
with gate control circuitry that incorporates relatively
complicated timing concerns.
As the main switch (M1) is being turned OFF and the
synchronous switch M2 is just turned ON with M1 not
being completely turned OFF, current is shunt from the
output bulk capacitor through M2 and M1 to ground. This
power loss lowers overall efficiency and possibly damages
the switching FETs. As a general practice, a certain amount
of dead time is introduced to make sure M1 is completely
turned OFF before M2 is being turned ON.
The previously mentioned situation occurs when the
regulator is operating in CCM, M2 is being turned OFF, M1
is just turned ON, and M2 is not being completely turned
OFF. A dead time is also needed to make sure M2 is
completely turned OFF before M1 is being turned ON.
As coil current is dropped to zero when the regulator is
operating in DCM, M2 should be OFF. If this does not
occur, the reverse current flows from the output bulk
capacitor through M2 and the inductor to the battery input,
causing damage to the battery. The ZLC comparator comes
with fixed offset voltage to switch M2 OFF before any
reverse current builds up. However, if M2 is switched OFF
too early, large residue coil current flows through the body
diode of M2 and increases conduction loss. Therefore,
determination of the offset voltage is essential for optimum
performance. With the implementation of the synchronous
rectification scheme, efficiency can be as high as 94% with
this device.
Cycle-by-Cycle Current Limit
In Figure 1, a SENSEFET is used to sample the coil
current as M1 is ON. With that sample current flowing
through a sense resistor, a sense-voltage is developed. The
threshold detector (I
LIM
) detects whether the
sense-voltage is higher than the preset level. If the sense
voltage is higher than the present level, the detector output
notifies the Control Logic to switch OFF M1, and M1 can
only be switched ON when the next cycle starts after the
minimum OFF-time (typically 0.12
mS). With proper
sizing of the SENSEFET and sense resistor, the peak coil
current limit is typically set at 1.5 A.
Voltage Reference
The voltage at REF is typically set at 1.20 V and can
output up to 2.5 mA with load regulation
2% at V
OUT
equal to 3.3 V. If V
OUT
is increased, the REF load
capability can also be increased. A bypass capacitor of
200 nF is required for proper operation when REF is not
loaded. If REF is loaded, a 1.0
mF capacitor at the REF pin
is needed.
True-Cutoff
The NCP1421 has a True-Cutoff function controlled by
the multi-function pin LBI/EN (pin 2). Internal circuitry
can isolate the current through the body diode of switch M2
to load. Thus, it can eliminate leakage current from the
battery to load in shutdown mode and significantly reduce
battery current consumption during shutdown. The
shutdown function is controlled by the voltage at pin 2
(LBI/EN). When pin 2 is pulled to lower than 0.3 V, the
controller enters shutdown mode. In shutdown mode, when
switches M1 and M2 are both switched OFF, the internal
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10
reference voltage of the controller is disabled and the
controller typically consumes only 50 nA of current. If the
pin 2 voltage is raised to higher than 0.5 V (for example, by
a resistor connected to V
IN)
, the IC is enabled again, and the
internal circuit typically consumes 8.5
mA of current from
the OUT pin during normal operation.
Low-Battery Detection
A comparator with 30 mV hysteresis is applied to
perform the low-battery detection function. When pin 2
(LBI/EN) is at a voltage (defined by a resistor divider from
the battery voltage) lower than the internal reference
voltage of 1.20 V, the comparator output turns on a 50
W
low side switch. It pulls down the voltage at pin 3 (LBO)
which has hundreds of k
W of pull-high resistance. If the
pin 2 voltage is higher than 1.20 V + 30 mV, the comparator
output turns off the 50
W low side switch. When this occurs,
pin 3 becomes high impedance and its voltage is pulled
high again.
APPLICATIONS INFORMATION
Output Voltage Setting
A typical application circuit is shown in Figure 26. The
output voltage of the converter is determined by the
external feedback network comprised of R1 and R2. The
relationship is given by:
VOUT
+
1.20 V
1
)
R1
R2
where R1
and R2 are the upper and lower feedback
resistors, respectively.
Low Battery Detect Level Setting
The Low Battery Detect Voltage of the converter is
determined by the external divider network that is
comprised of R3 and R4. The relationship is given by:
VLB
+
1.20 V
1
)
R3
R4
where R3
and R4 are the upper and lower divider resistors
respectively.
Inductor Selection
The NCP1421 is tested to produce optimum performance
with a 5.6
mH inductor at V
IN
= 2.5 V and V
OUT
= 3.3 V,
supplying an output current up to 600 mA. For other
input/output requirements, inductance in the range 3
mH to
10
mH can be used according to end application
specifications. Selecting an inductor is a compromise
between output current capability, inductor saturation
limit, and tolerable output voltage ripple. Low inductance
values can supply higher output current but also increase
the ripple at output and reduce efficiency. On the other
hand, high inductance values can improve output ripple
and efficiency; however, it is also limited to the output
current capability at the same time.
Another parameter of the inductor is its DC resistance.
This resistance can introduce unwanted power loss and
reduce overall efficiency. The basic rule is to select an
inductor with the lowest DC resistance within the board
space limitation of the end application. In order to help with
the inductor selection, reference charts are shown in
Figure 27 and 28.
Capacitors Selection
In all switching mode boost converter applications, both
the input and output terminals see impulsive
voltage/current waveforms. The currents flowing into and
out of the capacitors multiply with the Equivalent Series
Resistance (ESR) of the capacitor to produce ripple voltage
at the terminals. During the Syn-Rect switch-off cycle, the
charges stored in the output capacitor are used to sustain the
output load current. Load current at this period and the ESR
combine and reflect as ripple at the output terminals. For
all cases, the lower the capacitor ESR, the lower the ripple
voltage at output. As a general guideline, low ESR
capacitors should be used. Ceramic capacitors have the
lowest ESR, but low ESR tantalum capacitors can also be
used as an alternative.
PCB Layout Recommendations
Good PCB layout plays an important role in switching
mode power conversion. Careful PCB layout can help to
minimize ground bounce, EMI noise, and unwanted
feedback that can affect the performance of the converter.
Hints suggested below can be used as a guideline in most
situations.
Grounding
A star-ground connection should be used to connect the
output power return ground, the input power return ground,
and the device power ground together at one point. All
high-current paths must be as short as possible and thick
enough to allow current to flow through and produce
insignificant voltage drop along the path. The feedback
signal path must be separated from the main current path
and sense directly at the anode of the output capacitor.
Components Placement
Power components (i.e., input capacitor, inductor and
output capacitor) must be placed as close together as
possible. All connecting traces must be short, direct, and
thick. High current flowing and switching paths must be
kept away from the feedback (FB, pin 1) terminal to avoid
unwanted injection of noise into the feedback path.
Feedback Network
Feedback of the output voltage must be a separate trace
detached from the power path. The external feedback
network must be placed very close to the feedback (FB,
pin 1) pin and sense the output voltage directly at the anode
of the output capacitor.
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11
TYPICAL APPLICATION CIRCUIT
LBI/EN
FB
LBO
REF
BAT
GND
LX
OUT
1
2
3
4
8
7
6
5
NCP1421
R4
330 k
R2 200 k
Shutdown
Open Drain
Input
Low Battery
Open Drain
Output
C3
200 nF
R1
350 k
C2
22
m
F
+
V
OUT
=3.3 V
500 mA
C1
22
m
F
V
IN
L
6.5
m
H
Figure 26. Typical Application Schematic for 2 Alkaline Cells Supply
R3
220 k
C4
10 p*
*Optional
GENERAL DESIGN PROCEDURES
Switching mode converter design is considered a
complicated process. Selecting the right inductor and
capacitor values can allow the converter to provide
optimum performance. The following is a simple method
based on the basic first-order equations to estimate the
inductor and capacitor values for NCP1421 to operate in
Continuous Conduction Mode (CCM). The set component
values can be used as a starting point to fine tune the
application circuit performance. Detailed bench testing is
still necessary to get the best performance out of the circuit.
Design Parameters:
V
IN
= 1.8 V to 3.0 V, Typical 2.4 V
V
OUT
= 3.3 V
I
OUT
= 500 mA (600 mA max)
V
LB
= 2.0 V
V
OUT-RIPPLE
= 45 mV
p-p
at I
OUT
= 500 mA
Calculate the feedback network:
Select R2 = 200 k
R1
+
R2
VOUT
VREF
*
1
R1
+
200 k
3.3 V
1.20 V
*
1
+
350 k
Calculate the Low Battery Detect divider:
V
LB
= 2.0 V
Select R4 = 330 k
R3
+
R4
VLB
VREF
*
1
R3
+
300 k
2.0 V
1.20 V
*
1
+
220 k
Determine the Steady State Duty Ratio, D, for typical
V
IN
. The operation is optimized around this point:
VOUT
VIN
+
1
1
*
D
D
+
1
*
VIN
VOUT
+
1
*
2.4 V
3.3 V
+
0.273
Determine the average inductor current, I
LAVG,
at
maximum I
OUT
:
ILAVG
+
IOUT
1
*
D
+
500 mA
1
*
0.273
+
688 mA
Determine the peak inductor ripple current, I
RIPPLE-P,
and calculate the inductor value:
Assume I
RIPPLE-P
is 20% of I
LAVG
. The inductance of the
power inductor can be calculated as follows:
L
+
VIN
tON
2 IRIPPLE
*
P
+
2.4 V
0.75
m
S
2 (137.6 mA)
+
6.5
m
H
A standard value of 6.5
mH is selected for initial trial.
Determine the output voltage ripple, V
OUT-RIPPLE,
and
calculate the output capacitor value:
V
OUT-RIPPLE
= 40 mV
P-P
at I
OUT
= 500 mA
COUT
u
IOUT
tON
VOUT
*
RIPPLE
*
IOUT
ESRCOUT
where t
ON
= 0.75 uS and ESR
COUT
= 0.05
W,
COUT
u
500 mA
0.75
m
S
45 mV
*
500 mA
0.05
W +
18.75
m
F
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12
From the previous calculations, you need at least 18.75
mF in order to achieve the specified ripple level at the
conditions stated. Practically, a capacitor that is one level
larger is used to accommodate factors not taken into
account in the calculations. Therefore, a capacitor value of
22
mF is selected. The NCP1421 is internally compensated
for most applications, but in case additional compensation
is required, the capacitor C4 can be used as external
compensation adjustment to improve system dynamics.
In order to provide an easy way for customers to select
external parts for NCP1421 in different input voltage and
output current conditions, values of inductance and
capacitance are suggested in Figure 27, 28 and 29.
0
2
4
6
8
10
12
14
16
1.4
1.8
2.0
2.2
2.4
2.6
2.8
3.0
Figure 27. Suggested Inductance of V
OUT
= 3.3 V
Figure 28. Suggested Inductance of V
OUT
= 5.0 V
Figure 29. Suggested Capacitance for Output Capacitor
1.6
INPUT VOLTAGE (V)
INDUCT
OR V
ALUE (
m
H)
I
OUT
= 500 mA
0
3
6
9
12
15
18
21
1.6
2.2
2.5
2.8
3.1
3.4
3.7
4.0
1.9
INPUT VOLTAGE (V)
INDUCT
OR V
ALUE (
m
H)
I
OUT
= 500 mA
OUTPUT CURRENT (mA)
CAP
ACIT
OR V
ALUE (
m
F)
CAP
ACIT
OR ESR (m
W
)
V
OUT-RIPPLE
= 45 mV
V
OUT-RIPPLE
= 50 mV
V
OUT-RIPPLE
= 40 mV
25
33
50
100
40
35
30
25
20
15
10
5
0
200
250
300
350
400
450
500
550
600
Table 1. Suggestions for Passive Components
Output Current
Inductors
Capacitors
500 mA
Sumida CR43, CR54,CDRH6D28 series
Panasonic ECJ series
Kemet TL494 series
250 mA
Sumida CR32 series
Panasonic ECJ series
Kemet TL494 series
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PACKAGE DIMENSIONS
Micro8
DM SUFFIX
CASE 846A-02
ISSUE F
8X
8X
6X
mm
inches
SCALE 8:1
1.04
0.041
0.38
0.015
5.28
0.208
4.24
0.167
3.20
0.126
0.65
0.0256
S
B
M
0.08 (0.003)
A
S
T
DIM
MIN
MAX
MIN
MAX
INCHES
MILLIMETERS
A
2.90
3.10
0.114
0.122
B
2.90
3.10
0.114
0.122
C
---
1.10
---
0.043
D
0.25
0.40
0.010
0.016
G
0.65 BSC
0.026 BSC
H
0.05
0.15
0.002
0.006
J
0.13
0.23
0.005
0.009
K
4.75
5.05
0.187
0.199
L
0.40
0.70
0.016
0.028
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSION A DOES NOT INCLUDE MOLD
FLASH, PROTRUSIONS OR GATE BURRS. MOLD
FLASH, PROTRUSIONS OR GATE BURRS SHALL
NOT EXCEED 0.15 (0.006) PER SIDE.
4. DIMENSION B DOES NOT INCLUDE INTERLEAD
FLASH OR PROTRUSION. INTERLEAD FLASH OR
PROTRUSION SHALL NOT EXCEED 0.25 (0.010)
PER SIDE.
5. 846A-01 OBSOLETE, NEW STANDARD 846A-02.
-B-
-A-
D
K
G
PIN 1 ID
8 PL
0.038 (0.0015)
-T-
SEATING
PLANE
C
H
J
L
*For additional information on our Pb-Free strategy and soldering
details, please download the ON Semiconductor Soldering and
Mounting Techniques Reference Manual, SOLDERRM/D.
SOLDERING FOOTPRINT*
NCP1421
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14
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to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any
liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental
damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over
time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under
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may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees,
subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of
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PUBLICATION ORDERING INFORMATION
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Phone: 81-3-5773-3850
NCP1421/D
Micro8 is a trademark of International Rectifier.
SENSEFET is a trademark of Semiconductor Components Industries, LLC.
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