1
1

NR021506

HV9925

Initial Release

Features

Programmable Output Current to 50mA

PWM Dimming / Enable

Universal 85-264VAC Operation

Fixed OFF-Time Buck Converter

Internal 500V Power MOSFET

Over Temperature Protection with Hysteresis

Applications

Decorative Lighting

Low Power Lighting Fixtures

General Description

The HV9925 is a pulse width modulated (PWM) high-effi ciency

LED driver control IC with PWM dimming capabilities. It

allows effi cient operation of high brightness LED strings from

voltage sources ranging up to 400VDC. The HV9925 includes

an internal high-voltage switching MOSFET controlled with a

fi xed off-time T

OFF

of approximately 10祍. The LED string is

driven at constant current, thus providing constant light output

and enhanced reliability. Selecting a value of a current sense

resistor can externally program the output LED current of the

HV9925. The peak current control scheme provides good

regulation of the output current throughout the universal AC

line voltage range of 85 to 264VAC or DC input voltage of

20 to 400V. The HV9925 is designed with a built in thermal

shutdown to prevent excessive power dissipation in the IC.

Typical Application Circuit

Programmable-Current LED Lamp Driver IC

with PWM Dimming

HV9925

LED

BR1

SENSE

Enable

SENSE

GND

Drain

PWMD

NR021506

HV9925

Ordering Information

DEVICE

Package Options

8-Pin SOIC w/ Heat Slug

HV9925

HV9925SG-G

-G indicates package is RoHS compliant (`Green')

Absolute Maximum Ratings*

Symbol

Parameter

Min

Typ

Max

Units

Conditions

Electrical Characteristics

(The

denotes the specifi cations which apply over the full operating temperature range of -40癈 < T

*
*

denotes the specifi cations which apply over the full operating temperature range of -40癈 < T

denotes the specifi cations which apply over the full operating temperature range of -40癈 < T < +85癈, otherwise the specifi cations are at

T
T = 25癈. V

DRAIN

= 25癈. V

= 100V, unless otherwise noted)

Parameter

Value

Supply Voltage, V

-0.3 to +10V

PWMD, R

SENSE

Voltage

-0.3 to +10V

Supply Current, I

+5mA

Operating Ambient Temperature Range

-40癈 to +85癈

Operating Junction Temperature Range

-40癈 to +125癈

Storage Temperature Range

-65癈 to +150癈

Power Dissipation @ 25癈

800mW**

All voltages referenced to GND pin.
**The power dissipation is given for the standard minimum pad without a

heat slug, and based on R

=125癈/W. R

is the sum of the junction-to-

case and case-to-ambient thermal resistance, where the latter is deter-

mined by the user's board design. The junction-to-ambient thermal resis-

tance is R

= 105癈/W when the part is mounted on a 0.04 in

pad of 1 oz

2
2

copper, and R

= 60癈/W when mounted on a 1 in

pad of 1 oz copper.

2
2

regulator output

7.5

UVLO

undervoltage threshold

5.0

UVLO

undervoltage lockout hysteresis

200

Operating supply current

300

500

DD(EXT)

= 8.5V

Output (DRAIN)
V

Breakdown voltage

500

* --

DRAIN

supply voltage

ON resistance

100

200

DRAIN

= 50mA

DRAIN

Output capacitance

1.0

5.0

DRAIN

= 400V

SAT

DRAIN saturation current

100

150

Current Sense Comparator
V

Threshold voltage

0.44

0.47

0.50

BLANK

Leading edge blanking delay

200

300

400

* --

ON(MIN)

Minimum ON time

650

Pin Confi guration

top view

SO-8 + Heat Slug

(Heat Slug Potential is at ground)

SENSE

GND

PWMD

Drain

HV9925SG

DRAIN (6,7,8) � This is a drain terminal of the output switching

MOSFET and a linear regulator input.

(4) � This is a power supply pin for internal control circuits.

Bypass this pin with a 0.1uF low impedance capacitor.

SENSE

(1) � This is a source terminal of the output switching

MOSFET provided for current sense resistor connection.

GND (2) � This is a common connection for all circuits.

PWMD (3) � This is the PWM Dimming input to the IC.

NR021506

HV9925

Typical Performance Characteristics

(T = 25

J
J

C unless otherwise noted)

Threshold Voltage V

vs Temperature T

ON Resistance R

vs Temperature T

OFF Time T

OFF

vs Temperature T

Output Capacitance C

DRAIN

vs V

DRAIN

DRAIN Breakdown Voltage BV vs T

Output Characteristics I

DRAIN

vs V

DRAIN

490

500

510

520

530

540

550

560

570

580

-40

-15

110

Junction Temperature, 癈

A
I
N

B
r
e
a
k
d
o
w

n

V
o
l
t
a
g
e
,

V

100

120

140

160

180

DRAIN Voltage, V

I
N

u
r
r
e
n
t
,

m

100

1000

DRAIN Voltage, V

D
R
A
I
N

C
a
p
a
c
i
t
a
n
c
e
,

p
F

0.460

0.465

0.470

0.475

0.480

0.485

-40

-15

110

Junction Temperature, 癈

u
r
r
e
n
t

S

e
n
s
e

Thre

s
h
o
l
d
,

V

9.0

9.5

10.0

10.5

11.0

11.5

12.0

12.5

13.0

-40

-15

110

Junction Temperature, 癈

F
F

T
i
m

e
,

s

100

120

140

160

180

200

-40

-15

110

Junction Temperature, 癈

e
s
i
s
t
a
n
c
e
,

O

h
m

= 25

= 125

NR021506

HV9925

Functional Description

The HV9925 is a PWM peak current control IC for driving

a buck converter topology in continuous conduction mode

(CCM). The HV9925 controls the output current (rather than

output voltage) of the converter that can be programmed by

a single external resistor (R

SENSE

), for the purpose of driving a

string of light emitting diodes (LED). An external enable input

(PWMD) is provided that can be utilized for PWM dimming of

an LED string. The typical rising and falling edge transitions

of the LED current when using the PWM dimming feature of

the HV9925 are shown in Fig. 6 and Fig. 7.

When the input voltage of 20 to 400V appears at the DRAIN

pin, the internal linear regulator seeks to maintain a voltage

of 7.5VDC at the V

pin. Until this voltage exceeds the

internally programmed under-voltage threshold, no output

switching occurs. When the threshold is exceeded, the

integrated high-voltage switch turns on, pulling the DRAIN

low. A 200mV hysteresis is incorporated with the under-

voltage comparator to prevent oscillation.

When the voltage at R

SENSE

exceeds 0.47V, the switch turns

off and the DRAIN output becomes high impedance. At the

same time, a one-shot circuit is activated that determines

the off-time of the switch (10祍 typ.).

A "blanking" delay of 300ns is provided upon the turn-on of

the switch that prevents false triggering of the current sense

comparator due to the leading edge spike caused by circuit

parasitics.

Application Information

Selecting L1 and D1

The required value of L1 is inversely proportional to the ripple

current I

in it. Setting the relative peak-to-peak ripple to

20~30% is a good practice to ensure noise immunity of the

current sense comparator.

(1)

is the forward voltage of the LED string. T

OFF

is the off-

time of the HV9925. The output current in the LED string (I

)

is calculated then as:

(2)

where V

is the current sense comparator threshold, and

SENSE

is the current sense resistor. The ripple current

introduces a peak-to-average error in the output current

setting that needs to be accounted for. Due to the constant

off-time control technique used in the HV9925, the ripple

current is nearly independent of the input AC or DC voltage

variation. Therefore, the output current will remain unaffected

by the varying input voltage.

Adding a fi lter capacitor across the LED string can reduce

the output current ripple even further, thus permitting a

reduced value of L1. However, one must keep in mind that

the peak-to-average current error is affected by the variation

of T

OFF

. Therefore, the initial output current accuracy might

be sacrifi ced at large ripple current in L1.

Another important aspect of designing an LED driver with

HV9925 is related to certain parasitic elements of the

circuit, including distributed coil capacitance of L1, junction

capacitance, and reverse recovery of the rectifi er diode D1,

capacitance of the printed circuit board traces C

PCB

and output

capacitance C

DRAIN

of the controller itself. These parasitic

elements affect the effi ciency of the switching converter and

could potentially cause false triggering of the current sense

comparator if not properly managed. Minimizing these

parasitics is essential for effi cient and reliable operation of

HV9925.

Coil capacitance of inductors is typically provided in the

manufacturer's data books either directly or in terms of the

self-resonant frequency (SRF).

where L is the inductance value, and C

is the coil capacitance.

L
L

Charging and discharging this capacitance every switching

cycle causes high-current spikes in the LED string. Therefore,

connecting a small capacitor C

(~10nF) is recommended to

bypass these spikes.

Using an ultra-fast rectifi er diode for D1 is recommended to

achieve high effi ciency and reduce the risk of false triggering

of the current sense comparator. Using diodes with shorter

reverse recovery time t

rr
rr

and lower junction capacitance C

achieves better performance. The reverse voltage rating V

of the diode must be greater than the maximum input voltage

of the LED lamp.

The total parasitic capacitance present at the DRAIN output

of the HV9925 can be calculated as:

(3)

When the switch turns on, the capacitance C

is discharged

into the DRAIN output of the IC. The discharge current is

limited to about 150mA typically. However, it may become

lower at increased junction temperature. The duration of the

leading edge current spike can be estimated as:

(4)

V T

OFF

SENSE

1
2

SRF

L C

1 2

)

DRAIN

PCB

SPIKE

SAT

NR021506

HV9925

In order to avoid false triggering of the current sense

comparator, C

must be minimized in accordance with the

following expression:

(5)

where T

BLANK(MIN)

is the minimum blanking time of 200ns, and

IN(MAX)

is the maximum instantaneous input voltage.

The typical DRAIN and R

SENSE

voltage waveforms are shown

in Fig. 3 and Fig. 4.

Estimating Power Loss

Discharging the parasitic capacitance C

into the DRAIN

output of the HV9925 is responsible for the bulk of the

switching power loss. It can be estimated using the following

equation:

(6)

where Fs is the switching frequency and I

SAT

is the saturated

DRAIN current of the HV9925. The switching loss is the

greatest at the maximum input voltage.

The switching frequency is given by the following:

(7)

where is the effi ciency of the power converter.

When the HV9925 LED driver is powered from the full-wave

rectifi ed AC input, the switching power loss can be estimated

as:

(8)

V is the input AC line voltage.

The switching power loss associated with turn-off transitions

of the DRAIN output can be disregarded. Due to the large

amount of parasitic capacitance connected to this switching

node, the turn-off transition occurs essentially at zero-

voltage.

Conduction power loss in the HV9925 can be calculated as:

(9)

where D = V

is the duty ratio, R

is the ON resistance,

is the internal linear regulator current.

When the LED driver is powered from the full-wave

rectifi ed AC line input, the exact equation for calculating the

conduction loss is more cumbersome. However, it can be

estimated using the following equation:

(10)

where V

where V is the input AC line voltage. The coeffi cients K

and K

can be determined from the minimum duty ratio

=0.71Vo/(V

=0.71Vo/(V ).

Fig. 1. Conduction Loss Coeffi cients K

and K

EMI Filter

As with all off-line converters, selecting an input fi lter is critical

to obtaining good EMI. A switching side capacitor, albeit of

small value, is necessary in order to ensure low impedance

to the high frequency switching currents of the converter. As

a rule of thumb, this capacitor should be approximately 0.1-

0.2 礔/W of LED output power. A recommended input fi lter is

shown in Figure 2 for the following design example.

Design Example 1

Let us design an HV9925 LED lamp driver meeting the

following specifi cations:

Input:

Universal AC, 85-264VAC

Output Current: 20mA

Load:

String of 10 LED (LW541C by OSRAM

= 4.1V max. each)

The schematic diagram of the LED driver is shown in

Fig.2.

SAT

BLANK MIN

IN MAX

(

)

(

)

(

)

C V

V I

SWITCH

P IN

IN SAT

OFF

D I

COND

(

)

K I

COND

d DD

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.1

0.2

0.3

0.4

0.5

0.6

0.7

Kd Dm

(

)

Kc Dm

(

)

SWITCH

OFF

SAT

(

)

(

)

NR021506

HV9925

Step 1. Calculating L1.

The output voltage V

= 10

�

41V (max.). Use equation

(1) assuming a 30% peak-to-peak ripple.

Select L1 68mH, I=30mA. Typical SRF = 170KHz. Calculate

the coil capacitance.

Step 2. Selecting D1

Usually, the reverse recovery characteristics of ultra-

fast rectifi ers at I

= 20~50mA are not provided in the

manufacturer's data books. The designer may want to

experiment with different diodes to achieve the best result.

Select D1 MUR160 with V

= 600V, t

rr
rr

20ns (I

= 20mA, I

= 100mA) and C

8pF (V

>50V).

Step 3. Calculating total parasitic capacitance using:

(3)

Step 4. Calculating the leading edge spike duration using:

(4), (5)

Step 5. Estimating power dissipation in HV9925 at 264VAC

using (8) and (10)

Let us assume that the overall effi ciency = 0.7.

Switching power loss:

Minimum duty ratio:

Conduction power loss:

Total power dissipation at V

AC(max)

Total power dissipation at V

Step 6. Selecting input capacitor C

Select C

ECQ-E4104KF by Panasonic (0.1礔, 400V,

Metalized Polyester Film).

Design Example 2

Let us now design a PWM-dimmable LED lamp driver using

the HV9925:

Input:

Universal AC, 85-135VAC

Output Current: 50mA

Load:

String of 12 LED (Power TOPLED

�

OSRAM, V

= 2.5V max. each)

The schematic diagram of the LED driver is shown in Fig.3.

We will use an aluminum electrolytic capacitor for C

in order

to prevent interruptions of the LED current at zero crossings

of the input voltage. As a"rule of thumb", 2~3F per each

watt of the input power is required for C

in this case.

Step 1. Calculating L1.

The output voltage V

= 12 � V

= 30V (max.). Use equation

(1) assuming a 30% peak-to-peak ripple.

Select L1 22mH, I = 60mA. Typical SRF = 270KHz. Calculate

the coil capacitance.

Step 2. Selecting D1

Select D1 ES1G with V

= 400V, t

rr
rr

35ns and C

< 10pF.

Step 3. Calculating total parasitic capacitance using: (3)

Step 4. Calculating the leading edge spike duration using

(4), (5)

Step 5. Estimating power dissipation in HV9925 at 135VAC

using (6), (7) and (9)

Switching power loss:

41 10

0 3 20

�

SRF

KHz

1 2

2 170

(

)

(

)

ns T

SPIKE

BLANK MIN

264

2 31

100

136

(

)

SWITCH

125

SWITCH

(

)

2 10

264

2 100

264

0 7

�

0 71 41

0 7 264

0 16

/( .

)

COND

(

)

0 25 20

210

0 63 200

264

�

TOTAL

125

180

Output Power

41 20

820

10 5

0 3 50

�

SRF

KHz

1 2

2 270

(

)

(

)

ns T

SPIKE

BLANK MIN

135

2 35

100

(

)

kHz

135

2 30

0 7

135

2 10

/ .

�

NR021506

HV9925

Minimum duty ratio:

Conduction power loss:

Total power dissipation in HV9925:

Step 6. Selecting input capacitor C

Select C

3.3礔, 250V.

Figure 3. 85-135VAC LED Lamp Driver with PWM Dimming

AC Line

85-264V

LED

-LED

IN2

VRD1

HV9925

SENSE

Figure 2. Universal 85-264VAC LED Lamp Driver

= 20mA, V

= 20mA, V = 50V) from Example 1

kHz

SWITCH

(

)

(

)

135

2 100

SWITCH

0 7 135

0 23

/( .

)

COND

200

0 7 85

0 5

0 7

(

)

COND

= 217

TOTAL

217

269

Output Power

1 5

AC Line

85-135V

LED

-LED

HV9925

100~200Hz

SENSE

协谢械泻褌褉芯薪薪褘泄 泻芯屑锌芯薪械薪褌: HV9925

协谢械泻褌褉芯薪薪褘泄泻芯屑锌芯薪械薪褌: HV9925