THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Document 600067 Rev 00

T H A T

C o r p o r a t i o n

Pre-trimmed Analog Engine

THAT

4305

FEATURES

Pre-trimmed BlackmerTM VCA &
RMS-level detector

Wide supply voltage range:
±4.5V ~±16V

Low supply current:
3.5 mA typ. (±15V)

Wide dynamic range:
117 dB (VCA)
60 dB (RMS-level detector)

APPLICATIONS

Compressors & Limiters

Gates & Expanders

AGCs

Line-operated dynamics processors

De-Essers

Duckers

Mixers

Level indicators

Companding noise reduction systems

Description

The THAT4305 is a single-chip Analog En-

gine optimized for low-cost applications. It incor-
porates a high-performance Blackmer voltage-
controlled amplifier (VCA) and log-responding
RMS-level sensor. The VCA and RMS detector
are pre-trimmed at wafer stage to deliver low dis-
tortion without further adjustment.

Available only in a small (QSOP) surface-

mount package, the 4305 is aimed at line-oper-
ated audio applications such as compressor/lim-
iters, gates, and other dynamic processors. The
part normally operates from a split supply volt-
age up to ±16Vdc, drawing only 3.5mA at ±15V.
This IC also works at supply voltages as low as
±4.5V, making it useful in some battery-operated
products as well.

The 4305 was developed specifically for use

in low-cost dynamics processors, drawing from
THAT's long history and experience with such
designs. Both VCA control ports and the detec-
tor input and output are available for the de-
signer to connect as s/he sees fit. As a result, the
part is extremely flexible and can be configured
for a wide range of applications including single-
and multi-band companders, digital overload
protectors, voltage-controlled faders, level indi-
cators, etc.

What really sets the 4305 apart from other

manufacturers' offerings is the transparent sound
of its Blackmer VCA, coupled with its accurate
true-RMS level detector. This makes the IC useful
in a wide range of analog audio products.

EC-

EC+

OUT

VCA

OUT

RMS

VCA

RMS

VCA

OUT

EC-

RMS

OUT

EC+

GND

VCC

VEE

Figure 1. THAT4305 equivalent block diagram

Pin Name

Pin Number

No Connection

RMS IN

No Connection

TIME

RMS OUT

GND

Vee

No Connection

EC+

EC-

VCA OUT

No Connection

VCA IN

No Connection

Table 1. THAT 4305 pin assignments

Package

Order Number

16 pin QSOP

4305Q16-U

Table 2. Ordering Information

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Page 2

THAT4305 Pre-trimmed Analog Engine

Operating Temperature Range (T

)

-40 to +85 ºC

Junction Temperature (T

)

-40 to +125 ºC

Power Dissipation (P

) at T

=85 ºC

400mW

Supply Voltages (V

, V

)

18V

VCA Control Voltage

±0.6 V

Storage Temperature Range (T

)

-40 to +125 ºC

Absolute Maximum Ratings

Power Supply

Positive Supply Voltage

Referenced to GND

+4.5

+16

Negative Supply Voltage

Referenced to GND

-4.5

-16

Supply Current

No Signal

=+15V, V

= -15V

3.5

=+15V, V

= -15V

-3.5

-5

=+5V, V

= -5V

=+5V, V

= -5V

-2

Voltage Controlled Amplifier (VCA)

Max. I/O Signal Current

IN(VCA)

+ i

OUT(VCA)

±1.8

peak

VCA Gain Range

-60

+60

Gain at 0V Control

C+

= E

C-

= 0V

-1.0

+1.0

Gain-Control Constant

C+

/Gain (dB)

-60 dB < gain < +60 dB

6.2

mV/dB

Gain-Control Tempco

CHIP

Ref T

CHIP

=27ºC

+0.33

%/ºC

Output Offset Voltage Change

OFF(OUT)

OUT

= 20 k

0 dB gain

+15 dB gain

+30 dB gain

Output Noise

N(OUT)

0 dB gain

22Hz~22kHz, R

OUT

=20 k

-97.5

-95

dBV

Total Harmonic Distortion

THD

= -5dBV, 1kHz, E

C+

= E

C-

= 0V

0.07

0.15

RMS level detector

Output Voltage at Reference i

O(0)

= 7.5

A RMS

-9

Output Error at Input Extremes

O(RMS)error

= 200 nA RMS

±1

±3

= 200

A RMS

±1

±3

Scale Factor Match to VCA

-20 dB < VCA gain < +20 dB

a< i

IN(RMS)

< 100

.95

1.05

Electrical Characteristics

Parameter

Symbol

Conditions

Min

Typ

Max

Units

SPECIFICATIONS

1. If the devices are subjected to stress above the Absolute Maximum Ratings, permanent damage may result. Sustained operation at
or near the Absolute Maximum Ratings conditions is not recommended. In particular, like all semiconductor devices, device reliabil-
ity declines as operating temperature increases.
2. Unless otherwise noted, T

=25ºC, V

=+15V, V

= -15V.

3. Reference is to output offset with -60dB VCA gain.

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

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Document 600067 Rev 00

Page 3

Theory of Operation

The THAT 4305 Dynamics Processor combines

THAT Corporation's proven exponentially controlled

BlackmerTM Voltage-Controlled Amplifier (VCA) and

log-responding RMS-Level Detector building blocks

in a small package optimized for low cost designs.

The part is fabricated using a proprietary, fully com-

plementary, dielectric-isolation process. This pro-

cess produces very high-quality bipolar transistors

(both NPNs and PNPs) with unusually low collector-

substrate capacitances. The 4305 takes advantage of

these devices to deliver wide bandwidth and excel-

lent audio performance while consuming very low

Rectifier Balance

±7.5

A DC

±1

±3

Timing Current

7.5

Filtering Time Constant

3467 X C

TIME

Output Tempco

CHIP

Ref T

CHIP

= 27 ºC

+0.33

%/ºC

Load Resistance

-250mV < V

OUTRMS

< +250mV

Capacitive Load

150

Electrical Characteristics (con't)

Parameter

Symbol

Conditions

Min

Typ

Max

Units

Surface Mount Package

Type

See Fig. 23 for dimensions

16 Pin QSOP

Thermal Resistance

SO package in ambient

105

ºC/W

Thermal Resistance

SO package soldered to board

ºC/W

Environmental Regulation Compliance

Complies with RoHS requirements

Soldering Reflow Profile

JEDEC JESD22-A113-D (250 ºC)

Package Characteristics

Parameter

Symbol

Conditions

Min

Typ

Max

Units

5k1

10u

20k

22p NPO

20k

10u

100p

6k8

VCA Out

VCA In

U1A
THAT4305

RMS In

RMS Out

U1B

THAT4305

+15V

-15V

VCA Out

VCA In

RMS In

RMS Out

Control Voltage

Vcc

Gnd

Vee

Ec-

Ec+

TIME

10u

+15V

Figure 2. Simplified application circuit

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Page 4

THAT4305 Pre-trimmed Analog Engine

current and operating over a wide range of power

supply voltages.

For details of the theory of operation of the VCA

and RMS Detector, we refer the interested reader to

THAT Corporation's data sheets on the 2180-Series

VCAs and the 2252 RMS Level Detector. Theory of

the interconnection of exponentially controlled VCAs

and log-responding level detectors is covered in

THAT Corporation's application note AN101A, The

Mathematics of Log-Based Dynamic Processors.

The VCA - in Brief

The VCA in the 4305 is based on THAT Corpora-

tion's highly successful complementary log-antilog

gain cell topology (the BlackmerTM VCA) as used in

THAT 2180-Series IC VCAs. VCA symmetry is

trimmed during wafer probe for minimum distor-

tion. No external adjustment is allowed.

Input signals are currents in the VCA's VCA

(pin 15). This pin is a virtual ground with a small dc

offset, so in normal operation an input voltage is

converted to input current via an appropriately sized

resistor (R3 in Figure 2). Because the dc current as-

sociated with dc offsets present at the input pin plus

any dc offset in the preceding stages will be modu-

lated by gain changes (thereby becoming audible as

thumps), the input pin is normally ac-coupled. This

blocks such offset currents and reduces dc offset

variation with gain.

The VCA output signal, VCA

OUT

(pin 13), is also

a current, inverted with respect to the input current.

In normal operation, the output current is converted

to a voltage via an external op-amp, where the cur-

rent-to-voltage conversion ratio is determined by the

feedback resistor connected between the op-amp's

output and its inverting input (R2 in Figure 2). The

resulting signal path through the VCA plus op-amp

is noninverting.

The VCA gain is controlled by the voltage applied

between E

C+

(pin 11) and E

C-

(pin 12). Note that any

unused control port should be connected to ground

(as E

C+

is in Figure 2). The gain (in decibels) is pro-

portional to (E

C+

- E

C-

). The constant of proportion-

ality is 6.2 mV/dB for the voltage at E

C+

(relative to

C-

). Note that neither E

C+

or E

C-

should be driven

more than ±0.6 V away from ground.

The VCA's noise performance varies with gain in

a predictable way, but due to the way internal bias

currents vary with gain, noise at the output is not

strictly the product of a static input noise times the

voltage gain commanded. At large attenuation, the

noise floor is usually limited by the input noise of

the output op-amp and its feedback resistor. At 0 dB

gain, the noise floor of ~ -97.5 dBV is the result of

the VCA's output noise current, converted to a volt-

age by the typical 20k I-V converter resistor (R2 in

Figure 2). In the vicinity of 0 dB gain, the noise in-

creases more slowly than the gain: approximately

5 dB noise increase for every 10 dB gain increase.

Finally, as gain approaches 30 dB, output noise be-

gins to increase directly with gain.

While the 4305's VCA circuitry is very similar to

that of the THAT 2180 Series VCAs, there are sev-

eral important differences, as follows.

1. Supply current for the 4305 VCA depends on

the supply voltage. At ±5 V, approximately 800 uA is

available for the sum of input and output signal cur-

rents. This increases to about 1.8 mA at ±15 V.

(Compare this to ~1.8 mA for a 2180 Series VCA

when biased as recommended.)

2. The SYM control port (similar to that on the

2180 VCA) is not brought out to an external pin; it is

driven from an internally trimmed current genera-

tor.

3. The control-voltage constant is approximately

6.2 mV/dB, due primarily to the higher internal op-

erating temperature of the 4305 compared to that of

the 2180 Series.

The RMS Detector - in Brief

The 4305's detector computes RMS level by rec-

tifying input current signals, converting the rectified

current to a logarithmic voltage, and applying that

voltage to a log-domain filter. The output signal is a

dc voltage proportional to the decibel-level of the

RMS value of the input signal current. Some ac com-

ponent (at twice the input frequency plus higher-

order even harmonics) remains superimposed on

the dc output. The ac signal is attenuated by a log

domain filter, which constitutes a single-pole rolloff

with cutoff determined by an external capacitor (C4

in Figure 2).

The rectifier is balanced to within ±3 dB, so a

small amount of fundamental (and higher odd-order

harmonics) ripple can be present at the detector out-

put. By design, this ripple contributes less total rip-

ple than the even-order products that are naturally

and inevitably present at the output of a perfectly

balanced detector.

As in the VCA, input signals are currents to the

RMS

pin (pin 2). This input is a virtual ground, so

a resistor (R1 in Figure 2) is normally used to con-

vert input voltages to the desired current. The level

detector is capable of accurately resolving signals

well below 10 mV (with a 5 k

input resistor). How-

ever, if the detector is to accurately track such

low-level signals, ac coupling (C1 in Figure 2) is re-

quired to prevent dc offsets from causing a dc cur-

rent to flow in the detector's input, which would

obscure low-level ac signal currents.

The log-domain filter cutoff frequency is usually

placed well below the frequency range of interest.

For an audio-band detector, a typical value would be

5 Hz, or a 32 ms time constant (

). The filter's time

constant is determined by an external timing capaci-

tor (C

TIME

) attached to the C

pin (pin 4), and an in-

ternal current source (I

) connected to C

. The

current source is internally fixed at 7.5 µA. The re-

sulting time constant in seconds is approximately

equal to 3467 times the value of the timing capacitor

in Farads. Note that, as a result of the mathematics

of RMS detection, the attack and release time con-

stants are fixed in their relationship to each other.

The RMS detector is capable of driving large

spikes of current into C

TIME

, particularly when the

audio signal input to the RMS detector increases

suddenly. This current is drawn from V

(pin 9),

fed through C

TIME

at pin 4, and returns to the power

supply through the ground end of C

TIME

. If not han-

dled properly through layout and bypassing, these

currents can mix with the audio in the circuit's

ground structure with unpredictable and undesir-

able results. As noted in the Applications section,

local bypassing from the V

pin to the ground end

of C

TIME

is strongly recommended in order to keep

these currents out of the ground structure of the cir-

cuit (see C4 in Figure 2.)

The dc output of the detector is scaled with the

same constant of proportionality as the VCA gain

control: 6.2 mV/dB. The detector's 0 dB reference

in0

, the input current which causes the detector's

output to equal 0V), is trimmed during wafer probe

to equal approximately 7.5 µA. The RMS detector

output stage is capable of sinking or sourcing

125 µA. It is also capable of driving up to 150 pF of

capacitance.

Frequency response of the detector extends

across the audio band for a wide range of input sig-

nal levels. Note, however, that it does fall off at high

frequencies at low signal levels like THAT's other

RMS detectors.

Differences

between

the

4305's

RMS

level

detector circuitry and that of the THAT 2252 RMS

detector include the following.

1. The rectifier in the 4305 RMS Detector is in-

ternally balanced by design, and cannot be balanced

via an external control. The 4305 will typically bal-

ance positive and negative halves of the input signal

within 10 %, but in extreme cases the mismatch may

reach +40 % or -30 % (±3 dB). However, even such

extreme-seeming mismatches will not significantly

increase ripple-induced distortion in dynamics pro-

cessors over that caused by balanced signal ripple

alone.

2. The time constant of the 4305's RMS detector

is determined by the combination of an external ca-

pacitor C

TIME

and an internal current source. The in-

ternal current source is set to about 7.5 µA. A

resistor is not normally connected directly to the C

pin on the 4305.

3. The 0 dB reference point, or level match, is

also set to approximately 7.5 µA. However, as in the

2252, the level match will be affected by any addi-

tional currents drawn from the C

pin.

Compressor (or Limiter) Configurations

The 4305 provides the two essential building

blocks required for a wide variety of dynamics pro-

cessing applications.

The part may be configured

into practically any type of dynamics processor sys-

tem.

Perhaps the most common application for the

4305 is as a compressor or limiter. These circuits

are intended to reduce gain above some determined

signal level in order to prevent subsequent stages

from being overloaded by too high a signal.

Com-

pressors generally have low to moderate compres-

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Document 600067 Rev 00

Page 5

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Page 6

THAT4305 Pre-trimmed Analog Engine

sion ratios, while limiters have high ratios. In such

applications, the signal path has static gain so long

as the input signal remains below some threshold,

but gain is reduced when the signal rises above the

threshold. Compression ratio is defined as the num-

ber of dB the input signal increases for a 1 dB in-

crease in output signal.

Feedforward Topologies

To make a compressor or limiter with a 4305,

typically, the input signal is applied to both the VCA

and the RMS detector. The RMS output signal is fed

forward to the VCA's negative control port (E

C-

) via a

dc-coupled op-amp based stage. This stage has gain

above some dc level (the threshold), and no trans-

mission below that level. This path, called the

"sidechain," -- from detector output to VCA control

port -- determines the compression behavior of the

circuit. As signal level rises, the dc voltage at the

RMS' output rises.

Once the dc level exceeds the

threshold, the rms output signal is transmitted

through the sidechain and presented to the VCA con-

trol port, lowering the gain to signals passing

through the VCA. As a result, the output signal level

is reduced, or compressed, relative to rising input

signal levels.

Varying the threshold setting of the sidechain will

vary the point at which compression begins. Varying

the gain between the RMS output and the VCA con-

trol input varies the compression ratio.

Feedforward compressor topologies are espe-

cially versatile because they cannot become unstable

due to oscillation in the control loop. Unity gain in

the sidechain produces infinite compression (where

the output remains constant regardless of increases

in the input signal). With feedforward, negative com-

pression ratios are easily achievable. (Negative com-

pression occurs when the output signal decreases

as the input signal increases.) This approximates the

effect of playing music backwards, since the attack

is suppressed and the release is increased in vol-

ume.

Many other variations of the feedforward concept

are possible. These include implementing more than

one threshold, different ratios, additional time con-

stants, ac-coupling of some (or all) of the detector

output signal, and many more. See AN101A, The

Mathematics of Log-Based Dynamic Processors, for

more details of how the sidechain gain determines

compression ratios.

Feedback Topologies

An alternative configuration for compressor/lim-

iter design is to feed the output signal into the RMS

detector. The RMS output is fed back (dc coupled)

to the VCA's negative control port to reduce signal

levels.

Similarly as with a feedforward designs, a

threshold in the sidechain serves to stop the com-

pression action at low signal levels.

The feedback topology behaves somewhat differ-

ently from feedforward. First, reaching infinite com-

pression requires infinite gain in the feedback loop

from RMS output to VCA control port. Of course, in-

finite gain is impossible, so practical feedback com-

pressors are usually limited to ratios no greater than

20 or so. Additionally, the gain in the feedback loop

alters the effective time constant of the detector,

shortening the attack as the ratio becomes higher.

This may or may not be appropriate, depending on

the desired effect.

Expander (Gate) Configurations

By changing the sign of the sidechain in a

feedforward compressor, it is possible to arrange

signal gain to decrease along with signal level, thus

producing an expander.

This is typically applied

below a threshold (so, the threshold detector's po-

larity is reversed from that of a compressor) to re-

duce noise or crosstalk during pauses in program

material.

This technique has long been used for

"cleaning up" individual drum tracks to reduce re-

verberation, interference from microphones picking

up adjacent drum sounds, and alter the attack/decay

characteristic of individual drum sounds.

Practical gates usually require very fast attack

times, and carefully programmable release times. In

a 4305, this is best accomplished by using the RMS

detector as a log rectifier with very short time con-

stants, and following the detector output with a

time-constant stage that applies the desired attack

and release behavior. This alters the 4305 detector's

natural response characteristics to peak, rather than

rms, time constants. We intend to produce an appli-

cation note showing examples of these circuits. Un-

til that is available, see DN 100, which shows a noise

gate application using THAT's 4301 Analog Engine.

Noise Reduction (Compander) Configurations

An additional application of the 4305 is for

noise reduction systems. In these applications, one

Analog Engine is configured for use as a compressor

to condition audio signals before feeding them into a

noisy channel. A second Analog Engine, configured

as an expander, is located at the receiver end of the

noisy channel. Most commonly, the compression/ex-

pansion ratio is modest (e.g. 2:1:2) and is linearly

applied across the entire signal dynamic range.

During low-level audio passages, the compressor

increases signal levels, bringing them up above the

noise floor of the channel. At the receiving end, the

expander reduces the signal back to its original

level, in the process attenuating channel noise.

During high-level audio passages, the compres-

sor decreases signal levels, reducing them to fit

within the headroom limits of the channel. The ex-

pander increases the signal back to its original level.

While the channel noise may be increased by this ac-

tion, in a well-designed compander, at such times

the noise floor will be masked by the high-level sig-

nal.

The 4305 facilitates the design of a wide variety

of companding noise reduction systems. The RMS

detector responds accurately over a wide range of

levels; the VCA responds accurately to a wide range

of gain commands; and all the detector and VCA in-

puts, outputs, and control ports are independently

accessible and fully configurable. All these features

mean that the 4305 will support a wide range of

compander designs, including simple 2:1 wide range

(level-independent) systems, level-dependent sys-

tems with thresholds and varying compression

slopes, systems including noise gating and/or limit-

ing,

and

systems

with

varying

degrees

pre-emphasis and filtering in both the signal and de-

tector paths. Furthermore, much of this can be ac-

complished by extensively conditioning the control

voltage sidechain rather than the audio signal itself.

The audio signal can pass through as little as one

VCA and one opamp, and still support multiple ra-

tios, thresholds, and time constants.

Note that the 4305 is fully compatible with other

Analog Engines from THAT Corporation. All our An-

alog Engines feature log-responding true-RMS level

detectors and exponentially controlled Blackmer

VCAs.

It is possible to compress (encode) signals

using the low-voltage, low-power 4315 or 4320 in a

handheld, battery-operated device such as a wireless

microphone or instrument belt pack, and expand

(decode) that signal using the 4305 in a rack-mount,

line-operated receiver.

The Mathematics of Log-Based Dynamics
Processors

At first, the logarithmic output of the RMS detec-

tor and the exponential control ports of the VCA can

be intimidating for designers unfamiliar with THAT

Corporation's offerings.

However, in fact, these

characteristics make developing audio processors

easy once a designer understands the concepts in-

volved. As noted earlier, AN101A: The Mathematics

of Log-Based Dynamics Processors, discusses these

concepts in some detail.

The following discussion

draws heavily from that application note.

The Feedforward Compressor

Figure 3 shows a conceptual diagram of a very

simple feedforward compressor.

Using the "log

math" principles explained in AN101A, we can state

that

Out

, and that

k In

= -

Note that the sign of k makes this a compressor

in which gain G

decreases as input signal level

increases. Combining these equations,

Out

k In

(

)

Rearranging yields

Out

C R

(

)

. .

This is the compression ratio.

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Document 600067 Rev 00

Page 7

-k

Out

RMS

Figure 3. Simplified feedforward compressor,

conceptual diagram.

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By inspection we can see that if k equals zero,

the compression ratio will be 1:1, and if k equals 1,

the compression ratio will be infinity:1.

Thus, we

can make a feedforward compressor/limiter by hav-

ing the gain of the sidechain vary from zero to one.

Note that if k>1, the compression ratio becomes

negative. Negative compression results with 1<k<2.

The Feedback Compressor

We can use the same approach for a feedback

compressor. Figure 4 shows a simplified theoretical

model of a feedback compressor. By inspection,

Out

, and

k Out

= -

. Therefore,

Out

k Out

, and

Out

k Out

. As such,

Out

C R

= + =

. . .

In this case, as mentioned earlier, infinite com-

pression requires infinite sidechain gain.

Fortu-

nately, compression ratios of between 10 and 20

limit effectively enough that infinite gain is not re-

quired.

Log-Based Expanders

Similarly, for the feedforward expander shown in

Figure 5, we can state that

Out

, and that

k In

Note that the sign of k makes this an expander in

which gain G

decreases as input signal level In

decreases. Combining these equations:

Out

k In

, and

Out

k In

= +

(

)

Rearranging yields

Out

E R

= + =

. .

This is the expansion ratio.

Adjusting the Level Match Point

In the equations so far, we have made the im-

plicit assumption that the decibel reference level ev-

erywhere is that of the rms-level detector.

This

assumption simplifies the math, but it may not cor-

respond to real-world reference levels such as 1V

rms (0 dBV) or 0.775V rms (0 dBu). Additionally, it

is possible to offset the VCA's inherent behavior of

producing unity (0 dB) current gain at 0 mV control

voltage (E

C+

-E

C-

) by selecting asymmetrical volt-

age-to-current and current-to-voltage converting re-

sistors (R3 and R2, respectively, in Figure 2).

Figure 6 allows for a VCA voltage gain offset of

, as well as an offset (LM

) to vary the "level

match" point of the RMS detector. Using similar "log

math" from AN101A, we can state for Figure 6:

Out

, where G

is the

VCA's control port gain in dB, and A

is any static

gain or attenuation (in dB) applied to the signal.

We can also state that:

k Out

L M

= -

(

) , where L.M. is a vary-

ing dc voltage intended to change the system's zero

dB reference point (often referred to as level match

Page 8

THAT4305 Pre-trimmed Low-power Analog Engine

RMS

Out

Figure 5. Simplified feedforward expander,

conceptual diagram

Out

RMS

-k

Out

Figure 4. Simplified feedback compressor,

conceptual diagram.

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Document 600067 Rev 00

Page 9

point) from that of the detector, and k is the gain of

the sidechain.

Substituting the second equation into the first,

Out

k L M

Out

( .

)

, or

Out

k L M

k Out

+ ×

- ×

, or

Out

k L M

+ ×

. .

(

)

Similarly, for the expander shown in Figure 7, we

can state that:

Out

k In

L M

(

) . Thus:

Out

k In

k L M

= +

- ×

(

)

In both Figures 6 and 7, we show two ways to ad-

just the point at which input and output levels of the

compressor or expander become equal. These are

a) adjusting the 0 dB reference voltage level of the

RMS detector by setting its voltage-to current con-

verting resistor (R1 in Figure 2), and b) varying the

static signal-path gain offset (A

). For the 4305, the

best practice is to achieve level match via a combina-

tion of setting the RMS detector's voltage-to-current

conversion, and offsetting the signal-path gain. Sig-

nal-path gain can be offset by altering the values of

the VCA's voltage-to-current and current-to-voltage

resistors. Both of these approaches have the advan-

tage of being temperature-independent.

Figure 8 shows the behavior of the compressor

of Figure 6 with varying sidechain gain k for a con-

stant A

and LM

. Note that as k increases, the

compression ratio increases, and the dynamic range

of output levels decreases. Low input signal levels

(including noise) are increased, and high input sig-

nal levels are decreased.

Figure 9 shows how the

same compressor behaves with varying LM

, but

fixed A

and k. Note that as LM

is reduced, all

output levels decrease, and vice-versa.

-120

-100

-80

-60

-40

-20

dB In

-80

-60

-40

-20

L.M.= -10
L.M.= -20
L.M.= -30
L.M.= -40

k=1

Out

Figure 9. Compressor input-output transfer

characteristics with varying LM

-120

-100

-80

-60

-40

-20

dB In

-120

-100

-80

-60

-40

-20

k=0
k=1
k=2
k=3

LM= -40

Out

Figure 8. Compressor input-output transfer

characteristics with varying k.

RMS

-A

Out

L.M.

Figure 7. Feedforward expander with level match.

Out

RMS

-k

Out

L.M.

Figure 6. Feedback compressor with level match

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Figures 10 and 11 show how the input-output

transfer characteristics of the expander of Figure 7

change while varying k (Fig 10) and LM (Fig 11). In

each case, all other parameters remain fixed.

Compressors with Thresholds

The compressor of Figure 6 and expander of

Figure 7 form the primary basis for the linear

companding systems used in many audio applica-

tions. However, they are limited in application, since

the compression ratio is linear over the entire dy-

namic range of the applied signal. While this is fine

for companding systems, it is impractical for an ef-

fects compressor, not least due to the way very

low-level signals -- including noise -- are raised in

gain by the compressor's actions. A more practical

approach to effects compressors is shown in Fig-

ure 12, which offers control over the threshold of

compression and gain offset (or "makeup gain") in

addition to the ratio.

In Figure 12, we've extended this approach to

model a compressor with more of the features one

would expect in an effects compressor/limiter. This

model has a threshold adjustment (T

), a threshold

(set by the ideal diode), a means to vary the

sidechain gain (k), and a make-up gain adjustment

). As with the previous equations, all these vari-

ables are expressed in decibels, in keeping with the

simple "log math" of AN101A.

As with all the previous figures,

Out

For input signals above the level determined by

the threshold setting,

So, for signals above this level,

k In

L M T A

= -

(

. .)

Substituting yields

Out

k In

L M T A

(

. .)

For input signals below the level set by the

threshold setting, the signal at the output of the

ideal diode threshold is 0 (dB), so

, thus,

Out

In the circuit of Figure 12, static gain offset is ap-

plied via a dc voltage summed into the sidechain

and applied to the VCA control port. This illustrates

an alternative method of varying VCA gain (different

from offsetting the V-I and I-V converting resistors as

mentioned earlier).

This is especially convenient

when the level match must be varied by the user, as

with a front-panel control. Note, however, that since

the VCA gain scale factor varies slightly with temper-

ature (+0.33%/ºC), the level match point will vary

Page 10

THAT4305 Pre-trimmed Low-power Analog Engine

-60

-40

-20

dB In

-140

-100

-60

-20

100

140

k=0
k=1
k=2
k=3

Out

Figure 10. Expander input-output transfer

characteristics with varying k.

-60

-50

-40

-30

-20

-10

dB In

-120

-100

-80

-60

-40

-20

L.M.= -10
L.M.= -20
L.M.= -30
L.M.= -40

Out

Figure 11. Expander input-output transfer

characteristics with varying LM

RMS

Out

-k

Ideal

Diode

Figure 12. Feedforward compressor with threshold (T), gain

(A), and ratio (k) adjustments

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

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Document 600067 Rev 00

Page 11

slightly with temperature unless the applied voltage

is appropriately compensated.

If we let T

=20, A

=0 dB, and k=0.75, this be-

havior yields the transfer function shown in Fig-

ure 13. As predicted by the above equations, this

results in a 4:1 compression ratio above the thresh-

old of -20 dB (relative to the RMS detector's 0 dB

reference level. The output level increases by 10 dB

over a 40 dB change in input level.

In Figure 14, we have again extended the basic

feedback compressor with a threshold adjustment

), a threshold (the ideal diode), a means to vary

the sidechain gain (k), and a make-up gain adjust-

ment (A

) just as in Figure 8.

However, in this

case, the detector level is based on the compressor's

output.

Once again we start with

Out

For output signals above the level determined by

the threshold setting,

Out

So for signals above the threshold,

k Out

= -

(

)

Substituting yields

Out

k Out

(

)

which can be reduced to

Out

(

)

For output signals below the level set by the

threshold setting, the signal at the output of the

ideal diode threshold is 0 (dB), so

, and

Out

If we let T

=10, A

=20, and k=10, this behav-

ior yields the transfer function shown in Figure 15.

The compression ratio of 11:1 allows a rise of only

about 4.5 dB over a 50 dB range.

-100

-50

dB Out

-100

-50

Out

Figure 15. Transfer function of a feedback compressor

Out

RMS

-k

Out

Ideal

Diode

Figure 14. Feedback compressor with threshold, gain, and

ratio adjustments

-100

-80

-60

-40

-20

dB In

-100

-90

-80

-70

-60

-50

-40

-30

-20

-10

Out
In

Figure 13. Transfer function of a feedforward compressor

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

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Applications

In this datasheet, we will show detailed circuits

for the 4305 in a relatively simple above-threshold

compressor, and in two simple companding sys-

tems.

As mentioned above, many other configura-

tions of the 4305 are possible. THAT intends to

publish additional circuits in forthcoming applica-

tions notes. Please check with THAT's applications

engineering department to see if your application

has been covered yet, and for personalized assis-

tance with specific designs.

Feedforward Compressor/Limiter

The circuit in Figure 16 shows a typical hard-

knee, feedforward compressor/limiter.

In addition

to compression ratio, the sidechain includes con-

trols for threshold and make-up gain as well.

The Signal Path

The input of the VCA (pin 15) is a virtual ground,

and R14 converts the input signal into a current

flowing into the VCA.

The maximum total signal

current, (I

+ I

OUT

) is 1.8 mA with ±15 V supplies,

so R14 is sized to keep the maximum current at

unity gain to below this level. With peak input volt-

age swing limited by the ±15 V supply rails, the

20 k

resistor at R14 limits maximum i

to about

750 µA. At 0 dB gain, this will cause the same cur-

rent at the VCA output, which is converted back to

15 V by U2 and R13 (also 20 k

). C2 provides AC

coupling, required to block any DC currents that

might otherwise flow into the VCA input. This pre-

vents changes in gain from modulating this current,

which could produce audible "thumps".

The compensation circuit of R28 and C16 is re-

quired for stability. The VCA must "see" a source im-

pedance no greater than 5 k

above 1 MHz. R28 in

parallel with R14 accomplishes this. C16 is chosen

to prevent the added noise gain of the lower source

impedance from increasing noise within the audio

band.

Note that such compensation is unnecessary

when

the

voltage-to-current

converting

resistor

(R14) is 5 k

or less. For example, if the input sig-

nal were limited to lower voltages, the input volt-

age-to-current converting resistor (R14) could be

reduced in value, possibly eliminating the need for

R28 and C16.

U2, along with C4 and R13 forms a trans-

impedance amplifier that converts the VCA's output

current into a voltage. C4 prevents the VCA's output

capacitance from destabilizing the op-amp in this

configuration.

Page 12

THAT4305 Pre-trimmed Low-power Analog Engine

33k

10k

430k

10k

R10

10k

Threshold

Increase

R11

10k

Compression

Ratio

10u

R13

20k

22p NPO

R14

20k

10u

R17

5k1

R18

5k1

100n

620k

R12

10k

Gain

Increase

C13

22u

C15

22p

1N4148

-40 dBu

20 dBu

+20 dB

-20 dB

1:1 CR

INF:1 CR

+15V

C16

100p

R28

6k2

-15V

VCA Out

VCA In

U1A

THAT4305

RMS In

RMS Out

U1B

THAT4305

22p

+15V

-15V

Out

Vee

Gnd

EC+

Vcc

EC-

TIME

Figure 16. 4305-based feedforward compressor

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Document 600067 Rev 00

Page 13

The Sidechain

As noted earlier, for most effects compressors, it

is best not to linearly compress the entire dynamic

range of the audio signal.

To this end, in the

sidechain of figure 16, we have added a threshold

amplifier with ±30 dB of threshold adjustment.

This restricts compression to signals above the

threshold, passing those below threshold without

any change. Additionally, we added a compression

ratio control adjustable from 1:1 to

:1.

Finally,

we've added a means to apply up to ±20 dB of

static, or "make-up" gain.

The input signal is fed into the RMS detector

through C1 and R1. Like the VCA, the input to the

detector is AC coupled. This prevents any DC cur-

rent flowing into the detector's input from being

measured by the detector.

If unchecked, such off-

sets would limit resolution at low levels, producing

an artificial "floor" to the detector's low-level re-

sponse.

As previously discussed (in the theory section),

the output of the detector is proportional to the log

of the RMS level of the input voltage. The output of

the detector will be approximately zero volts at the

"zero dB reference level" -- the point at which the

RMS value of the input current equals the timing

current (set at 7.5 µA for the 4305). We have chosen

a value of -10dBu, (245 mV

RMS

) for the zero dB ref-

erence level. The required input resistor can be cal-

culated as

RMS

245

7 5

32 6

Inverting threshold amplifier (U3) provides gain

of approximately -2 to the detector output signal

above threshold, and zero gain (A

=0) to signals be-

low threshold. The change in gain is accomplished

by D1 and D2, which allow negative-going output

signals to pass but block positive-going ones.

Be-

cause U3 is configured to invert, positive-going sig-

nals at the RMS output (indicating increasing ac

input levels) are passed onwards, while nega-

tive-going RMS outputs are blocked.

By feeding

variable dc into this stage via R7 and the threshold

pot R10, we can vary the point at which RMS output

signals begin to be passed through to the threshold

amplifier stage's output (at the junction of D2 and

R2.)

The scaling at the output of the detector is

+6.2 mV/dB, but because R2 is approximately twice

R17, the threshold amplifier (U3) has a gain of -2, so

the scaling at its output is -12.4 mV/dB. To swing

the threshold over ±30 dB, we can calculate the re-

quired value of R7 as follows:

0 0124

(

)

We can rearrange this to be

0 0124

, and therefore,

403 2

We chose 430 k

for R7.

U4 is a variable-gain inverter that serves to

buffer the VCA's control port, ensuring a low-

impedance drive at that point.

(High impedances,

even as little as 50 to 100 ohms, will increase VCA

distortion at high signal levels.)

Above threshold,

when U4's gain is -½, the net gain of the sidechain

(from RMS output to VCA control input) is unity,

and the compression ratio is

:1. The network of

R3, R9, and R11 in conjunction with R18, allows the

gain of U4 to vary from 0 to -1/2, and simultaneously

shapes the (linear) pot's response so that 50% rota-

tion results in 4:1 compression. 4:1 ratio at 50% ro-

tation is often considered a useful target. Note that

with the more conventional approach of connecting

the undriven end of the pot to ground, the compres-

sion ratio at 50% rotation would be 2:1.

Finally, R8 and R12 provide the means for add-

ing static, or "make-up" gain.

The control-voltage

sensitivity at the output of U4 is 6.2 mV/dB. There-

fore,

0 0062

5 1

(

)

We can rearrange this to be

5 1

0 0062

, and therefore,

5
0 0062

625

We've chosen 620 k

for R8 since it is the near-

est 5% value.

The signals in the sidechain, and at the output of

U4, are generally relatively slow moving, so the

sidechain does not usually require wide bandwidth.

Furthermore, noise on the VCA control port can

modulate the VCA signal, thus adding noise to the

signal path. Accordingly, we added C5 in order re-

duce the noise gain of U4 at high frequencies, which

slightly reduces the high-frequency noise floor of the

entire circuit. C5 may be omitted for non-critical ap-

plications.

As described in the Theory of Operation section

("The RMS Detector - In Brief"), the RMS detector is

capable of driving large spikes of current into the av-

eraging capacitor C

TIME

. To prevent these currents

from upsetting circuit grounds, it is necessary to by-

pass V

to a point very near the grounded end of

TIME

with a capacitor equal to or greater than the

value of C

TIME

. This is C13 in Figure 16. The

grounded ends of these two capacitors should be

connected together before being tied to the rest of

the ground system. Doing so will ensure that the

current spikes flow within the local loop consisting

of the two capacitors, and stay out of the ground sys-

tem.

Companding Systems

The Encoder

Figure 17 shows the 4305 configured as a simple

2:1 encoder or feedback compressor. The encoder in

a companding system is positioned before the noisy

channel (wireless link, storage system, etc.). The

static gain of this circuit is 1, or zero dB, and a

5.1 k

resistor (R3) along with a 220 pF capacitor

(C4) comprise the compensation network is re-

quired to keep the VCA's input amplifier stable for

all gains.

Since the RMS detector output is tied directly to

the VCA's E

C-

, the compression ratio will be 2:1.

Note that the use of the negative-sense control port,

C-

, makes this circuit a compressor. The RMS de-

tector timing capacitor is set for a release rate of

-125 dB per second by using a value of 10 uF.

As described in the Theory of Operation section

("The RMS Detector - In Brief"), the RMS detector is

capable of driving large spikes of current into the av-

eraging capacitor C

TIME

. To prevent these currents

from upsetting circuit grounds, it is necessary to by-

pass V

to a point very near the grounded end of

the C

TIME

with a capacitor (C5) equal to or greater

than the value of C

TIME

. The grounded ends of these

two capacitors should be connected together before

being tied to the rest of the ground system. Doing so

will ensure that the current spikes flow within the

local loop consisting of the two capacitors, and stay

out of the ground system.

The output of the RMS detector is zero volts

when the RMS input current is equal to the timing

current (internally set to ~7.5 mA). A voltage level

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Page 14

THAT4305 Pre-trimmed Analog Engine

22u

+15V

VCA Out

VCA In

U1A

THAT4305

RMS In

RMS Out

U1B
THAT4305

22u

10u

22u

22p

20k

5k1

20k

220p

U2A

Op-Amp

OUTPUT

INPUT

EC+

EC-

Figure 17. 4305 simple compander circuit - 2:1 encoder (compressor)

of -28.5 dBV was chosen as the desired zero dB ref-

erence. The RMS detector's input resistance can be

calculated as:

RMS in

7 5

28 5

5 1

This value also applies to the decoder.

The Decoder

Figure 18 shows the THAT4305 configured as a

1:2 expander intended to complement the encoder

in Figure 17.

This circuit also uses a static gain of zero dB.

Since the VCA is not stable unless it sees a high fre-

quency source impedance of 5 k

or less, the com-

pensation network of R13 and C21 ensures stability.

In this instance, the RMS detector output is con-

nected to E

C+

; this reverses the polarity of the con-

trol signal relative to the encoder, and makes this

circuit a 2:1 expander.

System Performance

Table 3 shows the transfer characteristics of this

companding system. The columns labeled Encoder

VCA Gain, Encoder Out, Decoder VCA Gain, and De-

coder Out use the equations derived previously in

the Theory sub-section entitled "The Mathematics of

Log Based Companding Systems". The values in the

column labeled RMS In are derived using the equa-

tion:

RMS In

Encoder Out

RMS In

(

)

The required encoder VCA gain range is -24 dB

to +36 dB, and the required decoder VCA gain

range is -36 dB to +14 dB. These gains are easily

within the capabilities of the 4305's VCA. The range

of RMS input currents is easily accommodated at

the high end, though accuracy may be slightly com-

promised at the lowest input levels.

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Document 600067 Rev 00

Page 15

C14

22u

+15V

VCA Out

VCA In

U3A
THAT4305

RMS In

RMS Out

U3B
THAT4305

C11

22u

C13

10u

C12

22u

C15

47p

20k

R10

5k1

20k

U4A

Op-Amp

INPUT

OUTPUT

R13
5k1

C21

220p

EC+

EC-

Figure 18. 4305 Typical Application Circuit - 1:2 Expander

Encoder

Encode

VCA

Gain

Encoder

Out/

Decoder

IRMS In

Decoder

VCA

Gain

Decoder

Out

(dBV)

(In dB)

(dBV)

(mA)

(In dB)

(dBV)

+20

-24

-4

0.1223

+10

-19

-9

0.0688

-14

0.0387

-10

-9

-19

0.0218

-10

-20

-4

-24

0.0122

-20

-30

-29

0.0069

-1

-30

-40

-34

0.0039

-6

-40

-50

-39

0.0022

-11

-50

-60

-44

0.0012

-16

-60

-70

-49

0.0007

-21

-70

-80

-54

0.0004

-26

-80

-90

-59

0.0002

-31

-90

-100

-64

0.0001

139

Table 3. 2:1 compander transfer characteristics

Figure 19 tracks signal levels through the en-

coder and decoder of Figures 17 and 18. The en-

coder reduces the dynamic range at its input by a

factor of 2, compressing 120dB into 60dB. The de-

coder expands this dynamic range back to track that

of the encoder's input signal.

Hi-fi Compander

While the previous circuits perform adequately

in some applications, a few minor changes can re-

sult in substantially improved overall performance.

The following compander implementation adds pre-

and de-emphasis to the signal path.

Signal path

pre-emphasis helps overcome the rising noise level

with frequency of an FM RF channel by raising the

level of the high frequency portions of the signal be-

fore it passes through the transmission channel.

Matching signal-path de-emphasis in the decoder

brings the frequency response back to flat while si-

multaneously lowering the noise floor of the chan-

nel.

This helps ensure that isolated low-frequency

signals mask the channel noise by reducing the per-

ception of high-frequency noise signals.

course,

the

drawback

signal-path

pre-emphasis is that it can cause overload in the

channel when high-level, high-frequency signals are

present. To guard against this problem, we have

added RMS pre-emphasis to both detectors.

This

mitigates high-frequency overload by lowering the

level-match point to high-frequency signals.

For a

given signal level, high-frequency signals are lowered

in level by the VCA more than low-frequency signals.

As an additional enhancement, we have included

a means to truncate the RMS detector's low-level

response. This improves low level tracking between

different detectors by forcing each detector to "bot-

tom out" at a predetermined level, eliminating the ef-

fects

different

low-level

behavior

from

one

detector to the next.

System Performance

The compander shown in Figures 20 and 21 im-

plements all of the aforementioned improvements.

Assuming

change

VCA

gain

the

pre-emphasis network of R3 and C7 produces

~20 dB of signal-path pre-emphasis starting at

~2 kHz and stopping at ~19 kHz.

Note that R3 and C7 also compensate the input

to the VCA, so additional components are not re-

quired to implement this feature. Signal fed to this

network is buffered by U2; while this buffer is not

always necessary, the pre-emphasis network must

be driven from a low source impedance to ensure

proper tracking between the encoder pre-emphasis

and the decoder de-emphasis. If driven from an

unbuffered

source,

the

pre-emphasis

network

should be adjusted to take into account the imped-

ance of that source.

We have included ~10 dB of RMS pre-emphasis

(provided by R5 and C8 in the encoder, and R11 and

C18 in the decoder) for the detectors in both the en-

coder and the decoder. The center frequency of this

pre-emphasis circuit is aligned with the center fre-

quency of the signal path pre-emphasis when evalu-

ated on a logarithmic frequency scale. This shifts the

level match of the encoder symmetrically about the

mid-point of the signal-path pre-emphasis, which

configures the system to take the best advantage of

the companding to avoid high-level high-frequency

overload in the transmission or storage channel.

R6 of the Hi-Fi encoder and R12 of the decoder

are intended to force each of the detectors to stop

responding to low level signals at the same point in

order to improve tracking. This floor occurs when

the RMS current through R1 equals that of R3, and

when the current through R10 equals that of R12.

Since the input of the RMS detector is at virtual

ground, the current through R3 and R12 will be

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Page 16

THAT4305 Pre-trimmed Analog Engine

-100

-80

-60

-40

-20

20
0
-20
-40

-60
-80
-100

Compression

Process

Expansion

Process

In(Cmp)

Out Exp

Out Cmp

In(Exp)

Figure 19. 2:1 compander transfer characteristics

RMSOut

, and

RMSOut

We'll choose a point about 4dB above the mini-

mum input level which is -56dBV.

RMS

dBV

316

(

)

. Therefore,

0 00316

8 87

23 6

52 0 0062

(

)) .

- -

, And

459

464

23 6

52 0 0062

0 00316

8 87

- -

(

)) .

The same is true for R12.

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Document 600067 Rev 00

Page 17

VCA Out

VCA In

U3A
THAT4305

RMS In

RMS Out

U3B
THAT4305

C11

C13
10u

C12

C15

22p

21k0

R10

8k87

84k5

R11

4k02

C17

3n3

C18

3n3

C14
22u

Op-Amp

V+

Decode In

Decode Out

2k32

R12

464k

R13
5k1

C21
220p

EC+

EC-

Signal Path

De-emphasis

RMS Pre-emphasis

Figure 21. 4305 hi-fi 2:1 decoder circuit

VCA Out

VCA In

U1A

THAT4305

RMS In 2

RMS Out

U1B
THAT4305

10u

22p

21k0

2k32

8k87

84k5

4k02

3n3

22u

Op-Amp

Vcc

464k

Encode Out

Encode In

EC+

EC-

Signal Path

Pre-emphasis

RMS

Pre-emphasis

Figure 20. 4305 hi-fi 2:1 encoder circuit

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Table 4 shows the transfer characteristics of this

companding system (neglecting the effects of R6 and

R12). As before, the columns labeled Encoder VCA

Gain, Encoder Out, Decoder VCA Gain, and Decoder

Out use the equations derived previously in the sec-

tion

titled

"The

Mathematics

Log

Based

Companding Systems". The values in the column la-

beled RMS In are derived using the equation:

RMS In

Encoder Out

RMS In

(

)

Figure 22 tracks signal levels through the en-

coder and decoder of Figures 20 and 21. The com-

pression and expansion ratios here are the same as

those of the previous circuits, but the frequency

shaping afforded by signal pre- and de-emphasis

and detector pre-emphasis make this a superior

sounding system.

In this application, the VCA gain ranges over

about ±30 dB, which is well within specification, as

is the RMS detector input current.

Other Dynamics Processor Configurations

We have said before that the building blocks con-

tained within the 4305 are applicable to a very wide

range of dynamics processor configurations. These

include companding noise reduction systems with

ratios other (higher or lower) than 2:1:2, multi-band

companders, etc. The 4305 is versatile enough to be

used as the heart of a compressor, expander, noise

gate, AGC, de-esser, frequency-sensitive compres-

sor, and many other dynamics processors. It is be-

yond the scope of this data sheet to provide specific

advice about any of these functional classes. We re-

fer the interested reader to THAT's applications

notebooks volumes 1 and 2, which contain many cir-

cuits based on THAT's other VCAs and RMS level

detectors, but are largely applicable to the 4305 with

only minor variations. Of course, look for more ap-

plications information aimed specifically at the 4305

in the future.

Closing Thoughts

The

design

dynamics

processors

and

companding systems is a very intricate art: witness

the proliferation of dynamics processors available in

the market today.

Many of these are based on

THAT's VCAs and level detectors, yet they all have

individual sonic characteristics. In the applications

section of this data sheet, we have offered a few ex-

amples only as starting points. THAT Corporation's

applications engineering department is ready to as-

sist customers with suggestions for tailoring and ex-

tending these basic circuits to meet specific needs.

Page 18

THAT4305 Pre-trimmed Analog Engine

-100

-80

-60

-40

-20

20
0
-20

-40
-60
-80

-100

Compression

Process

Expansion

Process

In(Cmp)

Out Exp

Out Cmp

In(Exp)

Figure 22. Hi-fi compander transfer characteristics

Encoder

VCA Gain

Encoder

Out/

Decoder

IRMS In

Decoder

VCA Gain

Decoder

Out

(dBV)

(In dB)

(dBV)

(mA)

(In dB)

(dBV)

-28

0.1841

-23

-1

0.1035

-18

-6

0.0582

-10

-13

-11

0.0327

-10

-20

-8

-16

0.0184

-20

-30

-3

-21

0.0104

-30

-40

-26

0.0058

-2

-40

-50

-31

0.0033

-7

-50

-60

-36

0.0018

-12

-60

-70

-41

0.0010

-17

-70

-80

-46

0.0006

-22

-80

-90

-51

0.0003

-27

-90

-100

-56

0.0002

-32

-100

Table 4. Hi-fi compander transfer characteristics

THAT Corporation; 45 Sumner Street; Milford, Massachusetts 01757-1656; USA

Tel: +1 508 478 9200; Fax: +1 508 478 0990; Web: www.thatcorp.com

Document 600067 Rev 00

Page 19

Package and Soldering Information

The THAT 4305 is available in a 16-pin QSOP

package. The package dimensions are shown in Fig-

ure 23 below, while the pinout is given in Table 1 on

page 1.

The 4305 is available only in a lead-free, "green"

package. The lead frame is copper, plated with suc-

cessive layers of nickel palladium, and gold. This

approach makes it possible to solder these devices

using lead-free and lead-bearing solders. The plastic

mold compound, and the material in which the

parts are packaged, contains no hazardous sub-

stances as specified in the RoHS directive. For more

information, including MDDS forms which disclose

the substances contained in our ICs and their pack-

aging, please visit

www.thatcorp.com/RoHShome.html.

The package has been qualified using reflow tem-

peratures as high as 260°C for 10 seconds. This

makes them suitable for use in a 100% tin solder

process. Furthermore, the 4305 has been qualified

to a JEDEC moisture sensitivity level of MSL1. No

special humidity precautions are required prior to

flow soldering the parts.

0-8º

ITEM

MILLIMETERS

INCHES

4.80 - 4.98

0.189 - 0.196

3.81 - 3.99

0.150 - 0.157

5.79 - 6.20

0.228 - 0.244

0.20 - 0.30

0.008 - 0.012

0.635 BSC

0.025 BSC

1.35 - 1.75

0.0532 - 0.0688

0.10 - 0.25

0.004 - 0.010

0.40 - 1.27

0.016 - 0.050

0.19 - 0.25

0.0075 - 0.0098

Figure 23. QSOP-16 surface mount package drawing

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