OPA2822

Dual Wideband, Low-Noise
OPERATIONAL AMPLIFIER

FEATURES

LOW INPUT NOISE VOLTAGE: 2.0nV/

HIGH UNITY GAIN BANDWIDTH: 500MHz

HIGH GAIN BANDWIDTH PRODUCT: 240MHz

HIGH OUTPUT CURRENT: 90mA

SINGLE +5V TO +12V OPERATION

LOW SUPPLY CURRENT: 4.8mA/ch

APPLICATIONS

xDSL DIFFERENTIAL LINE RECEIVERS

HIGH DYNAMIC RANGE ADC DRIVERS

LOW NOISE PLL INTEGRATORS

TRANSIMPEDANCE AMPLIFIERS

PRECISION BASEBAND I/Q AMPLIFIERS

ACTIVE FILTERS

DESCRIPTION

The OPA2822 offers very low 2.0nV/

Hz input noise in a

wideband unity gain stable voltage-feedback architecture.
Intended for xDSL receiver applications, the OPA2822 also
supports this low input noise with exceptionally low harmonic
distortion, particularly in differential configurations. Adequate
output current is provided to drive the potentially heavy load
of a passive filter between this amplifier and the codec.
Harmonic distortion for a 2Vp-p differential output operating
from +5V to +12V supplies is

100dBc through 1MHz input

frequencies. Operating on a low 4.8mA/ch supply current,
the OPA2822 can satisfy all xDSL receiver requirements
over a wide range of possible supply voltages--from a single
+5V condition, to

5V, on up to a single +12V design.

General-purpose applications on a single +5V supply will
benefit from high input and output voltage swing available on
this reduced supply voltage. Low-cost precision integrators
for PLLs will also benefit from the low voltage noise and
offset voltage. Baseband I/Q receiver channels can achieve
almost perfect channel match with noise and distortion to
support signals through 5MHz with > 14-bit dynamic range.

OPA2822 RELATED PRODUCTS

FEATURES

SINGLES

DUALS

TRIPLES

High Slew Rate

OPA680

OPA2680

OPA3680

R/R Input/Output

OPA353

OPA2353

1.3nV Input Noise

OPA686

OPA2686

1.5nV Input Noise

THS6062

xDSL Driver

OPA2677

xDSL Receiver

500

OPA2822

500

n:1

OPA2

822

www.ti.com

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

SBOS188A JANUARY 2002

PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.

OPA2822

SBOS188A

www.ti.com

Supply Voltage .................................................................................

6.5V

Internal Power Dissipation ........................... See Thermal Characteristics
Differential Input Voltage ..................................................................

1.2V

Input Voltage Range ............................................................................

Storage Temperature Range ......................................... 40

C to +125

Lead Temperature (SO-8) ............................................................. +260

Junction Temperature (T

) ........................................................... +175

ESD Rating (Human Body Model) .................................................. 2000V

(Machine Model) ........................................................... 200V

NOTE: (1) Stresses above these ratings may cause permanent damage.
Exposure to absolute maximum conditions for extended periods may degrade
device reliability. These are stress ratings only, and functional operation of the
device at these or any other conditions beyond those specified is not implied.

ABSOLUTE MAXIMUM RATINGS

(1)

PACKAGE/ORDERING INFORMATION

SPECIFIED

PACKAGE

TEMPERATURE

PACKAGE

ORDERING

TRANSPORT

PRODUCT

PACKAGE-LEAD

DESIGNATOR

(1)

RANGE

MARKING

NUMBER

MEDIA, QUANTITY

OPA2822U

SO-8 Surface Mount

C to +85

OPA2822U

Rails, 100

OPA2822U/2K5

Tape and Reel, 2500

OPA2822E

MSOP-8 Surface Mount

DGK

C to +85

D22

OPA2822E/250

Tape and Reel, 250

OPA2822E/2K5

Tape and Reel, 2500

NOTE: (1) For the most current specifications and package information, refer to our web site at www.ti.com.

ELECTROSTATIC
DISCHARGE SENSITIVITY

Electrostatic discharge can cause damage ranging from per-
formance degradation to complete device failure. Texas In-
struments recommends that all integrated circuits be handled
and stored using appropriate ESD protection methods.

ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet published speci-
fications.

Top View

PIN CONFIGURATION / MSOP PACKING MARKING

MSOP PACKAGE MARKING

Out B

In B

+In B

Out A

In A

+In A

OPA2822

D22

OPA2822

SBOS188A

www.ti.com

AC PERFORMANCE (Figure 1)
Small-Signal Bandwidth

G = +1, V

= 0.1Vp-p, R

= 0

400

MHz

typ

G = +2, V

= 0.1Vp-p

200

120

110

105

MHz

min

G = +10, V

= 0.1Vp-p

MHz

min

Gain-Bandwidth Product

240

150

130

125

MHz

min

Bandwidth for 0.1dB Gain Flatness

G = +2, V

< 0.1Vp-p

MHz

typ

Peaking at a Gain of +1

< 0.1Vp-p

typ

Large Signal Bandwidth

G = +2, V

= 2Vp-p

MHz

typ

Slew Rate

G = +2, 4V Step

170

110

105

100

min

Rise-and-Fall Time

G = +2, V

= 0.2V Step

1.5

typ

Settling Time to 0.02%

G = +2, V

= 2V Step

typ

0.1%

G = +2, V

= 2V Step

typ

Harmonic Distortion

G = +2, f = 1MHz, V

= 2Vp-p

2nd Harmonic

= 200

dBc

max

500

dBc

max

3rd Harmonic

= 200

100

dBc

max

500

105

dBc

max

Input Voltage Noise

f > 10kHz

2.0

2.2

2.3

2.5

nV/

max

Input Current Noise

f > 10kHz

1.6

2.0

2.1

2.3

pA/

max

Differential Gain

G = +2, PAL, V

= 1.4Vp, R

= 150

0.02

typ

Differential Phase

G = +2, PAL, V

= 1.4Vp, R

= 150

0.03

deg

typ

Channel-to-Channel Crosstalk

f = 1MHz, Input Referred

dBc

typ

DC PERFORMANCE

(4)

Open-Loop Voltage Gain (A

)

= 0V, R

= 100

100

min

Input Offset Voltage

= 0V

0.2

1.2

1.4

1.5

max

Average Offset Voltage Drift

= 0V

max

Input Bias Current

= 0V

12

max

Average Bias Current Drift (magnitude)

= 0V

nA/

max

Input Offset Current

= 0V

100

400

600

700

max

Average Offset Current Drift

= 0V

nA/

max

INPUT
Common-Mode Input Range (CMIR)

(5)

4.8

4.5

4.4

min

Common-Mode Rejection Ratio (CMRR)

110

min

Input Impedance

Differential-Mode

= 0

0.6

|| pF

typ

Common-Mode

= 0

|| pF

typ

OUTPUT
Voltage Output Swing

No Load

4.9

4.7

4.6

min

100

Load

4.7

4.5

4.4

min

Current Output, Sourcing

= 0, Linear Operation

+150

+90

+85

+80

min

Current Output, Sinking

= 0, Linear Operation

150

90

min

Short-Circuit Current

Output Shorted to Ground

220

typ

Closed-Loop Output Impedance

G = +2, f = 100kHz

0.01

typ

POWER SUPPLY
Specified Operating Voltage

typ

Maximum Operating Voltage Range

6.3

6.3

max

Max Quiescent Current

6V, both channels

9.6

11.2

11.3

11.4

max

Min Quiescent Current

6V, both channels

9.6

8.2

8.1

8.0

min

Power-Supply Rejection Ratio (PSRR)

Input Referred

min

THERMAL CHARACTERISTICS
Specified Operating Range U, N Package

40 to +85

typ

Thermal Resistance,

Junction-to-Ambient

SO-8

125

C/W

typ

MSOP

150

C/W

typ

NOTES: (1) Test Levels: (A) 100% tested at 25

C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.

C tested specifications. (3) Junction temperature = ambient at low temperature limit:

junction temperature = ambient +23

C at high temperature limit for over temperature tested specifications. (4) Current is considered positive-out-of node. V

is the

input common-mode voltage. (5) Tested < 3dB below minimum CMRR specification at

CMIR limits.

ELECTRICAL CHARACTERISTICS: V

= 402

, R

= 100

, and G = +2

(Figure 1 for AC performance only), unless otherwise noted.

OPA2822U, E

TYP

MIN/MAX OVER TEMPERATURE

C to

40

C to

MIN/

TEST

PARAMETER

CONDITIONS

+25

(2)

(3)

+85

(3)

UNITS

MAX

LEVEL

(1)

OPA2822

SBOS188A

www.ti.com

AC PERFORMANCE (Figure 3)
Small-Signal Bandwidth

G = +1, V

= 0.1Vp-p, R

= 0

350

MHz

typ

G = +2, V

= 0.1Vp-p

180

105

102

100

MHz

min

G = +10, V

= 0.1Vp-p

MHz

min

Gain-Bandwidth Product

G > 20

200

130

110

105

MHz

min

Peaking at a Gain of +1

< 0.1Vp-p

typ

Large Signal Bandwidth

G = +2, V

= 2Vp-p

MHz

typ

Slew Rate

G = +2, 2V Step

120

min

Rise-and-Fall Time

G = +2, V

= 0.2V Step

2.0

2.7

3.2

3.3

max

Settling Time to 0.02%

G = +2, V

= 2V Step

typ

0.1%

G = +2, V

= 2V Step

typ

Harmonic Distortion

G = +2, f = 1MHz, V

= 2Vp-p

2nd Harmonic

= 200

to V

/ 2

dBc

max

= 500

to V

/ 2

dBc

max

3rd Harmonic

= 100

to V

/ 2

dBc

max

= 1500

to V

103

dBc

max

Input Voltage Noise

f > 1MHz

2.1

2.3

2.4

2.6

nV/

max

Input Current Noise

f > 1MHz

1.5

1.9

2.0

2.1

pA/

max

DC PERFORMANCE

(4)

Open-Loop Voltage Gain

= 0V, R

= 200

to 2.5V

min

Input Offset Voltage

= 2.5V

0.3

1.3

1.5

1.6

max

Average Offset Voltage Drift

= 2.5V

5.5

max

Input Bias Current

= 2.5V

11

max

Average Bias Current Drift

= 2.5V

nA/

max

Input Offset Current

= 2.5V

100

400

600

700

max

Average Offset Current Drift

= 2.5V

nA/

max

INPUT
Least Positive Input Voltage

1.2

1.5

1.6

1.65

min

Most Positive Input Voltage

3.8

3.5

3.4

3.35

max

Common-Mode Rejection Ratio (CMRR)

= +2.5V

110

min

Input Impedance

Differential-Mode

= +2.5V

|| pF

typ

Common-Mode

= +2.5V

1.3

|| pF

typ

OUTPUT
Most Positive Output Voltage

No Load

3.9

3.8

3.6

3.5

min

= 100

to 2.5V

3.7

3.5

3.4

3.35

min

Least Positive Output Voltage

No Load

1.3

1.4

1.5

1.55

min

= 100

to 2.5V

1.4

1.5

1.6

1.65

min

Current Output, Sourcing

+150

+90

+85

+80

min

Current Output, Sinking

150

90

min

Short-Circuit Current

Output Shorted to Either Supply

200

typ

Closed-Loop Output Impedance

G = +1, f = 100kHz

0.01

typ

POWER SUPPLY
Specified Single-Supply Operating Voltage

typ

Maximum Single-Supply Operating Voltage

12.6

12.6

max

Max Quiescent Current

= +5V, both channels

10.2

10.4

max

Min Quiescent Current

= +5V, both channels

7.2

7.0

6.9

min

Power-Supply Rejection Ratio

Input Referred

typ

THERMAL CHARACTERISTICS
Specified Operating Range U, E Package

40 to +85

typ

Thermal Resistance,

Junction-to-Ambient

SO-8

125

C/W

typ

MSOP

150

C/W

typ

NOTES: (1) Test Levels: (A) 100% tested at 25

C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.

C tested specifications. (3) Junction temperature = ambient at low temperature limit:

junction temperature = ambient +23

C at high temperature limit for over temperature tested specifications.

ELECTRICAL CHARACTERISTICS: V

= +5V

= 402

, R

= 100

to V

/ 2, and G = +2

(Figure 3 for AC performance only), unless otherwise noted.

OPA2822U, E

TYP

MIN/MAX OVER TEMPERATURE

C to

40

C to

MIN/

TEST

PARAMETER

CONDITIONS

+25

(2)

(3)

+85

(3)

UNITS

MAX

LEVEL

(1)

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

NONINVERTING SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

G = +1

= 0

G = +2

G = +5

G = +10

= 0.1Vp-p

See Figure 1

INVERTING SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

G = 1

G = 10

G = 5

= 0.1Vp-p

= 604

See Figure 2

G = 2

NONINVERTING LARGE-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

G = +2

= 0.1Vp-p

= 0.5Vp-p

= 1Vp-p

= 2Vp-p

See Figure 1

INVERTING LARGE-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

G = 1

= 604

= 0.1Vp-p

= 0.5Vp-p

= 1Vp-p

= 2Vp-p

See Figure 2

400

300

200

100

200

300

400

NONINVERTING PULSE RESPONSE

Time (20ns/div)

Small-Signal Output Voltage (100mv/div)

2.0

1.5

1.0

0.5

1.0

1.5

2.0

Larege-Signal Output Voltage (500mv/div)

See Figure 1

Large-Signal Right Scale

Small-Signal Left Scale

G = +2

400

300

200

100

200

300

400

INVERTING PULSE RESPONSE

Time (20ns/div)

Small-Signal Output Voltage (100mv/div)

2.0

1.5

1.0

0.5

1.0

1.5

2.0

Larege-Signal Output Voltage (500mv/div)

See Figure 2

Large-Signal Right Scale

Small-Signal Left Scale

G = 1

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

100

105

HARMONIC DISTORTION vs LOAD RESISTANCE

Load Resistance (

)

100

Harmonic Distortion (dBc)

= 2Vp-p

f = 1MHz

2nd-Harmonic

3rd-Harmonic

See Figure 1

1MHz HARMONIC DISTORTION vs SUPPLY VOLTAGE

Supply Voltage (V)

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Harmonic Distortion (dBc)

100

105

2nd-Harmonic

3rd-Harmonic

See Figure 1

= 2Vp-p

= 200

105

HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

Harmonic Distortion (dBc)

= 2Vp-p

= 200

2nd-Harmonic

3rd-Harmonic

See Figure 1

HARMONIC DISTORTION vs OUTPUT VOLTAGE

Output Voltage Swing (Vp-p)

0.1

Harmonic Distortion (dBc)

100

105

= 200

f = 1MHz

2nd-Harmonic

3rd-Harmonic

See Figure 1

100

110

HARMONIC DISTORTION vs NONINVERTING GAIN

Gain (V/V)

Harmonic Distortion (dBc)

= 2Vp-p

= 200

f = 1MHz

2nd-Harmonic

3rd-Harmonic

See Figure 1

100

110

HARMONIC DISTORTION vs INVERTING GAIN

Gain (V/V)

Harmonic Distortion (dBc)

3rd-Harmonic

See Figure 2

= 2Vp-p

= 200

= 604

f = 1MHz

2nd-Harmonic

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

INPUT VOLTAGE AND CURRENT NOISE DENSITY

Frequency (Hz)

Voltage Noise nV/

Current Noise pA/

2nV/

1.6pA/

Voltage Noise

Current Noise

2-TONE, 3rd-ORDER

INTERMODULATION INTERCEPT

Frequency (MHz)

Intercept Point (+dBm)

1/2

OPA2822

402

CHANNEL-TO-CHANNEL CROSSTALK

Frequency (MHz)

0.1

100

500

Cross-Talk Input Referred (dB)

100

Input Referred

= 100

G = +2

GAIN FLATNESS

Frequency (MHz)

150

100

200

Deviation from 6dB Gain (0.1dB/div)

0.50

0.40

0.30

0.20

0.10

0.00

0.10

0.20

0.30

0.40

0.50

NG = 3.5

= 301

NG = 3.0

= 452

NG = 2.5

= 904

G = 2

Noise Gain

Adjusted

See Figure 12

NG = 2

RECOMMENDED R

vs CAPACITIVE LOAD

Capacitive Load (pF)

100

(

)

1000

100

For Maximally Flat Response,
See Figure 12

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (MHz)

100

500

Normalized Gain to Capacitive Load (dB)

1/2

OPA2822

is optional.

402

= 100pF

= 10pF

= 22pF

= 47pF

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

OUTPUT VOLTAGE AND CURRENT LIMITATIONS

(mA)

200

150

100

150

200

(V)

1W Internal

Power Limit

Single-Channel

1W Internal

Power Limit

Single-Channel

= 25

= 100

= 50

CLOSED-LOOP OUTPUT IMPEDANCE

vs FREQUENCY

Frequency (MHz)

0.1

100

Output Impedance (

)

0.1

0.01

0.001

1/2

OPA2822

402

NONINVERTING OVERDRIVE RECOVERY

Time (40ns/div)

Output Voltage

Input Voltage

= 100

G = +2

See Figure 1

Output

Left Scale

Input Right Scale

INVERTING OVERDRIVE RECOVERY

Time (40ns/div)

Input/Output Voltage

= 100

= 604

See Figure 2

Output

Input

G = 1

CMRR AND PSRR vs FREQUENCY

Frequency (Hz)

Common-Mode Rejection Ratio (dB)

Power-Supply Rejection Ratio (dB)

120

100

CMRR

+PSRR

PSRR

OPEN-LOOP GAIN AND PHASE

Frequency (Hz)

Open-Loop Gain (dB)

120

100

Open-Loop Phase (30

/div)

120

150

180

210

20log(A

)

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

SETTLING TIME

Time (ns)

Percent of Final Value (%)

0.25

0.20

0.15

0.10

0.05

0.10

0.15

0.20

0.25

See Figure 1

= 100

= 2V step

G = +2

TYPICAL DC DRIFT OVER TEMPERATURE

Ambient Temperature (

100

125

Input Offset Voltage (mV)

0.5

Input Bias and Offset Current (

Input Offset Voltage

10x Input Offset Current

Input Bias Current

SUPPLY AND OUTPUT CURRENT

vs TEMPERATURE

Ambient Temperature (

100

125

Output Current (25mA/div)

250

225

200

175

150

125

100

Supply Current (1mA/div)

Sourcing Output Current

Left Scale

Current Limited Output

Sinking Output

Current

Left Scale

Supply Current

(both channels)

Right Scale

COMMON-MODE INPUT RANGE AND

OUTPUT SWING vs SUPPLY VOLTAGE

Supply Voltage (

Voltage Range (V)

Positive Input

and Output

Negative Input

and Output

COMMON-MODE AND DIFFERENTIAL

INPUT IMPEDANCE

Frequency (Hz)

Input Impedance Magnitude 20Log (

) 10

Common-Mode

Differential

VIDEO DIFFERENTIAL GAIN/DIFFERENTIAL PHASE

Video Loads

Differential Gain (%)

Differential Phase (

)

0.30

0.25

0.20

0.15

0.10

0.05

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

= +25

C, Differential Gain

= 2, R

= 604

, R

= 400

, unless otherwise noted.

1/2

OPA2822

604

+6V

1/2

OPA2822

604

D =

604

DIFFERENTIAL SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

= +1

= +2

= +5

= +10

= 200mVp-p

100

105

110

115

DIFFERENTIAL DISTORTION vs OUTPUT VOLTAGE

Differential Output Voltage Swing (Vp-p)

Harmonic Distortion (dBc)

2nd-Harmonic

3rd-Harmonic

f = 1MHz

= 2

= 400

105

DIFFERENTIAL DISTORTION vs FREQUENCY

Frequency (MHz)

Harmonic Distortion (dBc)

2nd-Harmonic

3rd-Harmonic

= 4Vp-p

= 2

= 400

DIFFERENTIAL DISTORTION vs LOAD RESISTANCE

Load Resistance (

)

100

Harmonic Distortion (dBc)

100

105

= 4Vp-p

= 2

f = 1MHz

2nd-Harmonic

3rd-Harmonic

DIFFERENTIAL LARGE-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

= 200mVp-p

= 1Vp-p

= 2Vp-p

= 2

= 400

= 5Vp-p

DIFFERENTIAL PERFORMANCE

TEST CIRCUIT

OPA2822

SBOS188A

www.ti.com

TYPICAL CHARACTERISTICS: V

= +5V

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

NONINVERTING SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

= 0.1Vp-p

G = +1

= 0

G = +2

See Figure 3

G = +5

G = +10

INVERTING SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

0.5

100

500

Normalized Gain (dB)

= 0.1Vp-p

= 604

See Figure 4

G = 5

G = 2

G = 1

G = 10

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (MHz)

500

100

Normalized Gain to Capacitive Load (dB)

= 100pF

= 10pF

= 22pF

= 47pF

1/2

OPA2822

is optional.

402

+5V

804

402

0.01

0.4

0.3

0.2

0.1

0.2

0.3

0.4

NONINVERTING PULSE RESPONSE

Time (20ns/div)

Small-Signal Output Voltage (100mv/div)

2.0

1.5

1.0

0.5

1.0

1.5

2.0

Large-Signal Output Voltage (500mv/div)

See Figure 3

Large-Signal Right Scale

Small-Signal Left Scale

0.4

0.3

0.2

0.1

0.2

0.3

0.4

INVERTING PULSE RESPONSE

Time (20ns/div)

Small-Signal Output Voltage (100mv/div)

2.0

1.5

1.0

0.5

1.0

1.5

2.0

Large-Signal Output Voltage (500mv/div)

See Figure 4

Large-Signal Right Scale

Small-Signal Left Scale

RECOMMENDED R

vs CAPACITIVE LOAD

Capacitive Load (pF)

100

1000

Input Impedance Magnitude 20Log (

) 1000

100

For Maximally Flat Response,
See Figure 12

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TYPICAL CHARACTERISTICS: V

= +5V

(Cont.)

= +25

C, G

= +2, R

= 402

, R

= 100

, unless otherwise noted.

100

105

HARMONIC DISTORTION vs LOAD RESISTANCE

Load Resistance (

)

100

Harmonic Distortion (dBc)

2nd-Harmonic

3rd-Harmonic

See Figure 3

= 2Vp-p

f = 1MHz

100

HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

Harmonic Distortion (dBc)

2nd-Harmonic

3rd-Harmonic

See Figure 3

= 2Vp-p

= 200

HARMONIC DISTORTION vs OUTPUT VOLTAGE

Output Voltage Swing (Vp-p)

0.1

Harmonic Distortion (dBc)

100

105

= 200

f = 1MHz

2nd-Harmonic

3rd-Harmonic

See Figure 3

2-TONE, 3rd-ORDER

INTERMODULATION INTERCEPT

Frequency (MHz)

Intercept Point (+dBm)

1/2

OPA2822

402

+5V

804

402

57.6

0.1

TYPICAL DC DRIFT OVER TEMPERATURE

Ambient Temperature (

100

125

Input Offset Voltage (mV)

0.5

Input Bias and Offset Current (

Input Offset Voltage

10x Input Offset Current

Input Bias Current

SUPPLY AND OUTPUT CURRENT

vs TEMPERATURE

Ambient Temperature (

100

125

Output Current (25mA/div)

200

175

150

125

100

Supply Current (1mA/div)

Sourcing Output Current

Left Scale

Sinking Output Current

Left Scale

Supply Current

(both channels)

Right Scale

Current Limited Output

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APPLICATIONS INFORMATION

WIDEBAND NONINVERTING OPERATION

The OPA2822 provides a unique combination of features in
a wideband dual, unity gain stable voltage feedback amplifier
to support the extremely high dynamic range requirements of
emerging communications technologies. Combining low
2nV/

Hz input voltage noise with harmonic distortion perfor-

mance that can exceed 100dBc SFDR through 2MHz, the
OPA2822 provides the highest dynamic range input interface
for emerging high speed 14-bit (and higher) converters. To
achieve this level of performance, careful attention to circuit
design and board layout is required.

Figure 1 shows the gain of +2 configuration used as the basis
for the Electrical Characteristics table and most of the Typical
Characteristics at

6V operation. While the characteristics are

given using split

6V supplies, most of the electrical and typical

characteristics also apply to a single-supply +12V design where
the input and output operating voltages are centered at the
midpoint of the +12V supply. Operation at

5V will very nearly

match that shown for the

6V operating point. Most of the

curves were characterized using signal sources with 50

driv-

ing impedance, and with measurement equipment presenting a
50

load impedance. In Figure 1, the 50

shunt resistor at the

terminal matches the source impedance of the test signal

generator, while the 50

series resistor at the V

terminal

provides a matching resistor for the measurement equipment
load. Generally, data sheet voltage swing specifications are at
the output pin (V

in Figure 1), while output power (dBm)

specifications are at the matched 50

load. The total 100

load

at the output, combined with the total 804

total feedback

network load for the noninverting configuration of Figure 1,
presents the OPA2822 with an effective output load of 89

While this is a good load value for frequency response mea-
surements, distortion will improve rapidly with lighter output
loads. Keeping the same feedback network and increasing the
load to 200

will give a total load of 160

for the distortion

performance reported in the Electrical Characteristics table.

For higher gains, the feedback resistor (R

) was held at 402

and the gain resistor (R

) adjusted to develop the Typical

Characteristics.

Voltage feedback op amps, unlike current feedback designs,
can use a wide range of resistor values to set their gains. A low-
noise part like the OPA2822 will deliver low total output noise
only if the resistor values are kept relatively low. For the circuit
of Figure 1, the resistors contribute an input-referred voltage
noise component of 1.8nV/

Hz, which is approaching the value

of the amplifier's intrinsic 2nV/

Hz. For a more complete de-

scription of the feedback network's impact on noise, see the
section titled "Setting Resistor Values to Minimize Noise" later
in this data sheet. In general, the parallel combination of R

and

should be < 300

to retain the low-noise performance of the

OPA2822. However, setting these values too low can impair
distortion performance due to output loading, as shown in the
distortion versus load data in the Typical Characteristics.

WIDEBAND INVERTING OPERATION

Operating the OPA2822 as an inverting amplifier has several
benefits and is particularly appropriate as part of the hybrid
design in an xDSL receiver application. Figure 2 shows the
inverting gain of 1 circuit used as the basis of the inverting
mode Typical Characteristics.

1/2

OPA2822

+5V

0.1

6.8

402

Source

Load

0.1

FIGURE 1. Noninverting G = +2 Specification and Test

Circuit.

In the inverting case, only the R

element of the feedback

network appears as part of the total output load in parallel
with the actual load. For the 100

load used in the Typical

Characteristics, this gives an effective load of 86

in this

inverting configuration. Gain resistor R

is set to achieve the

desired inverting gain (in this case 604

for a gain of 1)

while an additional input matching resistor (R

) can be used

to set the total input impedance equal to the source if
desired. In this case, R

= 54.9

in parallel with the 604

gain setting resistor yields a matched input impedance of
50

. R

is needed only when the input must be matched to

a source impedance, as in the characterization testing done
using the circuit of Figure 2.

1/2

OPA2822

+5V

0.1

6.8

54.9

309

604

Source

Load

0.1

604

FIGURE 2. Inverting G = 1 Specifications and Test Circuit.

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To take full advantage of the OPA2822's excellent DC input
accuracy, the total DC impedance seen at of each of the
input terminals must be matched to get bias current cancel-
lation. For the circuit of Figure 2, this requires the grounded
309

resistor on the noninverting input. The calculation for

this resistor value assumes a DC-coupled 50

source

impedance along with R

and R

. While this resistor will

provide cancellation for the input bias current, it must be
well decoupled (0.1

F in Figure 2) to filter the noise contri-

bution of the resistor itself and of the amplifier's input
current noise.

As the required R

resistor approaches 50

at higher gains,

the bandwidth for the circuit in Figure 2 will far exceed the
bandwidth at the same gain magnitude for the noninverting
circuit of Figure 1. This occurs due to the lower "noise gain"
for the circuit of Figure 2 when the 50

source impedance is

included in the analysis. For example, at a signal gain of
12 (R

= 50

, R

= open, R

= 604

) the noise gain for the

circuit of Figure 2 will be 1 + 604

/(50

+ 50

) = 7, due to

the addition of the 50

source in the noise gain equation.

This will give considerably higher bandwidth than the nonin-
verting gain of +12.

SINGLE-SUPPLY NONINVERTING OPERATION

The OPA2822 can also support single +5V operation with
its exceptional input and output voltage swing capability.
While not a rail-to-rail input/output design, both inputs and
outputs can swing to within 1.2V of either supply rail. For a
single amplifier channel, this gives a very clean 2Vp-p
output capability on a single +5V supply, or 4Vp-p output for
a differential configuration using both channels together.
Figure 3 shows the AC-coupled noninverting gain of +2
used as the basis of the Electrical Characteristics table and
most of the Typical Characteristics for single +5V supply
operation.

The key requirement of broadband single-supply operation is
to maintain input and output signal swings within the usable
voltage range at both input and output. The circuit of Figure
3 establishes an input midpoint bias using a simple resistive
divider from the +5V supply (two 804

resistors). These two

resistors are selected to provide DC bias current cancellation
because their parallel combination matches the DC imped-
ance looking out of the inverting node, which equals R

. The

gain setting resistor is not part of the DC impedance looking
out of the inverting node, due to the blocking capacitor in
series with it. The input signal is then AC-coupled into the
midpoint voltage bias. The input impedance matching resis-
tor (57.6

) is selected for testing to give a 50

input match

(at high frequencies) when the parallel combination of the
biasing divider network is included. The gain resistor (R

) is

AC-coupled giving a DC gain of +1. This centers the output
also at the input midpoint bias voltage (V

/2). While this

circuit is shown using a +5V supply, this same circuit may be
applied for single-supply operation as high as +12V.

SINGLE-SUPPLY INVERTING OPERATION

For those single +5V Typical Characteristics that require
inverting gain of 1 operation, the test circuit in Figure 4 was
used.

FIGURE 3. AC-Coupled, G = +2, Single-Supply Operation:

Specification and Test Circuit.

1/2

OPA2822

+5V

100

804

57.6

402

0.1

6.8

0.1

As with the circuit of Figure 2, the feedback resistor (R

) has

been increased to 604

to reduce the loading effect it has

in parallel with the 100

actual load. The noninverting input

is biased at V

/2 (2.5V in this case) using the two 1.21k

resistors for R

. The parallel combination of these two

resistors (605

) provides input bias current cancellation by

matching the DC impedance looking out of the inverting
input node. The noninverting input bias is also well de-
coupled using the 0.1

F capacitor to both reduce both

power supply noise and the resistor and bias current noise
at this input.

FIGURE 4. AC-Coupled, G = 1, Single-Supply Operation:

Specification and Test Circuit.

1/2

OPA2822

+5V

100

1.21k

0.1

6.8

54.9

Source

604

0.1

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The gain resistor (R

) is set to equal the feedback resistor (R

)

at 604

to achieve the desired gain of 1 from V

to V

. A DC

blocking capacitor is included in series with the R

to reduce the

DC gain for the noninverting input bias and offset voltages to +1.
This places the V

/2 bias voltage at the output pin and reduces

the output DC offset error terms. The signal input impedance is
matched to the 50

source using the additional R

resistor set

to 54.9

. At higher frequencies, the parallel combination of R

and R

provides the input impedance match at 50

. This is

principally used for test and characterization purposes--system
applications do not necessarily require this input impedance
match, particularly if the source device is physically near the
OPA2822 and/or does not require a 50

input impedance

match. At higher gains, the signal source impedance will start to
materially impact the apparent noise gain and hence bandwidth
of the OPA2822.

ADSL RECEIVE AMPLIFIER

One of the principal applications for the OPA2822 is a low-power,
low-noise receive amplifier in ADSL modem designs. Applica-
tions ranging from single +5V,

5V, and up to single +12V

supplies can be well supported by the OPA2822. For higher
supplies, consider the dual, low-noise THS6062 ADSL receive
amplifier that can support up to

15V supplies. Figure 5 shows a

typical ADSL receiver design where the OPA2822 is used as an
inverting summing amplifier to provide both driver output signal
cancellation and receive channel gain. In the circuit of Figure 5,
the driver differential output voltage is shown as V

, while the

receiver channel output is shown as V

The two sets of resistors, R

and R

, are set to provide the

desired gain from the transformer windings for the signal
arriving on the line side of the transformer, and also to provide
nominal cancellation for the driver output signal (V

) to the

receiver output. Typically, the two R

resistors are set to

provide impedance matching through the transformer. This is
accomplished by setting R

= 0.5 · (R

), where N is the

turns ratio used for the line driver design. If R

is set in this

fashion, and the actual twisted pair line shows the expected R

impedance value, the voltage swing produced at V

will be cut

in half at the transformer input. In this case, setting R

= 2 · R

will achieve cancellation of the driver output signal at the
output of the receiver. Essentially, the driver output voltage
produces a current in R

that is exactly matched by the current

pulled out of R

due to the attenuated and inverted version of

the output signal at the transformer input. In actual practice, R

and R

are usually RC networks to achieve cancellation over

the frequency varying line impedance.

As the transformer turns ratio changes to support different
line driver and supply voltage combinations, the impact of
receiver amplifier noise will change. Typically, DSL systems
incur a line referred noise contribution for the receiver that
can be computed for the circuit of Figure 5. For example,
targeting an overall gain of 1 from the line to the receiver
output, and picking the input resistor R

, the remaining

resistors will be set by the driver cancellation and gain
requirements. With the resistor values set, a line referred
noise contribution due to the OPA2822 can be computed. R

will be set to 2x the value of R

, and the feedback resistor will

be set to recover the gain loss through the transformer. Table
I shows the total line referred noise floor (in dBm/Hz) using
three different values for R

over a range of transformer turns

ratio (where the amplifier gain is adjusted at each turns ratio).

FIGURE 5. Example ADSL Receiver Amplifier.

1:n

Driver

1/2

OPA2822

+5V

1/2

OPA2822

Line

= 200

= 500

= 1000

151.5

150.2

148.5

1.5

149.1

147.6

145.8

147.2

145.6

143.7

2.5

145.6

144.0

142.1

144.3

142.7

140.7

3.5

143.2

141.5

139.5

142.2

140.5

138.4

4.5

141.3

139.5

137.5

140.4

138.7

136.6

TABLE I. Line Referred Noise dBm/Hz, Due to Receiver Op Amp.

Table I shows that a lower transformer turns ratio results in
reduced line referred noise, and that the resistor noise will
start to degrade the noise at higher values--particularly in
going from 500

to 1k

. In general, line referred noise floor

due to the receiver channel will not be the limit to ADSL
modem performance, if it is lower than 145dBm.

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FIGURE 7. Frequency Response for the Filter in Figure 6.

FIGURE 6. 6th-Order Bandpass Filter.

1/2

OPA2822

1/2

OPA2822

+5V

225

158

150pF

12pF

18pF

180pF

66pF

150

107

2.1k

140

1.0nF

2.2pF

143

300

1.8nF

1.3k

FIGURE 9. Frequency Response for the Filter of Figure 8.

Frequency (Hz)

1.0E+04

1.0E+05

1.0E+06

1.0E+08

1.0E+07

Gain (dB)

Frequency (Hz)

1.0E+04

1.0E+05

1.0E+06

1.0E+07

Gain (dB)

ACTIVE FILTER APPLICATIONS

As a low-noise, low-distortion unity gain stable voltage feed-
back amplifier, the OPA2822 provides an ideal building block
for high-performance active filters. With two channels avail-
able, it can be used either as a cascaded two-stage active
filter or as a differential filter. Figure 6 shows a 6th-order
bandpass filter cascaded with two 2nd-order Sallen-Key
sections, with transmission zeroes along with a passive post
filter made up of a high-pass and a low-pass section. The first
amplifier provides a 2nd-order high-pass stage while the
second amplifier provides the 2nd-order low-pass stage.
Figure 7 shows the frequency response for this example
filter.

A differential active filter is shown in Figure 8. This circuit
shows a single-supply, 2nd-order high-pass filter with the
corner frequencies set to provide the required high-pass
function for an ADSL CPE modem application. To use this
circuit, the hybrid would be implemented as a passive sum-
ming circuit at the input to this filter. For +5V only ADSL
designs, it is preferable to implement a portion of the filtering
prior to the amplifier, thus limiting the amplitude of the
uncancelled line driver signals. This type of receiver stage
would typically then drive a low-pass filter prior to the codec
setting the high frequency cutoff of the ADC (Analog-to-
Digital Converter) input signal. Figure 9 shows the frequency
response for the high-pass circuit of Figure 8.

1/2

OPA2822

+5V

1/2

OPA2822

2.2

365

730

FIGURE 8. Single-Supply, 2nd-Order High-Pass Active

Filter.

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HIGH DYNAMIC RANGE ADC DRIVER

Numerous circuit approaches exist to provide the last stage
of amplification before the ADC in high-performance applica-
tions. For very high dynamic range applications where the
signal channel can be AC-coupled, the circuit shown in
Figure 10 provides exceptional performance. Most very high
performance ADCs > 12-bit performance require differential
inputs to achieve the dynamic range. The circuit of Figure 10
converts a single-ended source to differential via a 1:2 turns
ratio transformer, which then drives the inverting gain setting
resistors (R

). These resistors are fixed at 100

to provide

input matching to a 50

source on the transformer primary

side. The gain can then be adjusted by setting the feedback
resistor values. For best performance, this circuit operates
with a ground centered output on

5V supplies, although a

+12V supply can also provide excellent results. Since most
high-performance converters operate on a single +5V sup-
ply, the output is level shifted through an AC blocking
capacitor to the common-mode input voltage (V

) for the

converter input, and then low-pass filtered prior to the input
of the converter. This circuit is intended for inputs from 10kHz
to 10MHz, so the output high-pass corner is set to 1.6kHz,
while the low-pass cutoff is set to 20MHz. These are example
cutoff frequencies; the actual filtering requirements would be
set by the specific application.

The 1:2 turns ratio transformer also provides an improvement
in input referred noise figure. Equation 1 shows the Noise
Figure (NF) calculation for this circuit, where R

has been

constrained to provide an input match to R

(through the

transformer) and then R

is set to get the desired overall

gain. With these constraints (and 0

on the noninverting

inputs), the noise figure equation simplifies considerably.

i nR

KTR

(

)

log

(1)

where R

= 1/2 n

n = Transformer Turns Ratio

= R

= Op Amp Input Voltage Noise

= Inverting Input Current Noise

KT = 4E 21 [T = 290

Gain (dB) = 20log[n

]

FIGURE 10. Single-Ended to Differential High Dynamic Range ADC Driver.

1/2

OPA2822

+5V

1:2

+5V

Noise

Figure

Defined

Here

1/2

OPA2822

100

14-Bit

ADC

0.1

100pF

0.1

500

= 50

= 2

REQUIRED

TOTAL GAIN

LOG GAIN

AMPLIFIER GAIN

NOISE FIGURE

(V/V)

(dB)

(

)

(dB)

12.0

11.2

14.0

2.5

10.4

15.6

9.9

16.9

3.5

9.5

18.1

9.1

19.1

4.5

8.9

20.0

8.6

TABLE II. Noise Figure versus Gain with n = 2 Transformer.

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FREQUENCY RESPONSE CONTROL

Voltage-feedback op amps like the OPA2822 exhibit de-
creasing closed-loop bandwidth as the signal gain is in-
creased. In theory, this relationship is described by the Gain
Bandwidth Product (GBP) shown in the electrical character-
istics. Ideally, dividing GBP by the noninverting signal gain
(also called the Noise Gain, NG) will predict the closed-loop
bandwidth. In practice, this holds true only when the phase
margin approaches 90

, as it does in higher gain configura-

tions. At low gains, most high-speed amplifiers will show a
more complex response with lower phase margin and higher
bandwidth than predicted by the GBP. The OPA2822 is
compensated to give a slightly peaked frequency response
at a gain of +2 (see the circuit in Figure 1). The 200MHz
typical bandwidth at a gain of +2 far exceeds that predicted
by dividing the GBP of 240MHz by the gain of 2. The
bandwidth predicted by the GBP is more closely correct as
the gain increases. As shown in the Typical Characteristics,
at a gain of +10, the 3dB bandwidth of 24MHz matches that
predicted by dividing the GBP by 10.

Inverting operation offers some interesting opportunities to
increase the available signal bandwidth. When the source
impedance is matched by the gain resistor (Figure 10 for
example), the signal gain is (1 + R

) while the noise gain

is (1 + R

/2R

). This reduces the noise gain almost by half,

extending the signal bandwidth and increasing the loop gain.
For instance, setting R

= 500

in Figure 10 will give a signal

gain for the amplifier of 5V/V. However, including the 50

source impedance reflected through the 1:2 transformer will
give an additional 100

source impedance for the noise gain

analysis for each of the amplifiers. This reduces the noise gain
to 1 + 500

/200

= 3.5V/V and results in an amplifier

bandwidth of at least 240MHz/3.5 = 68MHz.

DRIVING CAPACITIVE LOADS

One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC, including additional
external capacitance which may be recommended to im-
prove ADC linearity. A high-speed, high open-loop gain
amplifier like the OPA2822 can be very susceptible to de-
creased stability and closed-loop frequency response peak-
ing when a capacitive load is placed directly on the output
pin. When the amplifier's open-loop output resistance is
considered, this capacitive load introduces an additional pole
in the signal path that can decrease the phase margin.
Several external solutions to this problem have been sug-
gested. When the primary considerations are frequency
response flatness with low noise and distortion, the simplest
and most effect solution is to isolate the capacitive load from
the feedback loop by inserting a series isolation resistor
between the amplifier output and the capacitive load. This
does not eliminate the pole from the loop response, but
instead shifts it and adds a zero at a higher frequency. The
additional zero acts to cancel the phase lag from the capaci-
tive load pole, thus increasing the phase margin and improv-
ing stability.

OPERATING SUGGESTIONS

SETTING RESISTOR VALUES TO MINIMIZE NOISE

Getting the full advantage of the OPA2822's low input noise
requires careful attention to the external gain setting and DC
biasing networks. The feedback resistor is part of the overall
output load (which can start to degrade distortion if set too
low). With this in mind, a good starting point for design is to
select the feedback resistor as low as possible (consistent
with loading distortion concerns), then continue with the
design, and set the other resistors as needed. To retain full
performance, setting the feedback resistor in the range of
200

to 750

can provide a good start to the design.

Figure 11 shows the full output noise analysis model for any
op amp.

4kT

1/2

OPA2822

4kT = 1.6E 20J

at 290

4kTR

FIGURE 11. Op Amp Noise Analysis Model.

The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
terms. Equation 2 shows the general form of this output noise
voltage expression using the terms shown in Figure 11.

kTR

I R

kTR NG

BN S

BI F

(

)

(

)

(2)

Dividing this expression by the noise gain (NG = 1 = R

)

will give the total equivalent spot noise voltage referred to the
noninverting input, as shown in Equation 3:

kTR

I R

kTR

BN S

BI F

(

)

(3)

Inserting high resistor values into Equation 3 can quickly
dominate the total equivalent input referred voltage noise. A
250

source impedance on the noninverting input will add as

much noise as the amplifier itself. If the noninverting input is
a DC bias path (as in inverting or in some single-supply
applications), it is critical to include a noise shunting capaci-
tor with that resistor to limit the added noise impact of those
resistors (see the example in Figure 2).

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The Typical Characteristics show the recommended R

ver-

sus capacitive load and the resulting frequency response at
the load. For the OPA2822 operating at a gain of +2, the
frequency response at the output pin is already slightly
peaked without the capacitive load, requiring relatively high
values of R

to flatten the response at the load. One way to

reduce the required R

value is to use the noise gain

adjustment circuit of Figure 12.

1/2

OPA2822

402

Source

FIGURE 12. Noise Gain Tuning for Noninverting Circuit.

100

Capacitive Load (pF)

100

1000

(

)

NG = 3, R

= 452

NG = 2, R

FIGURE 13. Required R

vs Noise Gain.

DISTORTION PERFORMANCE

The OPA2822 is capable of delivering exceptionally low
distortion through approximately 5MHz signal frequency.
While principally intended to provide very low noise and
distortion through the maximum ADSL frequency of 1.1MHz,
the OPA2822 in a differential configuration can deliver lower
than 85dBc distortions for a 4Vp-p swing through 5MHz. For
applications requiring extremely low distortion through higher
frequencies, consider higher slew rate amplifiers such as the
OPA687 or OPA2681.

As the Typical Characteristics show, until the fundamental
signal reaches very high frequencies or power levels, the limit
to SFDR will be second harmonic distortion rather than the
negligible third harmonic component. Focusing then on the
second harmonic, increasing the load impedance improves
distortion directly. However, operating differentially offers the
most significant improvement in even order distortion terms.
For example, the Electrical Characteristics show that a single
channel of the OPA2822, delivering 2Vp-p at 1MHz into a
200

load, will typically show a second harmonic product at

92dBc versus the third harmonic at 102dBc. Changing the
configuration to a differential driver where each output still
drives 2Vp-p results in a 4Vp-p total differential output into a
400

differential load, giving the same single ended load of

200

for each amplifier. This configuration drops the second

harmonic to 103dBc and the third to approximately 105dBc--
an overall dynamic range improvement of more than 10dB.

For general distortion analysis, remember that the total
loading on the amplifier includes the feedback network; in the
noninverting configuration, this is the sum of R

+ R

, while

in the inverting configuration this additional loading is simply
R

. Increasing the output voltage swing increases the har-

monic distortion directly. A 6dB increase in the output swing
will generally increase the second harmonic 12dB and the

The resistor across the two inputs, R

, can be used to

increase the noise gain while retaining the desired signal
gain. This can be used either to improve flatness at low gains
or to reduce the required value of R

in capacitive load

driving applications. This circuit was used with R

adjusted

to produce the gain flatness curve in the Typical Character-
istics. As shown in that curve, an R

of 452

will give an NG

of 3 giving exceptional frequency response flatness at a
signal gain of +2. Equation 4 shows the calculation for R

given a target noise gain (NG) and signal gain (G).

R G

NG G

(4)

where R

= Total Source Impedance on Noninverting Input

[25

in Figure 12]

G = Signal Gain [1 + (R

)]

NG = Noise Gain Target

Using this technique to get initial frequency response flat-
ness will significantly reduce the required series resistor
value to get a flat response at the capacitive load. Using the
best case noise gain of 3 with a signal gain of 2 allows the
required R

to be reduced, as shown in Figure 13. Here, the

required R

versus Capacitive Load is replotted along with

data from the Typical Characteristics. This demonstrates that
the use of R

= 452

across the inputs results in much

lower required R

values to achieve a flat response.

OPA2822

SBOS188A

www.ti.com

third harmonic 18dB. Increasing the signal gain will also
generally increase both the second and third harmonics
because the loop gain decreases at higher gains. Again, a
6dB increase in voltage gain will increase the second har-
monic distortion by approximately 6dB. The distortion char-
acteristic curves for the OPA2822 show little change in the
third harmonic distortion versus gain. Finally, the overall
distortion generally increases as the fundamental frequency
increases due to the rolloff in the loop gain with frequency.
Conversely, the distortion will improve going to lower fre-
quencies, down to the dominant open-loop pole at approxi-
mately 50kHz. This will give essentially unmeasurable levels
of harmonic distortion in the audio band.

The OPA2822 exhibits an extremely low 3rd-order harmonic
distortion. This also gives exceptionally good 2-tone 3rd-
order intermodulation intercept as shown in the Typical
Characteristics. This intercept curve is defined at the 50

load when driven through a 50

matching resistor to allow

direct comparisons to RF MMIC devices. This network at-
tenuates the voltage swing from the output pin to the load by
6dB. If the OPA2822 drives directly into the input of a high-
impedance device, such as an ADC, this 6dB attenuation
does not occur. Under these conditions, the intercept will
improve by at least 6dBm. The intercept is used to predict the
intermodulation spurs for two closely spaced frequencies. If
the two test frequencies, f

and f

, are specified in terms of

average and delta frequency, f

= (f

+ f

)/2 and

= |f

the two, 3rd-order, close-in spurious tones will appear at f

3 ·

. The difference between two equal test-tone power

levels and the spurious intermodulation power levels is given
by

dBc = 2 · (IM3 P

), where IM3 is the intercept taken

from the Typical Specification and P

is the power level in

dBm at the 50

load for either one of the two closely spaced

test frequencies. For example, at 1MHz in a gain of +2
configuration, the OPA2822 exhibits an intercept of 57dBm
at a matched 50

load. If the full envelope of the two

frequencies needs to be 2Vp-p, each tone will be set to
4dBm. The 3rd-order intermodulation spurious tones will
then be 2 · (57 4) = 106dBc below the test-tone power level
(102dBm). If this same 2Vp-p 2-tone envelope were deliv-
ered directly into the input of an ADC without the matching
loss or loading of the 50

network, the intercept would

increase to at least 63dBm. With the same signal and gain
conditions but now driving directly into a light load, the
spurious tones would then be at least 2 · (63 4) = 118dBc
below the test-tone power levels.

DC ACCURACY AND OFFSET CONTROL

The OPA2822 can provide excellent DC signal accuracy due
to its high open-loop gain, high common-mode rejection, high
power-supply rejection, and low input offset voltage and bias
current offset errors. To take full advantage of the low input
offset voltage (

1.2mV maximum at 25

C), careful attention

to input bias current cancellation is also required. The high-
speed input stage for the OPA2822 has relatively high input
bias current (8

A typical into the pins) but with a very close

match between the two input currents, typically 100nA input

offset current. The total output offset voltage may be reduced
considerably by matching the source impedances looking out
of the two inputs. For example, one way to add bias current
cancellation to the circuit of Figure 1 would be to insert a
175

series resistor into the noninverting input from the 50

terminating resistor. If the 50

source resistor is DC coupled,

this will increase the source impedance for the noninverting
input bias current to 200

. Since this is now equal to the

impedance looking out of the inverting input (R

|| R

), the

circuit will cancel the bias current effects, leaving only the
offset current times the feedback resistor as a residual DC
error term at the output. Using a 402

feedback resistor, the

output DC error due to the input bias currents will now be less
than 0.7

A · 402

= 0.28mV over the full temperature range.

This is significantly lower than the contribution due the input
offset voltage. At a gain of +2, the maximum input offset
voltage is 1.5mV, giving a total maximum output offset of
(

3mV

0.28mV) =

3.3mV over the 40

C to +85

temperature range (for the circuit of Figure 1, including the
additional 175

resistor at the noninverting input).

THERMAL ANALYSIS

The OPA2822 will not require heatsinking or airflow under
most operating conditions. Maximum desired junction tem-
perature will limit the maximum allowed internal power dissi-
pation as described below. In no case should the maximum
junction temperature be allowed to exceed +175

Operating junction temperature (T

) is given by T

+ P

The total internal power dissipation (P

) is the sum of the

quiescent power (P

) and additional power dissipated in the

output stage (P

) to deliver load power. Quiescent power is

simply the specified no-load supply current times the total
supply voltage across the part. P

will depend on the required

output signal and load but would, for a grounded resistive load,
be at a maximum when the output is fixed at a voltage equal
to half of either supply voltage (assuming equal bipolar sup-
plies). Under this condition P

= V

/(4 · R

) where R

includes feedback network loading.

Note that it is the power dissipated in the output stage and not in
the load that determines internal power dissipation. As a worst
case example, compute the maximum T

for the OPA2822E with

both channels operating at A

= +2, R

= 100

, R

= 400

5V, and at the specified maximum T

= 85

= 10V · 11.4mA + 2 · (5

)/(4 · (100 || 804)) = 255mW.

Maximum T

= 85

C + 0.255W · 150

C/W = 123

C. This

calculation represents a worst case combination of condi-
tions to reach a maximum possible operating junction tem-
perature. Under most operating conditions, the junction tem-
perature will be far lower than the 123

C calculated here.

The output current is limited in the OPA2822 to protect
against damage under short-circuit conditions. This current
limited output of approximately 220mA exceeds the rated
typical output current of 150mA. The typical and minimum
output current limits are set for linear operation while the
maximum output shown in the Typical Characteristics is
nonlinear limited performance.

OPA2822

SBOS188A

www.ti.com

BOARD LAYOUT

Achieving optimum performance with a high-frequency am-
plifier like the OPA2822 requires careful attention to board
layout parasitics and external component types. Recommen-
dations that will optimize performance include:

a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output and
inverting input pins can cause instability: on the noninverting
input, it can react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted capacitance,
a window around the signal I/O pins should be opened in all
of the ground and power planes around those pins. Other-
wise, ground and power planes should be unbroken else-
where on the board.

b) Minimize the distance (< 0.25") from the power-supply
pins to high frequency 0.1

F decoupling capacitors. At the

device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the device pins and the decoupling capacitors. The primary
power-supply connections (on pins 4 and 8) should always
be decoupled with these capacitors. Larger (2.2

F to 6.8

decoupling capacitors, effective at lower frequency, should
also be used on the main supply pins. These may be placed
somewhat farther from the device and may be shared among
several devices in the same area of the PC board.

c) Careful selection and placement of external components
will preserve the high-frequency performance of the OPA2822.
Resistors should be a very low reactance type. Surface-mount
resistors work best and allow a tighter overall layout. Metal film
and carbon composition axially leaded resistors can also provide
good high-frequency performance. Again, keep their leads and
PC board trace length as short as possible. Never use wirewound
type resistors in a high-frequency application. Since the output pin
and inverting input pin are the most sensitive to parasitic capaci-
tance, always position the feedback and series output resistor, if
any, as close as possible to the output pin. Other network
components, such as noninverting input termination resistors,
should also be placed close to the package. Even with a low
parasitic capacitance shunting the external resistors, excessively
high resistor values can create significant time constants that can
degrade performance. Good axial metal film or surface mount
resistors have approximately 0.2pF in shunt with the resistor. For
resistor values > 1.5k

, this parasitic capacitance can add a pole

and/or zero below 500MHz that can effect circuit operation. Keep
resistor values as low as possible consistent with parasitic load,
distortion, and noise considerations. The 402

feedback used in

the Typical Characteristics is a good starting point for design.

d) Connections to other wideband devices on the board may
be made with short direct traces or through onboard transmission
lines. For short connections, consider the trace and the input to
the next device as a lumped capacitive load. Relatively wide
traces (50mils to 100mils) should be used, preferably with ground
and power planes opened up around them. Estimate the total
capacitive load and set R

from the plot of recommended R

versus capacitive load. If a long trace is required, and the 6dB
signal loss intrinsic to a doubly terminated transmission line is
acceptable, implement a matched impedance transmission line

using microstrip or stripline techniques (consult an ECL design
handbook for microstrip and stripline layout techniques). A 50

environment is normally not necessary onboard, and in fact a
higher impedance environment will improve distortion as shown
in the distortion versus load plots. With a characteristic board
trace impedance defined based on board material and trace
dimensions, a matching series resistor into the trace from the
output of the OPA2822 is used as well as a terminating shunt
resistor at the input of the destination device. Remember also that
the terminating impedance will be the parallel combination of the
shunt resistor and the input impedance of the destination device;
this total effective impedance should be set to match the trace
impedance. Multiple destination devices are best handled as
separate transmission lines, each with their own series and shunt
terminations. If the 6dB attenuation of a doubly-terminated trans-
mission line is unacceptable, a long trace can be series-termi-
nated at the source end only. Treat the trace as a capacitive load
in this case and set the series resistor value as shown in the plot
of "R

vs Capacitive Load". This will not preserve signal integrity

as well as a doubly-terminated line. If the input impedance of the
destination device is low, there will be some signal attenuation
due to the voltage divider formed by the series output into the
terminating impedance.

e) Socketing a high-speed part like the OPA2822 is not
recommended. The additional lead length and pin-to-pin ca-
pacitance introduced by the socket can create an extremely
troublesome parasitic network which can make it almost impos-
sible to achieve a smooth, stable frequency response. Best
results are obtained by soldering the OPA2822 onto the board.

INPUT AND ESD PROTECTION

The OPA2822 is built using a very high-speed complementary
bipolar process. The internal junction breakdown voltages are
relatively low due to these very small geometry devices. These
breakdowns are reflected in the Absolute Maximum Rating
table. All device pins are protected with internal ESD protection
diodes to the power supplies, as shown in Figure 14.

External

Internal
Circuitry

FIGURE 14. Internal ESD Protection.

These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes can
typically support 30mA continuous current. Where higher cur-
rents are possible (e.g. in systems with

15V supply parts

driving into the OPA2822), current limiting series resistors
should be added into the two inputs. Keep these resistor values
as low as possible since high values degrade both noise
performance and frequency response.

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