OPA686

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OPA686

Wideband, Low Noise,

Voltage Feedback OPERATIONAL AMPLIFIER

APPLICATIONS

HIGH DYNAMIC RANGE ADC PREAMP

LOW NOISE, WIDEBAND,
TRANSIMPEDANCE AMPLIFIER

WIDEBAND, HIGH GAIN AMPLIFIER

LOW NOISE DIFFERENTIAL RECEIVER

VDSL LINE RECEIVER

ULTRASOUND CHANNEL AMPLIFIER

IMPROVED REPLACEMENT FOR THE
CLC425

FEATURES

HIGH BANDWIDTH: 250MHz (G = +10)

LOW INPUT VOLTAGE NOISE: 1.3nV/

VERY LOW DISTORTION: 90dBc (5MHz)

HIGH SLEW RATE: 600V/

HIGH DC ACCURACY

LOW SUPPLY CURRENT: 12mA

HIGH GAIN BANDWIDTH PRODUCT:
1600MHz

STABLE FOR GAINS

DESCRIPTION

The OPA686 combines very high gain bandwidth and large
signal performance with very low input voltage noise while
dissipating a low 12mA supply current. The classical differen-
tial input stage, along with two stages of forward gain and a
high power output stage, combine to make the OPA686 an
exceptionally low distortion amplifier with excellent DC accu-
racy and output drive. The voltage feedback architecture allows
all standard op amp applications to be implemented with very
high performance.

The combination of low input voltage and current noise, along
with a 1.6GHz gain bandwidth product, make the OPA686 an
ideal amplifier for wideband transimpedance stages. As a volt-
age gain stage, the OPA686 is optimized for a flat response at a
gain of +10 and is guaranteed stable down to a noise gain of +7.

High Gain, 20MHz Transimpedance Amplifier

A new external compensation technique can be used to give a
very flat frequency response below the minimum stable gain
for the OPA686, further improving its already exceptional
distortion performance. Using this compensation makes the
OPA686 one of the premier 12- to 14-bit analog-to-digital
converter input drivers. The supply current for the OPA686 is
precisely trimmed to 12.4mA at +25

C. This, along with

carefully defined supply current tempcos in the input and
output stages, combine to provide exceptional performance
over the full specified temperature range.

OPA686

1997 Burr-Brown Corporation

PDS-1370D

Printed in U.S.A. May, 2000

OPA686

+5V

50k

0.1

100pF

50k

0.2pF

10pF

Photodiode

Supply decoupling

not shown.

100

Frequency (MHz)

0.1

100

20·log (Z

) 5dB/div

20log (50k

) = 94dB

OPA686 RELATED PRODUCTS

INPUT NOISE

GAIN BANDWIDTH

SINGLES

DUALS

VOLTAGE (nV/

Hz)

PRODUCT (MHz)

OPA643

2.3

800

OPA2686

1.3

1600

OPA687

0.95

3600

For most current data sheet and other product

information, visit www.burr-brown.com

SBOS063

OPA686

OPA686U, N

TYP

GUARANTEED

C to

40

C to

MIN/

TEST

PARAMETER

CONDITIONS

+25

(2)

(3)

+85

(3)

UNITS

MAX LEVEL

(1)

SPECIFICATIONS: V

= 453

, R

= 100

and G =+10, unless otherwise noted. Figure 1 for AC performance.

AC PERFORMANCE (Figure 1)
Closed-Loop Bandwidth

G = +7, R

= 50

, V

= 200mVp-p

425

MHz

typ

G = +10, R

= 50

, V

= 200mVp-p

250

200

170

140

MHz

min

G = +20, R

= 50

, V

= 200mVp-p

100

MHz

min

Gain Bandwidth Product

+40

1600

1250

1100

1000

MHz

min

Bandwidth for 0.1dB Gain Flatness

G = +10, R

= 100

, V

= 200mVp-p

MHz

min

Peaking at a Gain of +7

typ

Harmonic Distortion

G = +10, f = 5MHz, V

= 2Vp-p

2nd Harmonic

= 100

dBc

max

= 500

dBc

max

3rd Harmonic

= 100

dBc

max

= 500

110

105

100

dBc

max

Two-Tone, 3rd-Order Intercept

G = +10, f = 10MHz

dBm

min

Input Voltage Noise

f > 1MHz

1.3

1.5

1.6

1.7

nV/

max

Input Current Noise

f > 1MHz

1.8

2.3

2.4

2.5

pA/

max

Rise/Fall Time

0.2V Step

1.4

1.75

2.5

max

Slew Rate

2V Step

600

500

400

310

min

Settling Time to 0.01%

2V Step

typ

0.1%

2V Step

max

2V Step

max

Differential Gain

G = +10, NTSC, R

= 150

0.02

typ

Differential Phase

G = +10, NTSC, R

= 150

0.02

deg

typ

DC PERFORMANCE

(4)

Open-Loop Voltage Gain (A

)

= 0V

min

Input Offset Voltage

= 0V

0.35

1.0

1.2

1.5

max

Average Offset Voltage Drift

= 0V

max

Input Bias Current

= 0V

17

max

Input Bias Current Drift

= 0V

100

nA/

max

Input Offset Current

= 0V

0.5

1.0

1.5

1.8

max

Input Offset Current Drift

= 0V

nA/

max

INPUT
Common-Mode Input Range (CMIR)

(5)

3.2

3.0

2.9

2.8

min

Common-Mode Rejection (CMR)

1V, Input Referred

100

min

Input Impedance

Differential-Mode

= 0V

6 || 2

|| pF

typ

Common-Mode

= 0V

2.9 || 1

|| pF

typ

OUTPUT
Output Voltage Swing

400

Load

3.5

3.2

3.1

3.0

min

100

Load

3.3

3.0

2.8

min

Current Output, Sourcing

= 0V

min

Current Output, Sinking

= 0V

60

min

Closed-Loop Output Impedance

G = +10, f = 100kHz

0.008

typ

POWER SUPPLY
Specified Operating Voltage

typ

Maximum Operating Voltage

max

Max Quiescent Current

12.4

12.9

13.9

max

Min Quiescent Current

12.4

11.9

11.9

min

Power Supply Rejection Ratio

+PSRR, PSRR

| = 4.5 to 5.5, Input Referred

min

THERMAL CHARACTERISTICS
Specified Operating Range: U, N Package

40 to +85

typ

Thermal Resistance,

Junction-to-Ambient

8-Pin, SO-8

125

C/W

typ

5-Pin, SOT23

150

C/W

typ

NOTES: (1) Test Levels: (A) 100% tested at 25

C. Over temperature limits by characterization and simulation. (B) Limits set by characterization and simulation.

C guaranteed specifications. (3) Junction temperature = ambient at low temperature

limit: junction temperature = ambient +23

C at high temperature limit for over temperature guaranteed specifications. (4) Current is considered positive out-of-node.

is the input common-mode voltage. (5) Tested < 3dB below minimum specified CMR at

CMIR limits.

OPA686

The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes
no responsibility for the use of this information, and all use of such information shall be entirely at the user's own risk. Prices and specifications are subject to change
without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant
any BURR-BROWN product for use in life support devices and/or systems.

PIN CONFIGURATIONS

Inverting Input

Non-Inverting Input

DNC

Output

DNC: Do Not Connect

NC: No Connection

Output

Non-Inverting Input

Inverting Input

Pin Orientation/Package Marking

A86

Top View

SOT23-5

Top View

SO-8

PACKAGE
DRAWING

TEMPERATURE

PACKAGE

ORDERING

TRANSPORT

PRODUCT

PACKAGE

NUMBER

(1)

RANGE

MARKING

NUMBER

(2)

MEDIA

OPA686U

SO-8 Surface Mount

182

C to +85

OPA686U

Rails

OPA686U/2K5

Tape and Reel

OPA686N

5-Lead SOT23-5

331

C to +85

A86

OPA686N/250

Tape and Reel

OPA686N/3K

Tape and Reel

NOTES: (1) For detailed drawing and dimension table, please see end of data sheet, or Appendix C of Burr-Brown IC Data Book. (2) Models with a slash (/ ) are
available only in Tape and Reel in the quantities indicated (e.g., /2K5 indicates 2500 devices per reel). Ordering 2500 pieces of "OPA686U/2K5" will get a single
2500-piece Tape and Reel. For detailed Tape and Reel mechanical information, refer to Appendix B of Burr-Brown IC Data Book.

PACKAGE/ORDERING INFORMATION

ABSOLUTE MAXIMUM RATINGS

Power Supply ...............................................................................

6.5V

Internal Power Dissipation ...................................... See Thermal Analysis
Differential Input Voltage ..................................................................

1.2V

Input Voltage Range ............................................................................

Storage Temperature Range: U, N ................................ 40

C to +125

Lead Temperature (soldering, 10s) .............................................. +300

Junction Temperature (T

) ........................................................... +175

ELECTROSTATIC
DISCHARGE SENSITIVITY

This integrated circuit can be damaged by ESD. Burr-Brown
recommends that all integrated circuits be handled with appropriate
precautions. Failure to observe proper handling and installation
procedures can cause damage.

ESD damage can range from subtle performance degradation to
complete device failure. Precision integrated circuits may be more
susceptible to damage because very small parametric changes
could cause the device not to meet its published specifications.

OPA686

TYPICAL PERFORMANCE CURVES: V

At T

= +25

C, G = +10, R

= 453

, and R

= 100

, unless otherwise noted.

NON-INVERTING SMALL-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

Normalized Gain (3dB/div)

0.5

100

500

G = +50

See Figure 1

= 50

= 0.2Vp-p

G = +20

G = +7

G = +10

NON-INVERTING LARGE-SIGNAL

FREQUENCY RESPONSE

Frequency (MHz)

Gain (3dB/div)

0.5

100

500

= 50

G = +10V/V

= 0.2Vp-p

= 1Vp-p

= 2Vp-p

= 5Vp-p

See Figure 1

INVERTING LARGE-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Gain (3dB/div)

0.1

100

500

= R

= 50

G = 20V/V

= 0.2Vp-p

= 1Vp-p

= 2Vp-p

= 5Vp-p

See Figure 2

100

1.5

1.0

0.5

1.0

1.5

NON-INVERTING PULSE RESPONSE

Time (5ns/div)

Output Voltage (100mV/div)

Output Voltage (500mV/div)

G = +10V/V

Large Signal ±1V

Small Signal ±100mV

Right Scale

Left Scale

See Figure 1

100

1.5

1.0

0.5

1.0

1.5

INVERTING PULSE RESPONSE

Time (5ns/div)

Output Voltage (100mV/div)

Output Voltage (500mV/div)

G = 20V/V

Large Signal ±1V

Small Signal ±100mV

Right Scale

Left Scale

See Figure 2

INVERTING SMALL-SIGNAL FREQUENCY RESPONSE

Frequency (MHz)

Normalized Gain (3dB/div)

0.5

100

500

= R

= 50

= 0.2Vp-p

G = 12

G = 50

G = 20

See Figure 2

OPA686

TYPICAL PERFORMANCE CURVES: V

(CONT)

At T

= +25

C, G = +10, R

= 453

, and R

= 100

, unless otherwise noted. See Figure 1.

140

120

100

POWER SUPPLY and OUTPUT CURRENT

vs TEMPERATURE

Temperature (

100

125

Power Supply Current (mA)

Output Current (mA)

Power Supply Current

Output Current Sourcing

Output Current Sinking

1.3

1.1

0.9

0.7

0.5

0.3

0.1

INPUT DC ERRORS vs TEMPERATURE

Temperature (

100

125

(mV)

Input Bias and Input Offset Current (

Input Bias Current

Offset Voltage

Input Offset Current

120

150

180

210

240

270

OPEN-LOOP GAIN and PHASE

Frequency (Hz)

100

10M

100M

10k

100k

Open-Loop Gain (10dB/div)

Open-Loop Phase (30°/div)

| A

1.0

0.1

0.01

0.001

CLOSED-LOOP OUTPUT IMPEDANCE

Frequency (Hz)

10M

100M

10k

100k

Output Impedance (

)

OPA686

450

DIFFERENTIAL and COMMON-MODE

INPUT IMPEDANCE

Frequency (Hz)

100

10M

10k

100k

Input Impedance (

)

Common-Mode

Differential

110

100

CMRR and PSRR

Frequency (Hz)

100

10M

100M

10k

100k

Power Supply Rejection Ratio (dB)

CMRR

+PSRR

PSRR

OPA686

APPLICATIONS INFORMATION

WIDEBAND, NON-INVERTING OPERATION

The OPA686 provides a unique combination of features--
low input voltage noise along with a very low distortion
output stage--to give one of the highest dynamic range op
amps available. Its very high Gain Bandwidth Product (GBP)
can be used either to deliver high signal bandwidths at high
gains, or to deliver very low distortion signals at moderate
frequencies and lower gains. To achieve the full perfor-
mance of the OPA686, careful attention to PC board layout
and component selection is required as discussed in the
remaining sections of this data sheet.

Figure 1 shows the non-inverting gain of +10 circuit used as
the basis of the Electrical Specifications and most of the
Typical Performance Curves. Most of the curves were char-
acterized using signal sources with 50

driving impedance,

and with measurement equipment presenting a 50

load

impedance. In Figure 1, the 50

shunt resistor at the V

terminal matches the source impedance of the test generator,
while the 50

series resistor at the V

terminal provides a

matching resistor for the measurement equipment load.
Generally, data sheet voltage swing specifications are at the
output pin (V

in Figure 1), while output power (dBm)

specifications are at the matched 50

load. The total 100

load at the output, combined with the 503

total feedback

network load, presents the OPA686 with an effective output
load of 83

for the circuit of Figure 1.

Voltage feedback op amps, unlike current feedback designs,
can use a wide range of resistor values to set their gain. The
circuit of Figure 1, and the specifications at other gains, use
the constraint that R

should always be set to 50

and R

adjusted to get the desired gain. Using this guideline will
guarantee that the noise added at the output due to Johnson
noise of the resistors will not significantly increase the total
noise over that due to the 1.3nV/

Hz input voltage noise for

the op amp itself.

WIDEBAND, INVERTING GAIN OPERATION

Operating the OPA686 as an inverting amplifier has several
benefits and is particularly appropriate when a matched
input impedance is required. Figure 2 shows the inverting
gain circuit used as the basis of the inverting mode Typical
Performance Curves.

FIGURE 1. Non-Inverting, G = +10 Specification and Test

Circuit.

FIGURE 2. Inverting, G = 20 Characterization Circuit.

Driving this circuit from a 50

source, and constraining the

gain resistor (R

) to equal 50

, will give both a signal

bandwidth and noise advantage. R

acts as both the input

termination resistor and the gain setting resistor for the
circuit. Although the signal gain (V

) for the circuit of

Figure 2 is double that for Figure 1, the noise gains are in
fact equal when the 50

source resistor is included. This

has the interesting effect of doubling the equivalent GBP of
the amplifier. This can be seen in comparing the G = +10
and G = 20 small-signal frequency response curves. Both
show approximately 250MHz bandwidth, but the inverting
configuration of Figure 2 gives 6dB higher signal gain. If
the signal source is actually the low impedance output of
another amplifier, R

should be increased to the minimum

load resistance value allowed for that amplifier and R

should be adjusted to achieve the desired gain. For stable
operation of the OPA686, it is critical that this driving
amplifier show a very low output impedance at frequencies
beyond the expected closed-loop bandwidth for the OPA686.

WIDEBAND, HIGH SENSITIVITY, TRANSIMPEDANCE
DESIGN

The high Gain Bandwidth Product (GBP) and the low input
voltage and current noise for the OPA686 make it an ideal
wideband transimpedance amplifier for low to moderate
transimpedance gains. Very high transimpedance gains
(> 100k

) will benefit from the low input noise current of

a FET-input op amp such as the OPA655. Unity gain
stability in the op amp is not required for application as a

OPA686

+5V

0.1µF

6.8µF

453

Source

Load

0.1µF

OPA686

+5V

6.8µF

0.1µF

6.8µF

0.1µF

Source

Load

OPA686

transimpedance amplifier. One transimpedance design ex-
ample is shown on the front page of the data sheet. Designs
that require high bandwidth from a large area (high capaci-
tance) detector with relatively low transimpedance gain will
benefit from the low input voltage noise offered by the
OPA686. This input voltage noise will be peaked up over
frequency at the output by the diode source capacitance, and
can, in many cases, become the limiting factor to input
sensitivity. The key elements of the design are the expected
diode capacitance (C

) with the reverse bias voltage (V

)

applied, the desired transimpedance gain, R

, and the GBP

of the OPA686 (1600MHz). Figure 3 shows a design using
a 50pF source capacitance diode and a 10k

transimpedance

gain. With these three variables set (and including the
parasitic input capacitance for the OPA686 added to C

), the

feedback capacitor value (C

) may be set to control the

frequency response.

Where:

= Equivalent input noise current if the output noise is

bandlimited to F < 1/(2

)

= Input current noise for the op amp inverting input

= Input voltage noise for the op amp

= Diode capacitance

= Bandlimiting frequency in Hz (usually a post filter

prior to further signal processing)

Evaluating this expression up to the feedback pole frequency
at 15.5MHz for the circuit of Figure 3, gives an equivalent
input noise current of 6.4pA/

Hz. This is much higher than

the 1.8pA/

Hz for just the op amp itself. This result is being

dominated by the last term in the equivalent input noise
expression. It is essential in this case to use a low voltage
noise op amp. For example, if a slightly higher input noise
voltage, but otherwise identical op amp were used instead of
the OPA686 in this application (say 2.0nV/

Hz ), the total

input-referred current noise would increase to 9.5pA/

Hz.

LOW GAIN COMPENSATION FOR IMPROVED SFDR

Where a low gain is desired, and inverting operation is
acceptable, a new external compensation technique may be
used to retain the full slew rate and noise benefits of the
OPA686 while giving increased loop gain and the associ-
ated improvement in distortion offered by the decompen-
sated architecture. This technique shapes the loop gain for
good stability while giving an easily controlled second-
order low pass frequency response. Considering only the
noise gain (non-inverting signal gain) for the circuit of
Figure 4, the low frequency noise gain, (NG

) will be set by

the resistor ratios while the high frequency noise gain (NG

)

will be set by the capacitor ratios. The capacitor values set
both the transition frequencies and the high frequency noise
gain. If this noise gain, determined by NG

= 1+C

, is set

to a value greater than the recommended minimum stable
gain for the op amp and the noise gain pole, set by 1/R

is placed correctly, a very well controlled, 2nd-order low
pass frequency response will result.

FIGURE 3. Wideband, Low Noise, Transimpendance

Amplifier.

To achieve a maximally flat 2nd-order Butterworth fre-
quency response, the feedback pole should be set to:

1/(2

) =

(GBP/(4

))

Adding the common-mode and differential mode input ca-
pacitance (1.0 + 2.0)pF to the 50pF diode source capacitance
of Figure 3, and targeting a 10k

transimpedance gain using

the 1600MHz GBP for the OPA686, will require a feedback
pole set to 15.5MHz. This will require a total feedback
capacitance of 1.0pF. Typical surface-mount resistors have
a parasitic capacitance of 0.2pF, leaving the required 0.8pF
value shown in Figure 3 to get the required feedback pole.

This will give an approximate 3dB bandwidth equal to:

3dB

(GBP/2

)Hz

The example of Figure 3 will give approximately 23MHz
flat bandwidth using the 0.8pF feedback compensation.

If the total output noise is bandlimited to a frequency less
than the feedback pole frequency, a very simple expression
for the equivalent input noise current can be derived as:

FIGURE 4. Broadband Low Gain Inverting External Com-

pensation.

500

27pF

OPA686

+5V

2.9pF

250

+ 4

(

)

10k

Supply Decoupling

Not Shown

50pF

OPA686

+5V

0.8pF

OPA686

Frequency (MHz)

Gain (3dB/div)

100

500

170MHz

To choose the values for both C

and C

, two parameters and

only three equations need to be solved. The first parameter
is the target high frequency noise gain NG

, which should be

greater than the minimum stable gain for the OPA686. Here,
a target NG

of 10.5 will be used. The second parameter is

the desired low frequency signal gain, which also sets the
low frequency noise gain NG

. To simplify this discussion,

we will target a maximally flat second-order low pass
Butterworth frequency response (Q = 0.707). The signal
gain of 2 shown in Figure 4 will set the low frequency noise
gain to NG

= 1 + R

(= 3 in this example). Then, using

only these two gains and the GBP for the OPA686
(1600MHz), the key frequency in the compensation can be
determined as:

Physically, this Z

(10.6MHz for the values shown above) is

set by 1/(2

· R

+ C

)) and is the frequency at which the

rising portion of the noise gain would intersect unity gain if
projected back to 0dB gain. The actual zero in the noise gain
occurs at NG

· Z

and the pole in the noise gain occurs at

· Z

. Since GBP is expressed in Hz, multiply Z

by 2

and use this to get C

by solving:

Finally, since C

and C

set the high frequency noise gain,

determine C

by [Using NG

= 10.5]:

The resulting closed-loop bandwidth will be approximately
equal to:

For the values shown in Figure 4, the f

3dB

will be approxi-

mately 130MHz. This is less than that predicted by simply
dividing the GBP product by NG

. The compensation

network controls the bandwidth to a lower value while
providing the full slew rate at the output and an excep-
tional distortion performance due to increased loop gain at
frequencies below NG

· Z

. The capacitor values shown

in Figure 4 are calculated for NG

= 3 and NG

= 10.5 with

no adjustment for parasitics.

Figure 5 shows the measured frequency response for the
circuit of Figure 4. This shows the expected gain of
2 (6dB) with exceptional flatness through 70MHz and a
3dB bandwidth of 170MHz. Measured distortion into a
100

load shows > 5dB improvement through 20MHz over

the performance shown in the Typical Performance Curves.
Into a 500

load, the 5MHz, 2Vp-p, 2nd harmonic improves

from 90dBc to 96dBc.

GBP

1 2

FIGURE 5. G = 2 Frequency Response with External

Compensation.

LOW NOISE FIGURE, HIGH DYNAMIC RANGE "IF"
AMPLIFIER

The low input noise voltage of the OPA686 and its high
two-tone intercept can be used to good advantage as a fixed
gain IF amplifier. While input noise figures in the 10dB
range (for a matched 50

input) are easily achieved with

just OPA686 alone, Figure 6 shows a technique which
reduces the noise figure even further while providing a
broadband, low gain IF amplifier stage using the OPA686.

FIGURE 6. Low Noise Figure IF Amplifier.

Bringing the signal in through a step-up transformer to the
inverting input gain resistor has several advantages for the
OPA686. First, grounding the non-inverting input elimi-
nates the contribution of the non-inverting input current
noise to the output noise. Secondly, the non-inverting
input voltage noise of the op amp is actually attenuated if
reflected to the input side of R

. Using the 1:2 (turns ratio)

step up transformer reflects the 50

source impedance at

· R

(

)

GBP

(= 2.86pF)

(= 27.2pF)

(= 130MHz)

OPA686

+5V

200

2pF

20pF

Source

Load

1:2

OPA686

the primary through to the secondary as a 200

source

impedance and likewise, the 200

resistor is reflected

through to the transformer primary as a 50

input match-

ing impedance. The noise gain (NG) to the amplifier
output is then 1+ 1000/400 = 3.5V/V. Taking the op amp's
1.3nV/

Hz input voltage noise times this noise gain to the

output, then reflecting this noise term to the input side of
the R

resistor, divides it by 5. This gives a net gain of 0.7

for the non-inverting input voltage noise when reflected to
the input point for the op amp circuit. This is further
reduced when referred back to the transformer primary.

The 14dB gain to the matched load for the circuit of Figure
6 is precisely controlled (

0.2dB) and gives a 6dB noise

figure at the input of the transformer. The DC noise gain for
this circuit (3.5) is below the specified minimum stable gain.
This will improve the distortion performance at frequencies
below 20MHz from those shown in the Typical Performance
Curves. Adding the inverting compensation capacitors holds
this configuration stable as described in the previous section.
Measured results show 140MHz small-signal bandwidth for
the circuit of Figure 6 with

0.1dB flatness through 50MHz.

The OPA686 will easily deliver a 2Vp-p ADC full-scale
input at the matched 50

load. Two-tone testing at 20MHz

for the circuit of Figure 6 (1Vp-p for each test tone) shows
that the two-tone intermodulation intercept has improved to
40dBm versus the 35dBm shown in the Typical Perfor-
mance Curves, giving a 72dBc SFDR for the two 4dBm test
tones at the load .

DESIGN-IN TOOLS

DEMONSTRATION BOARDS

Two PC boards are available to assist in the initial evaluation
of circuit performance using the OPA686 in its two package
styles. Both of these are available free as an unpopulated PC
board delivered with descriptive documentation. The sum-
mary information for these boards is shown in the table
below.

BOARD

LITERATURE

PART

REQUEST

PRODUCT

PACKAGE

NUMBER

OPA686U

8-Pin SO-8

DEM-OPA68xU

MKT-351

OPA686N

5-Lead SOT23-5

DEM-OPA6xxN

MKT-348

Contact the Burr-Brown applications support line to request
any of these boards.

MACROMODELS AND APPLICATIONS SUPPORT

Computer simulation of circuit performance using SPICE is
often useful when analyzing the performance of analog
circuits and systems. This is particularly true for video and
RF amplifier circuits where parasitic capacitance and induc-
tance can have a major effect on circuit performance. A
SPICE model for the OPA686 is available through either the
Burr-Brown Internet web page (http://www.burr-brown.com)
or as one model on a disk from the Burr-Brown Applications
department (1-800-548-6132). The Applications department

is also available for design assistance at this number. These
models do a good job of predicting small-signal AC and
transient performance under a wide variety of operating
conditions. They do not do as well in predicting the har-
monic distortion characteristics. These models do not at-
tempt to distinguish between the package types in their
small-signal AC performance.

OPERATING SUGGESTIONS

SETTING RESISTOR VALUES TO MINIMIZE NOISE

The OPA686 provides a very low input noise voltage while
requiring a low 12mA quiescent current. To take full advan-
tage of this low input noise, careful attention to the other
possible noise contributors is required. Figure 7 shows the
op amp noise analysis model with all the noise terms
included. In this model, all the noise terms are taken to be
noise voltage or current density terms in either nV/

Hz or

pA/

Hz.

The total output spot noise voltage can be computed as the
square root of the squared contributing terms to the output
noise voltage. This computation adds all the contributing
noise powers at the output by superposition, then takes the
square root to get back to a spot noise voltage. Equation 1
shows the general form for this output noise voltage using
the terms shown in Figure 7.

Equation 1

Dividing this expression by the noise gain (NG = 1+R

)

will give the equivalent input-referred spot noise voltage at
the non-inverting input as shown in Equation 2.

Equation 2

4kT

OPA686

4kT = 1.6E 20J

at 290°K

4kTR

(

)

+ 4

kTR

(

)

(

)

+ 4

kTR

(

)

+ 4

kTR

+ 4

kTR

FIGURE 7. Op Amp Noise Analysis Model.

OPA686

Inserting high resistor values into Eq. 2 can quickly domi-
nate the total equivalent input referred noise. A 105

source

impedance on the non-inverting input will add a Johnson
voltage noise term equal to that of the amplifier itself. As a
simplifying constraint, set R

= R

in Eq. 2 and assume an

/2 source impedance at the non-inverting input (where R

is the signal's source impedance with another matching R

to ground on the non-inverting input). This results in Eq. 3,
where NG > 10 has been assumed to further simplify the
expression.

Equation 3

Evaluating this expression for R

= 50

will give a total

equivalent input noise of 1.7nV/

Hz. Note that the NG has

dropped out of this expression. This is valid only for NG > 10
as will typically be required by stability considerations.

FREQUENCY RESPONSE CONTROL

Voltage feedback op amps exhibit decreasing closed-loop
bandwidth as the signal gain is increased. In theory, this
relationship is described by the Gain Bandwidth Product
(GBP) shown in the specifications. Ideally, dividing GBP by
the non-inverting signal gain (also called the Noise Gain, or
NG) will predict the closed-loop bandwidth. In practice, this
only holds true when the phase margin approaches 90

, as it

does in high gain configurations. At low gains (increased
feedback factor), most high speed amplifiers will exhibit a
more complex response with lower phase margin. The
OPA686 is compensated to give a maximally flat 2nd-order
Butterworth closed-loop response at a non-inverting gain of
+10 (Figure 1). This results in a typical gain of +10 band-
width of 250MHz, far exceeding that predicted by dividing
the 1600MHz GBP by 10. Increasing the gain will cause the
phase margin to approach 90

and the bandwidth to more

closely approach the predicted value of (GBP/NG). At a gain
of +40, the OPA686 will show the 40MHz bandwidth
predicted using the simple formula and the typical GBP of
1600MHz.

Inverting operation offers some interesting opportunities to
increase the available GBP. When the source impedance is
matched by the gain resistor (Figure 2), the signal gain is
(1+R

) while the noise gain for bandwidth purposes is

(1 + R

/2R

). This cuts the noise gain almost in half, increas-

ing the minimum stable gain for inverting operation under
these condition to 12 and the equivalent GBP to 3.2GHz.

DRIVING CAPACITIVE LOADS

One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an A/D converter, including
additional external capacitance which may be recommended
to improve A/D linearity. A high speed, high open-loop gain
amplifier like the OPA686 can be very susceptible to de-
creased stability and closed-loop response peaking when a

capacitive load is placed directly on the output pin. When
the amplifier's open-loop output resistance is considered,
this capacitive load introduces an additional pole in the
signal path that can decrease the phase margin. Several
external solutions to this problem have been suggested.
When the primary considerations are frequency response
flatness, pulse response fidelity and/or distortion, the sim-
plest and most effective solution is to isolate the capacitive
load from the feedback loop by inserting a series isolation
resistor between the amplifier output and the capacitive
load. This does not eliminate the pole from the loop re-
sponse, but rather shifts it and adds a zero at a higher
frequency. The additional zero acts to cancel the phase lag
from the capacitive load pole, thus increasing the phase
margin and improving stability.

The Typical Performance Curves show the recommended
R

vs Capacitive Load and the resulting frequency response

at the load. Parasitic capacitive loads greater than 2pF can
begin to degrade the performance of the OPA686. Long PC
board traces, unmatched cables, and connections to multiple
devices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA686 output pin
(see Board Layout Guidelines).

The criterion for setting this R

resistor is a maximum

bandwidth, flat frequency response at the load. For the
OPA686 operating in a gain of +10, the frequency response
at the output pin is very flat to begin with, allowing relatively
small values of R

to be used for low capacitive loads. As the

signal gain is increased, the unloaded phase margin will also
increase. Driving capacitive loads at higher gains will re-
quire lower R

values than those shown for a gain of +10.

DISTORTION PERFORMANCE

The OPA686 is capable of delivering an exceptionally low
distortion signal at high frequencies over a wide range of
gains. The distortion plots in the Typical Performance Curves
show the typical distortion under a wide variety of condi-
tions. Most of these plots are limited to 110dB dynamic
range. The OPA686's distortion, driving a 500

load does

not rise above 90dBc until either the signal level exceeds
2.0Vp-p and/or the fundamental frequency exceeds 5MHz.
Distortion in the audio band is < 120dBc.

Generally, until the fundamental signal reaches very high
frequencies or powers, the 2nd harmonic will dominate the
distortion with negligible a 3rd harmonic component. Focus-
ing then on the 2nd harmonic, increasing the load impedance
improves distortion directly. Remember that the total load
includes the feedback network; in the non-inverting configu-
ration, this is sum of R

+ R

, while in the inverting

configuration, it is just R

(Figures 1 and 2). Increasing

output voltage swing increases harmonic distortion directly.
A 6dB increase in output swing will generally increase the
2nd harmonic 12dB and the 3rd harmonic 18dB. Increasing
the signal gain will also increase the 2nd harmonic distor-
tion. Again, a 6dB increase in gain will increase the 2nd and
3rd harmonic by approximately 6dB even with constant

( )

+ 5

(

)

+ 4

OPA686

output power and frequency. Finally, the distortion increases
as the fundamental frequency increases due to the rolloff in
the loop gain with frequency. Conversely, the distortion will
improve going to lower frequencies down to the dominant
open-loop pole at approximately 100kHz. Starting from the
82dBc 2nd harmonic for a 5MHz, 2Vp-p fundamental into
a 200

load at G = +10 (from the Typical Performance

Curves), the 2nd harmonic distortion for frequencies lower
than 100kHz will be approximately 82dBc 20log(5MHz/
100kHz) = 116dBc.

The OPA686 has extremely low 3rd-order harmonic distor-
tion. This also gives a high two-tone, 3rd-order
intermodulation intercept as shown in the Typical Perfor-
mance Curves. This intercept curve is defined at the 50

load when driven through a 50

matching resistor to allow

direct comparisons to RF MMIC devices. This matching
network attenuates the voltage swing from the output pin to
the load by 6dB. If the OPA686 drives directly into the input
of a high impedance device, such as an ADC, the 6dB
attenuation is not taken. Under these conditions, the inter-
cept will increase by a minimum 6dBm. The intercept is
used to predict the intermodulation spurious for two, closely-
spaced frequencies. If the two test frequencies, f

and f

, are

specified in terms of average and delta frequency, f

+ f

)/2 and

f = |f

|/2, the two 3rd-order, close-in

spurious tones will appear at f

3 ·

f. The difference

between two equal test-tone power levels and these
intermodulation spurious power levels is given by

dBc = 2 · (IM3 P

) where IM3 is the intercept taken from

the Typical Performance Curve and P

is the power level in

dBm at the 50

load for one of the two closely-spaced test

frequencies. For instance, at 5MHz the OPA686 at a gain of
+10 has an intercept of 48dBm at a matched 50

load. If the

full envelope of the two frequencies needs to be 2Vp-p, this
requires each tone to be 4dBm. The 3rd-order intermodulation
spurious tones will then be 2 · (48 4) = 88dBc below the
test-tone power level (84dBm). If this same 2Vp-p, two-
tone envelope were delivered directly into the input of an
ADC--without the matching loss or the loading of the 50

network--the intercept would increase to at least 54dBm.
With the same signal and gain conditions, but now driving
directly into a light load, the spurious tones will then be at
least 2 · (54 4) = 100dBc below the 4dBm test-tone power
levels centered on 5MHz.

DC ACCURACY AND OFFSET CONTROL

The OPA686 can provide excellent DC signal accuracy due
to its high open-loop gain, high common-mode rejection,
high power supply rejection, and low input offset voltage
and bias current offset errors. To take full advantage of its
low

1.5mV input offset voltage, careful attention to input

bias current cancellation is also required. The low noise
input stage of the OPA686 has a relatively high input bias
current (10

A typical into the pins) but with a very close

match between the two input currents--typically

100nA

input offset current. The total output offset voltage may be
reduced considerably by matching the source impedances
looking out of the two inputs. For example, one way to add

bias current cancellation to the circuit of Figure 1 would be
to insert a 20

series resistor into the non-inverting input

from the 50

terminating resistor. When the 50

source

resistor is DC-coupled, this will increase the source resis-
tances for the non-inverting input bias current to 45

. Since

this is now equal to the resistance looking out of the
inverting input (R

|| R

), the circuit will cancel the gains for

the bias currents to the output leaving only the offset current
times the feedback resistor as a residual DC error term at the
output. Using the 453

feedback resistor, this output error

will now be less than

0.9

A · 453

0.4mV over the full

temperature range.

A fine-scale, output offset null, or DC operating point
adjustment, is often required. Numerous techniques are
available for introducing a DC offset control into an op amp
circuit. Most of these techniques eventually reduce to setting
up a DC current through the feedback resistor. One key
consideration to selecting a technique is to insure that it has
a minimal impact on the desired signal path frequency
response. If the signal path is intended to be non-inverting,
the offset control is best applied as an inverting summing
signal to avoid interaction with the signal source. If the
signal path is intended to be inverting, applying the offset
control to the non-inverting input can be considered. For a
DC-coupled inverting input signal, this DC offset signal will
set up a DC current back into the source that must be
considered. An offset adjustment placed on the inverting op
amp input can also change the noise gain and frequency
response flatness. Figure 8 shows one example of an offset
adjustment for a DC-coupled signal path that will have
minimum impact on the signal frequency response. In this
case, the input is brought into an inverting gain resistor with
the DC adjustment an additional current summed into the
inverting node. The resistor values setting this offset adjust-
ment are much larger than the signal path resistors. This will
insure that this adjustment has minimal impact on the loop
gain and hence, the frequency response.

FIGURE 8. DC-Coupled, Inverting Gain of 20, with

Output Offset Adjustment.

±200mV Output Adjustment

= = 20

Supply Decoupling

Not Shown

0.1µF

20k

10k

0.1µF

+5V

OPA686

+5V

OPA686

THERMAL ANALYSIS

The OPA686 will not require heatsinking or airflow in most
applications. Maximum desired junction temperature will
set the maximum allowed internal power dissipation as
described below. In no case should the maximum junction
temperature be allowed to exceed +175

Operating junction temperature (T

) is given by T

+ P

. The total internal power dissipation (P

) is the sum of

quiescent power (P

) and additional power dissipated in

the output stage (P

) to deliver load power. Quiescent

power is simply the specified no-load supply current times
the total supply voltage across the part. P

will depend on

the required output signal and load but would, for a grounded
resistive load, be at a maximum when the output is fixed at
a voltage equal to 1/2 either supply voltage (for equal bipolar
supplies). Under this worst-case condition, P

= V

/(4 ·

) where R

includes feedback network loading.

Note that it is the power in the output stage and not in the
load that determines internal power dissipation.

As a worst-case example, compute the maximum T

using an

OPA686N (SOT23-5 package) in the circuit of Figure 1
operating at the maximum specified ambient temperature of
+85

C and driving a grounded 100

load at +2.5V

= 10V (13.9mA) + 5

/(4 · (100

|| 500

)) = 214mW

Maximum T

= +85

C + (0.21W · 150

C/W) = 117

BOARD LAYOUT

Achieving optimum performance with a high frequency
amplifier like the OPA686 requires careful attention to
board layout parasitics and external component types. Rec-
ommendations that will optimize performance include:

a) Minimize parasitic capacitance to any AC ground for
all of the signal I/O pins. Parasitic capacitance on the
output and inverting input pins can cause instability: on the
non-inverting input, it can react with the source impedance
to cause unintentional bandlimiting. To reduce unwanted
capacitance, a window around the signal I/O pins should be
opened in all of the ground and power planes around those
pins. Otherwise, ground and power planes should be unbro-
ken elsewhere on the board.

b) Minimize the distance (< 0.25") from the power
supply pins to high frequency 0.1

F decoupling capaci-

tors. At the device pins, the ground and power plane layout
should not be in close proximity to the signal I/O pins. Avoid
narrow power and ground traces to minimize inductance
between the pins and the decoupling capacitors. The power
supply connections should always be decoupled with these
capacitors. Larger (2.2

F to 6.8

F) decoupling capacitors,

effective at lower frequency, should also be used on the
main supply pins. These may be placed somewhat farther
from the device and may be shared among several devices in
the same area of the PC board.

c) Careful selection and placement of external compo-
nents will preserve the high frequency performance of
the OPA686. Resistors should be a very low reactance type.

Surface-mount resistors work best and allow a tighter over-
all layout. Metal-film and carbon composition, axially-leaded
resistors can also provide good high frequency performance.
Again, keep their leads and PC board trace length as short as
possible. Never use wirewound type resistors in a high
frequency application. Since the output pin and inverting
input pin are the most sensitive to parasitic capacitance,
always position the feedback and series output resistor, if
any, as close as possible to the output pin. Other network
components, such as non-inverting input termination resis-
tors, should also be placed close to the package. Where
double-side component mounting is allowed, place the feed-
back resistor directly under the package on the other side of
the board between the output and inverting input pins. Even
with a low parasitic capacitance shunting the external resis-
tors, excessively high resistor values can create significant
time constants that can degrade performance. Good axial
metal-film or surface-mount resistors have approximately
0.2pF in shunt with the resistor. For resistor values > 1.5k

this parasitic capacitance can add a pole and/or a zero below
500MHz that can effect circuit operation. Keep resistor
values as low as possible consistent with load driving con-
siderations. It has been suggested here that a good starting
point for design would be set the R

be set to 50

. Doing

this will automatically keep the resistor noise terms low, and
minimize the effect of their parasitic capacitance.

d) Connections to other wideband devices on the board
may be made with short direct traces or through on-
board transmission lines. For short connections, consider
the trace and the input to the next device as a lumped
capacitive load. Relatively wide traces (50mils to 100mils)
should be used, preferably with ground and power planes
opened up around them. Estimate the total capacitive load
and set R

from the plot of recommended R

vs Capacitive

Load. Low parasitic capacitive loads (< 5pF) may not need
an R

since the OPA686 is nominally compensated to

operate with a 2pF parasitic load. Higher parasitic capacitive
loads without an R

are allowed as the signal gain increases

(increasing the unloaded phase margin). If a long trace is
required, and the 6dB signal loss intrinsic to a doubly-
terminated transmission line is acceptable, implement a
matched impedance transmission line using microstrip or
stripline techniques (consult an ECL design handbook for
microstrip and stripline layout techniques). A 50

environ-

ment is normally not necessary on board, and in fact, a
higher impedance environment will improve distortion as
shown in the distortion versus load plots. With a character-
istic board trace impedance defined based on board material
and trace dimensions, a matching series resistor into the
trace from the output of the OPA686 is used as well as a
terminating shunt resistor at the input of the destination
device. Remember also that the terminating impedance will
be the parallel combination of the shunt resistor and the
input impedance of the destination device; this total effec-
tive impedance should be set to match the trace impedance.
If the 6dB attenuation of a doubly-terminated transmission
line is unacceptable, a long trace can be series-terminated at
the source end only. Treat the trace as a capacitive load in
this case and set the series resistor value as shown in the plot

OPA686

of R

vs Capacitive Load. This will not preserve signal

integrity as well as a doubly-terminated line. If the input
impedance of the destination device is low, there will be
some signal attenuation due to the voltage divider formed by
the series output into the terminating impedance.

e) Socketing a high speed part like the OPA686 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an ex-
tremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA686
onto the board.

INPUT AND ESD PROTECTION

The OPA686 is built using a very high speed complemen-
tary bipolar process. The internal junction breakdown volt-
ages are relatively low for these very small geometry de-
vices. These breakdowns are reflected in the Absolute Maxi-
mum Ratings table. All device pins are protected with
internal ESD protection diodes to the power supplies as
shown in Figure 9.

External

Internal
Circuitry

FIGURE 9. Internal ESD Protection.

These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with

15V supply

parts driving into the OPA686), current-limiting series resis-
tors should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.

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